DESIGN OF ULTRA-WIDEBAND BANDPASS FILTERS WITH NARROW NOTCHED BANDS USING A RING RESONATOR

Transcription

1 Chuyên san Công nghệ thông tin và Truyền thông - Số 04 (4-2014) DESIGN OF ULTRA-WIDEBAND BANDPASS FILTERS WITH NARROW NOTCHED BANDS USING A RING RESONATOR Nguyen Tran Quang 1, Doan Minh Tan 1, Ta Chi Hieu 1 Abstract In this paper, compact ultra-wideband (UWB) bandpass filters with a/two notched bands are proposed based on a ring resonator. Broad passband and narrow notched bands with controlled even/odd-mode resonator frequencies can be adjusted conveniently by changing the characteristic impedance of the ring resonator. Some transmission zeros are introduced to improve the selectivity and harmonic suppression of the proposed ultra-wideband bandpass filters with notched bands. To verify the presented concept, three prototypes with 3-dB fractional bandwidth greater than 120 % are designed and fabricated. The results show that simple structure, good in/out-of-band performance can be achieved in the proposed ultra-wideband bandpass filters. Bài báo trình bày về mạch lọc dải thông băng siêu rộng với một và hai khe chặn can nhiễu trên cấu trúc đơn vòng cộng hưởng. Tần số dải thông và khe chặn can nhiễu có thể điều chỉnh thuận tiện bởi việc thay đổi đặc tính trở kháng của vòng cộng hưởng, một số điểm truyền 0 được tạo ra nhằm tăng tính chọn lọc của mạch lọc cũng như triệt tiêu các hài bậc cao. Để chứng minh tính đúng đắn của các kết quả nghiên cứu, một vài bộ lọc băng thông siêu rộng với các đặc tính tốt đã được thiết kế. 1. INTRODUCTION Recently, more and more attentions have been paid to the development of ultra-wideband (UWB) systems since Federal Communications Commission (FCC) s decision to permit the unlicensed operation band from 3.1 to 10.6 GHz in February 2002 [1]. As one of the most important microwave components, ultra-wideband bandpass filters (UWB-BPF) with high performance and low cost are imperatively needed in UWB communication systems. Various UWB filters employing multiple-mode resonators, complementary split-ring resonator (CSRR) and multilayer aperture-coupled patches are designed and analyzed [2]-[8]. In [9]-[16], cascaded low-pass/high-pass filters, electromagnetic (EM) loaded bandgap and T/Y-shaped resonators are used to extend the upper stopband bandwidth and improve the selectivity of the passband. In addition, some other UWB filters and differential filters are realized based on the transversal signal-interference concept in [17]-[24], by introducing intentionally a passband constructive interference and out-of-band signal energy cancellations to produce power transmission zeros, high-selectivity filtering responses and harmonic suppression can be achieved in these filter structures. In addition, due to the frequency band of the UWB indoor (1)Le Quy Don Technical University 96

2 Tạp chí Khoa học và Kỹ thuật - Học viện KTQS - Số 160 (4-2014) limit ( GHz) covers existing wireless communication systems, such as the 3.5-GHz band WiMax, GHz wireless local-area network (WLAN) and some 8.0-GHz band satellite-communication systems, it is desirable to introduce single or multiple notch bands to avoid the interferences from existing wireless communication systems in the design of UWB- BPF. A number of UWB filters with a/multi notched bands are investigated and discussed in [25]-[32]. However, little research has described the application of the ring resonator in the UWB filters with single/multi-notched bands based on transversal signal-interference concept. Lately, the authors have presented an effective and simple method for wideband filter design [36]-[38]. The UWB-BPF with notched band using stubs loaded multi-mode ring resonator was presented [39]. However, for those structure the attenuation in the upper stopband still remains high and the selectivity needs to be further improved. In this paper, compact UWB- BPF with a/two notched bands are proposed based on a ring resonator and open/shorted stubs. A broad passband with 3-dB fractional bandwidth greater than 120% can be realized with a three-quarter wavelength ring resonator and a shorted stub. Two transmission zeros are used to improve the selectivity and harmonic suppression based on transversal signal-interference concept. In addition, another two UWB-BPF with a/two notched bands for WiMax band, WLAN band and satellite-communication systems 8.0-GHz band are proposed using the threequarter wavelength ring resonator and an open stub. The bandwidth and center frequencies of the notched bands can be easily adjusted by changing the electrical length and characteristic impedance of the open stub. All the structures are designed and fabricated on the dielectric substrate with e r = 2.65, h = 0.5 mm, and tanδ = The design parameters obtained by Ansoft HFSS 10.0, and the measurement is accomplished using Agilent 8753 ES network analyzer. Good agreement can be found between the theoretical and measured results. 2. DESIGN OF PROPOSED UWB BPF WITH SINGLE CENTRALLY LOADED OPEN STUB Figs. 1 and show the top view and the circuit of the ultra-wideband bandpass filter with a three quarter wavelength ring resonator, two different transmission paths with electrical length 2θ 1 and θ 1, characteristic impedance Z 1 are connected ports 1, 2 symmetrically. A centrally loaded shorted stub with electrical length θ 2 and characteristic impedance Z 2 is attached to center of the path 1. Due to the symmetry of the ring resonator, the even/odd-mode equivalent circuits of the ring resonator are shown in Figs. 2 and [33] for theoretical analysis, respectively Even/Odd-Mode Analysis When the even/odd-mode signals are excited form port 1 to port 2, a virtual open/short stub appears at the center of the three quarter wavelength ring resonator, and the even/odd-mode input admittance Y ine /Y ino of the structures in Figs. 2- is given by (1) and (2). 97

3 Chuyên san Công nghệ thông tin và Truyền thông - Số 04 (4-2014) Figure 1. Proposed ultra-wideband bandpass filter with the three quarter wavelength ring resonator and a shorted stub Top view equivalent circuit. Figure 2. Even-mode equivalent circuit of the ring resonator odd-mode equivalent circuit of the ring resonator. Y ine = j tan θ 1/2 Z 1 + j tan θ 1 (Z 1 cot θ 2 )/2Z 2 Z 1 + (Z 2 1 tan θ 1 cot θ 2 )/2Z 2 (1) Y ino = j cot θ 1 Z 1 j cot θ 1/2 Z 1 (2) The resonance frequencies for the even/odd modes can be calculated when Y ine /Y ino = 0 or Z ine /Z ino = 0, and the even/odd-mode resonance frequencies for the ring resonator under weak coupling in Fig. 3 are shown in Fig. 3. The bandwidth for the ring resonator is mainly determined by the even-mode resonance frequencies f even1 and f even2, and the transmission zero located at the even-mode frequency f even3 near the upper end of the passband helps to improve the resonator selectivity. Moreover, another two odd-mode resonance frequencies f odd1 (θ 1 = ), f odd2 (θ 1 = ) can be realized, the odd-mode resonance frequency f odd2 is mainly caused by the phase difference of the two transmission paths at 2f 0 [20]-[22]. In addition, the even/odd-mode resonator frequencies versus θ 2 and different characteristic impedance Z 1 and Z 2 are shown in Figs. 3(c)-(e). And the odd-mode resonator frequencies (f odd1, f odd2 do not change with θ 2, Z 1 and Z 2. The even-mode resonator frequency f even1 98

4 Tạp chí Khoa học và Kỹ thuật - Học viện KTQS - Số 160 (4-2014) moves towards f 0 as Z 1 increases, while away from f 0 as θ 2, Z 2 increase. Moreover, the evenmode resonator frequencies f even2 and f even3 move towards f 0 as θ 2, Z 2 increase, while away from f 0 as Z 1 increases. In this way, the bandwidth for the passband of the UWB filter with the ring resonator can be controlled conveniently by changing the characteristic impedance Z 1, Z 2 and θ 2 when Z 0 and θ 1 are fixed (c) (d) (e) Figure 3. Equivalent circuit of the ring resonator under weak coupling S 21 under weak coupling, Z 1 = 60Ω, Z 2 = 70Ω, θ 2 = 20 0 (c) even/odd-mode resonator frequencies versus θ 2 at f 0, Z 1 = 60Ω, Z 2 = 70Ω(d) even/odd-mode resonator frequencies versus Z 1, Z 2 = 70Ω, θ 2 = 20 0 (e) even/odd-mode resonator frequencies versus Z 2, Z 1 = 60Ω, θ 2 = 20 0 (C 0 = 0.02pF, f 0 = 7.7GHz). 99

5 Chuyên san Công nghệ thông tin và Truyền thông - Số 04 (4-2014) Figure 4. Simulated Q L for the passband versus θ 2, Z 1 and Z 2. Versus Z 1, Z 2 = 70Ω versus Z 2, Z 1 = 60Ω(Z 0 = 50Ω, f 0 = 7.7GHz, θ 1 = 90 0 ). The dependence of the loaded quality factor Q L of the passband on θ 2, Z 1 and Z 2 is shown in Fig. 4. It can be seen that the Q L of the passband decreases as Z 1 increases, and increases as θ 2, Z 2 increase. In addition, the loaded quality factor Q L and the 3-dB bandwidth f 3dB for the passband can be related by [34]: Q L = f 0 / f 3dB (3) And the loaded quality factor Q L can be expressed as Q L = f 3dB (θ 2, Z 1, Z 2 ). Once Z 0 and θ 1 are determined, we can adjust θ 2, Z 1 and Z 2 to satisfy the demand of Q L. Obviously, the required 3-dB bandwidth f 3dB and the transmission characteristic for the passband can be simultaneously obtained and further optimized based on the above discussion Proposed Ultra-wideband Bandpass Filter with Single Ring Resonator Based on the above theoretical analysis, the final parameters for the filter circuit of Fig. 1 are given as follows: Z 0 = 50Ω, Z 1 = 60Ω, Z 2 = 70Ω, f 0 = 7.7 GHz,θ 1 = 90 0, θ 2 = In addition, the structure parameters for the ultra-wideband bandpass filter (15mm 12mm, 0.56λ g0 0.45λ g0 ) in Fig. 1 are listed below: L 1 = 10.35mm, L 2 = 2.1mm, W 0 = 1.37mm, W 1 = 1mm, W 2 = 0.8mm, d = 0.7mm. The photograph, measured results and simulated results are shown in Fig. 5, good agreement can be found between the measured and simulated results, the measured insertion loss is less than 0.95 db while the return loss is over 14 db from 3.7 GHz to 11.6 GHz (3-dB fractional bandwidth is approximately 123.4%, GHz). Three transmission zeros (located at 13, 14.6 and 15.7 GHz) are realized to improve the selectivity and harmonic suppression. Furthermore, over 15-dB upper stopband is achieved from GHz to 18.5 GHz (2.4f 0 ). The measured group delay is less than 0.20 ns from 2.7 GHz to 11.5 GHz. 100

6 Tạp chí Khoa học và Kỹ thuật - Học viện KTQS - Số 160 (4-2014) Figure 5. The photograph measured and simulated results of the UWB bandpass filter. Table 1. Comparisons of Measured Results for Several Ultra-wideband Bandpass Filter Structures. Table 1 illustrates the comparisons of measured results for several ultra-wideband bandpass filter structures. Compared with other UWB filters [3]-[17], the effective circuit size of the proposed UWB bandpass filter is only 0.19λ g0 0.19λ g0, and the bandwidth for the UWB bandpass filter based on the ring resonator is 123.4% compared with the other UWB filter structures based on the transversal signal-interaction concept in [18], [20]. Moreover, to extend the upper stopband, some lowpass/bandstop networkss can be cascaded to further improve the upper performance of the UWB filter with the single ring resonator [9]-[13]. 3. DESIGN OF PROPOSED ULTRA-WIDEBAND BANDPASS FILTERS WITH A/TWO NOTCHED BANDS 3.1. Proposed Ultra-wideband Bandpass Filter with A Notched Band To cancel the interference for WLAN signals ( GHz), the UWB bandpass filter with a notched band is further proposed as Figs. 6-, an shunt open stub with characteristic 101

7 Chuyên san Công nghệ thông tin và Truyền thông - Số 04 (4-2014) (c) (d) Figure 6. Proposed ultra-wideband bandpass filter with single notched band Top view equivalent circuit (c) even-mode equivalent circuit of the ring resonator (d) odd-mode equivalent circuit of the ring resonator (L 1 = 9.95mm, L 2 = 2.84mm, L 3 = 3.11mm, L 4 = 2.5mm, L 5 = 2.4mm, W 0 = 1.37mm, W 1 = 1.25mm, W 2 = 1.1mm, W 3 = 0.6mm, d = 0.7mm, 20mm 12mm, 0.74λ g0 0.45λ g0 ). impedance Z 3 electrical length θ 3 is connected in the center of the path 1. And the even/oddmode equivalent circuits for the ring resonator with the open/shorted stubs are shown in Figs. 6(c)-(d), the even-mode input admittance Y ine of Fig. 6(c) can be illustrated as Y ine = j tan θ 1/2 Z 1 + j tan θ 1 (Z 1 cot θ 2 )/2Z 2 Z 1 + (Z 2 1 tan θ 1 cot θ 2 )/2Z 2 (4) +(Z 1 tan θ 3 )/2Z 3 (Z1 2 tan θ 1 tan θ 3 )/2Z 3 And the odd-mode input admittance Y ino of Fig. 6(d) is the same as (2), similarly, when the Y ine /Z ine = 0, the even-mode resonator frequencies of Fig. 6(c) can be calculated. Fig. 7 shows the even/odd-mode resonator frequencies for the ring resonator with shorted/open stubs under weak coupling, and a transmission zero (f tz1 ) located at 5.6 GHz, and the third harmonic (3f tz1 ) of the transmission zero can be used to further improve the upper stopband. In addition, the even/odd-mode resonator frequencies for the ring resonator of Fig. 1 do not change compared with Fig The simulated frequency responses of the notched band versus θ 3 and Z 3 are shown in Figs.

8 Tạp chí Khoa học và Kỹ thuật - Học viện KTQS - Số 160 (4-2014) (c) (d) Figure 7. S 21 under weak coupling, C 0 = 0.02pF, Z 1 = 55Ω, Z 2 = 60Ω, Z 3 = 75Ω, θ 2 = 30 0, θ 3 = notched band versus θ 3, Z 1 = 55Ω, Z 2 = 60Ω, Z 3 = 75Ω, θ 2 = 30 0 (c) notched band versus Z 3, Z 1 = 55Ω, Z 2 = 60Ω, θ 2 = 30 0, θ 3 = (c) notched band versus θ 3 and Z 3, Z 1 = 55Ω, Z 2 = 60Ω, θ 2 = 30 0 (C 0 = 0.02pF, Z 0 = 50Ω, f 0 = 7.7GHz). 7-(d). One can see that the center frequency of the notched band decreases as θ 3 increases, and increases as Z 3 increases, thus the center frequency and the bandwidth of the notched band for WLAN band can be easily adjusted by changing θ 3 and Z 3 while Z 0, Z 1, Z 2, θ 1 and θ 2 are fixed Proposed Ultra-wideband Bandpass Filter with Two Notched Bands The UWB bandpass filter with two notched bands to cancel the interferences for 3.5- GHz band WiMax and some 8.0-GHz band satellite-communication systems are shown in Figs. 8-, a stepped impedance resonator [35] with characteristic impedance Z 3, Z 4 and electrical length θ 3, θ 4 is introduced to realize the two notched bands, and the even -mode 103

9 Chuyên san Công nghệ thông tin và Truyền thông - Số 04 (4-2014) (c) (d) Figure 8. Proposed ultra-wideband bandpass filter with dual notched bands Top view equivalent circuit (c) even-mode equivalent circuit of the ring resonator (d) odd-mode equivalent circuit of the ring resonator (L 1 = 9.95mm, L 2 = 2.84mm, L 3 = 3.16mm, L 4 = 3.6mm, L 5 = 2.4mm, L 6 = 3.5mm, L 7 = 1.5mm, L 8 = 2.5mm, W 0 = 1.37mm, W 1 = 1.25mm, W 2 = 1.1mm, W 3 = 1.4mm, W 4 = 0.5mm, d = 0.7mm, 20mm 12mm, 0.74λ g0 0.45λ g0 ) input admittance Y ine of Fig. 8(c) can be illustrated as Y ine = j tan θ 1/2 tan θ 1 (Z 1 cot θ 2 )/2Z 2 + j (5) Z 1 Z1 + (Z1 2 tan θ 1 cot θ 2 )/2Z 2 +(Z 1 Z 3 tan θ 4 + Z 1 Z 4 tan θ 3 )/(2Z 3 Z 4 2Z3 2 tan θ 3 tan θ 4 ) (Z1Z 2 3 tan θ 4 + Z1Z 2 4 tan θ 3 )/(2Z 3 Z 4 2Z3 2 tan θ 3 tan θ 4 ) And the odd-mode input admittance Yino of Fig. 6(d) is also the same as (2), similarly, when the Yine/Zine = 0, the even-mode resonator frequencies of Fig. 8(c) can be calculated. The even/odd-mode resonator frequencies for the ring resonator in Fig. 8 under weak coupling is shown as Fig. 9, two transmission zeros are located at 3.5 GHz (f tz1 ) and 8.0 GHz (f tz2 ), respectively. In addition, the two notched bands versus the characteristic impedance Z 3, Z 4 and electrical length θ 3, θ 4 are shown in Figs. 9-(e), the center frequency of the first notched band decreases as θ 3, Z 3 increase, and the center frequency of the second notched band also decrease as θ 4, Z 4 increase. In addition, the center frequency of each notched band can be adjusted independently when the other notched band is fixed. Moreover, the bandwidth of the notched bands can also be adjusted by changing the characteristic impedance Z 3, Z 4 and electrical length θ 3, θ 4 when Z 0, Z 1, Z 2, θ 1 andθ 2 are fixed. 104

10 Tạp chí Khoa học và Kỹ thuật - Học viện KTQS - Số 160 (4-2014) (c) (d) (e) Figure 9. S 21 under weak coupling, C 0 = 0.02pF, Z 1 = 55Ω, Z 2 = 30Ω, Z 3 = 43Ω, Z 4 = 90Ω, θ 2 = 30 0, θ 3 = 123 0, θ 4 = the first notched band versus θ 3 and Z 3, Z 1 = 55Ω, Z 2 = 60Ω, Z 4 = 90Ω, θ 2 = 30 0, θ 4 = (c) the first notched band versus θ 4 and Z 4, Z 1 = 55Ω, Z 2 = 60Ω, Z 3 = 43Ω, θ 2 = 30 0, θ 3 = (d) the second notched band versus θ 4 and Z 4, Z 1 = 55Ω, Z 2 = 60Ω, Z 3 = 43Ω, θ 2 = 30 0, θ 3 = (e) the second notched band versus θ 3 and Z 3, Z 1 = 55Ω, Z 2 = 60Ω, Z 4 = 90Ω, θ 2 = 30 0, θ 4 = (C 0 = 0.02pF, Z 0 = 50Ω, f 0 = 7.7GHz) Measured and Simulated Results of Proposed UWB Bandpass Filters with A/Two Notched Bands Based on the above theoretical analysis, the final parameters for the UWB filter circuit with a notched band of Fig. 6 are: Z 0 = 50Ω, Z 1 = 55Ω, Z 2 = 60Ω, Z 3 = 75Ω, f 0 = 105

11 Chuyên san Công nghệ thông tin và Truyền thông - Số 04 (4-2014) 7.7GHz, θ 1 = 90 0, θ 2 = 30 0, θ 3 = ; and the final parameters for the UWB filter circuit with dual notched bands of Fig. 8 are: Z 0 = 50Ω, Z 1 = 55Ω, Z 2 = 60Ω, Z 3 = 43Ω, Z 4 = 90Ω, f 0 = 7.7GHz, θ 1 = 90 0, θ 2 = 30 0, θ 3 = 125 0, θ 4 = The photographs, measured results and simulated results of the UWB filters with a/two notched bands are shown in Fig. 10. For the UWB filter with a notched band (Fig. 10), the measured 3-dB fractional bandwidth is approximately 126% ( GHz), a notched band located at 5.7 GHz with 10-dB bandwidth of 7.0% ( GHz). The group delay is 2.0 ns in 5.7 GHz. Furthermore, over 15-dB upper stopband is achieved from 12.6 GHz to 19.2 GHz (2.5f 0 ). The measured group delay is less than 0.22 ns from 2.1 GHz to 12 GHz. For the UWB filter with two notched bands (Fig. 10), the measured 3-dB fractional bandwidth is approximately 127% ( GHz), two notched bands located at 3.5 and 8.1 GHz with 10-dB bandwidth of 10% ( GHz) and 4.3% ( GHz). The group delay is 2.1 ns in 3.5 GHz and 1.6 ns in 8.1 GHz. The measured group delay is less than 0.25 ns from 2.2 GHz to 11.6 GHz. The slight frequency discrepancies between the measured and simulated results are mainly caused by the limited fabrication precision and measurement errors. In addition, Table 2 illustrates the comparisons of measured results for several ultra-wideband bandpass filter structures with single/dual notched bands. Compared with other UWB filters [25]-[32], the proposed UWB filters with a/two notched bands have very compact effective circuit sizes (0.35λ g0 0.19λ g0, 0.35λ g0 0.32λ g0 ) and simpler structures. And some transmission zeros can be realized to improve the upper stopband based on transversal signal-interaction concept. Figure 10. The photographs, measured and simulated results of the UWB bandpass filters with notched bands Single notched band dual notched bands. 4. CONCLUSION In this paper, compact UWB bandpass filters with a/two notched bands are proposed based on only a ring resonator with open/shorted stubs. Several transmission zeros can be realized to improve the upper stopband for the filter structures. Compared with former UWB structures, the proposed UWB filters have more compact effective circuit size, wider bandwidth, simpler 106

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14 Tạp chí Khoa học và Kỹ thuật - Học viện KTQS - Số 160 (4-2014) [38] Minh Tan. Doan, W. Q. Che and P. L. Nguyen, A novel wideband bandpass filter using open stubs multiple-mode ring resonator, In Proc. Int. Conf on Advanced Technologies for Communications., pp Oct [39] Minh Tan. Doan, Novel Ultra-wideband bandpass filter with notched band using stubs loaded multiple-mode ring resonator, In Proc. Int. Conf on Advanced Technologies for Communications., pp Oct Manuscript received ; accepted Tran Quang Nguyen graduated Faculty of Radio-Electronics Engineering, Le Qui Don Technical University, Hanoi, Vietnam in 2000, completed the master training course specialized the technique of electronic at Le Quy Don Technical University in He is currently a scientific and technological management assistant of Vietnamese Defence Ministry s General Technical Department, and a fellow of The Scientific and Technological Military Institute. MinhTan Doan received the B.Eng and M. Eng degrees in Radio Electronic Engineering from Le Quy Don Technical University in 2000 and 2003, respectively, Ph.D. degree in Department of Communication Engineering Nanjing University of Science and Technology, Nanjing, China in He is currently a lecturer of Faculty of Radio-Electronics Engineering, Le Qui Don Technical University, Hanoi, Vietnam. His research interests include the design of microwave filters and associated RF modules for microwave and signal processing for communications. Chi Hieu Ta was born in Vinh Phuc in He graduated the Military Technical Academy in 1994 with honor. He got his MSc degree in electronic engineering in the National Defense Academy of Japan in 2002 and his PhD degree in signal processing in the University of Strathclyde, United Kingdom in He is currently working at the Faculty of Radio Electronics, Military Technical Academy. His research interests include precoding and equalization for MIMO systems, microwave engineering and computational electromagnetics. 109