Devices selection or the construction o a microwave transmission link at 2.45 GHz E. ZIRINTSIS, C. PAVLATOS, C.A. CHRISTODOULOU 2, V. M. MLADENOV 3 IT Faculty, Hellenic American University, 2 Kaplanon Str., 06 80 Athens, GREECE e-mail: ezirintsis@hau.gr, cpavlatos@hau.gr 2 School o Electrical and Computer Engineering, National Technical University o Athens, 9 Iroon Politechniou Str., 57 80 Athens, GREECE e-mail: christ_th@yahoo.gr 3 Department o Theoretical Electrical Engineering, Technical University o Soia Soia 000, Kliment Ohridski blvd. 8; BULGARIA e-mail: valerim@tu-soia.bg Abstract: Today s communication systems allow voice and data to be transmitted using both digital and analogue transmission. Principles o analogue transmission are demonstrated using double sideband with carrier (DSBC) amplitude modulation (AM). A complete transmitting and receiving system is constructed, including a voltage controlled oscillator (VCO), a modulator, three ampliiers, two ilters and a detector, at microwave requency, and a audio and power ampliier or the low requency signal. Two antennas were also constructed. The carrier requency was selected to be 2.45 GHz and the transmitted inormation had a bandwidth o a ew khz (voice). Keywords: Analogue transmission; double sideband with carrier (DSBC); amplitude modulation (AM), voltage controlled oscillator VCO), microwave transmission. Introduction The great interest in microwave requencies basically arises rom the ever-increasing need or more radio-requency-spectrum space and the rather unique uses to which microwave requencies can be applied. Today, the majority o applications o microwaves are related to radars and communication systems. Radar systems are used or detecting and locating air, ground, or sea-going targets, by airport traiccontrol radars, missile tracking radars, irecontrol radars, and other weapons systems. Radar is also used or weather prediction and remote sensing applications. Microwave communications systems handle a large raction o international and other long-haul telephone traic, in addition to television programs and military communications. Although the requencies involved are high, still the basic principles o communications apply. Aim o this paper is to demonstrate the construction and the devices selection or a transmitting and receiving system. The constructed system includes: a voltage controlled oscillator (VCO), a modulator, RF ampliiers, a diode detector, an audio ampliier and an audio power ampliier. For the design o the band pass ilter used or the work presented here, the EEso s series IV Libra computer aided design package (CAD) was used in order to simulate any microwave structure [], and Protel Easytrax, a low-cost, powerul, sotware package or producing printed circuit board (PCB) artwork. ISSN: 790-57 287 ISBN: 978-960-474-39-7
2. Sotware packages 2. HP EEso s series IV Libra For the design o the band pass ilter used the EEso s series IV Libra computer aided design package (CAD) was used. It is a powerul package, being able to simulate with great accuracy any microwave structure []. The manuacturer s manual book appreciates that the design simulation results compared to the real structure can have a deviation not more than 5 % maximum. This accuracy is good enough to make this package being widely used. The Series IV project design environment includes the ollowing windows: Deault window, Schematic Window or system network block diagram description, LineCalc analysis and synthesis window, Test window or system network simulation and measurement, Design Layout capability, Momentum analysis, Graphic display windows or viewing simulation results. 2.2 Protel Easytrax Protel Easytrax is a low-cost, yet powerul, sotware package or producing printed circuit board (PCB) artwork. Easy to learn and easy to use, Easytrax puts proessional-quality PCB layout tools. It has many advanced eatured including component library support and comprehensive plotting acilities. This package was used or the construction o the micro strips incorporated in this work. 3. Transmitting system A communication system consists o a transmitter, a channel and a receiver. The purpose o the transmitter is to change the inormation so that transmission over a speciied channel is possible. Here the several parts o the transmitter are described and the entire transmitting system is designed. 3. Oscillators The oscillator is an electronic generator that operates rom a dc power source and can produce ac signal requencies ranging up to millimetre waves. A well - designed oscillator will have a uniorm output, varying in neither requency nor amplitude [2, 3]. A Voltage Controlled Oscillator (VCO) was chosen rom the Mini-Circuits catalogue instead. That was the JTOS-3000, which has wide requency range, linear tuning characteristics, excellent harmonic suppression and low phase noise. This model was based on surace mount technology. It had 4 pins and its actual dimensions were (a) (b) (c) = =20.04 3.3 6.35 (mm). According to manuacturer s dimension datasheets, the appropriate substrate had to be chosen or the circuit to be constructed. Protel design package was used to produce the layout o the micro strip which is depicted in Figure, in scale, according to the dimension datasheets. (pin2) (pin5) Figure : Oscillator s micro strip layout (in scale) Pin 2 was used or the power supply (V in ) and pin 5 or the tuning voltage (V tune ). The microwave output was taken at pin 3 (V out ). Care was taken to taper the ground plane at the output so that the signal would be not aected by the extra capacitance which would be created. All the other pins were connected to ground. According to the Mini-Circuits technical datasheets, the oscillator should be biased in a particular way in order to minimise noise. There were some active bias circuits or improved power line iltering but these should be adopted i the oscillator was going to be used in a highly external generated noise signals environment. That was not the case, thereore to improve the phase noise perormance o the VCO under external load conditions the ollowing steps were taken: (pin3) ISSN: 790-57 288 ISBN: 978-960-474-39-7
A low ESR capacitor o 0 µf was placed on the V cc and V tune lines Decoupling o the lines was enhanced urther by including a choke o 0 µh in series with them IN 4 ERA-SM C 3 R bias V cc RF Choke OUT That coniguration is depicted in Figure 2. C block 2 C block V cc 0µH V tune 0µH 0µF pin 2 pin 3 VCO Out Figure 3: The ERA-SM biasing circuit pin 5 0µF Figure 2: Decoupling o power supply and tuning voltage with LC type iltering 3.2 Modulator ampliier For the transmission o a signal through a channel, modiication should take place. By varying the ampliier s gain with the modulation signal applied to its biasing circuit, the carrier s amplitude would vary in phase. The ampliier used or the modulation was a monolithic Broadband DC to 8 GHz, Surace Mount ampliier, the ERA-SM; it was chosen rom Mini-Circuits catalogue. Its main eatures were miniature microwave ampliier, requency range rom DC to 8 GHz, usable up to 0 GHz and up to 8.5 dbm output power The discrete components such as chip capacitors, resistors and transistors are connected in place by soldering. According to the datasheets, the ampliier s biasing circuit was as shown in Figure 3. As or the oscillator the RT/duroid 5880 substrate was chosen. The ampliier had a 50 Ω input and output impedance. Thereore the micro strip lines had the same width as or the oscillator case, w = 2.434 mm. Again the PROTEL drawing package was used to draw the circuit. The layout is shown in Figure 4. Figure 4: Modulator s layout With the ampliier s input connected to the oscillator, its output was connected through a 20 db attenuator to the Spectrum Analyser. The biasing voltage was not the modulating signal since the reading would be more diicult to be taken. This would be due to the act that the output response would vary with respect to the modulation signal and luctuation would exist. Instead, a steady voltage was applied making sure that the voltage at pin 3 would not exceed 4. Volts and the current would be no more than 75 ma maximum. The latter was achieved by choosing the correct value or R bias. The graph o Figure 5 was printed when the ampliier had the maximum possible gain. A simulation was run at the Test bench window and the graph o S 2 was obtained (Figure 6b). It can be seen that indeed the maximum possible gain at 2.45 GHz was 4.6 db. That was just 0.0 db higher than the readings taken. That showed that the gain could not be increased up to the maximum point (according to the manuacturer,.8 db). ISSN: 790-57 289 ISBN: 978-960-474-39-7
Figure 5: Ampliier s gain (a) adjacent resonators are parallel to each other, as mentioned earlier. This parallel arrangement is convenient or micro strip ilters with bandwidths o up to about 5 %. Micro strip coupling theory shows that maximum coupling between two parallel micro strips is achieved when the length o the coupled regions is λ g /4 (g, stands or guide) or some odd multiple thereo, where λ g is the wavelength o the electromagnetic waves in the micro strip. Resonator theory shows that or resonance to occur, the resonators must be λ g /2 or any multiple. For the implementation o the present work, a micro strip parallel-coupled band pass ilter was designed. The design process was carried out with the use o the HP EEso s-libra Series IV computer aided design (CAD) package. The design was simulated using the CAD system and its requency response determined. Comparisons were then made with the design speciications to veriy the validity o the design method. The design method was based largely on one given by [3] and used a commonly available substrate with a dielectric constant (ε r ) o 0.5 and a thickness (h) o.575 mm. The ollowing speciications were set, which can also be seen in the Figure 3 below (not in scale): Substrate RT/Duroid 600.5 (ε r ) = 0.5, thickness (h) =.575 mm Micro strip parallel-coupled band pass ilter Chebyshev requency response characteristic Pass band ripple not greater than 0.0 db Centre requency o 2.45 GHz Pass band ripple bandwidth o 4 % (i.e. rom 2.40-2.499 GHz) Insertion loss greater than 35 db at 6 % o the centre requency (i.e. 2.3 and 2.6 GHz) The centre requency was chosen arbitrarily at 2.45 GHz, since that requency was included in the licence ree range o spectrum. (b) Figure 6: a) Schematic o ampliier s circuit (CAD package) and b) its response S 2 A band pass ilter can be constructed [4] rom a series o micro strip resonators positioned so that 3.3 Transmitting and receiving antenna Two antennas were constructed, or the transmitting and the receiving end o the system. ISSN: 790-57 290 ISBN: 978-960-474-39-7
There were three options: a) the Horn, b) the Dipole and c) the Monopole antenna. Both (a) and (b) were more diicult to construct thereore the monopole antenna was chosen. A monopole antenna is shown in Figure 8. 35 the ground is not a perect conductor, the vertical quarter-wave antenna will lose a considerable amount o power in the resistance o the grounding system. The radiation pattern or a quarter-wave vertical antenna is shown in Figure 9. The antenna radiates equally well in all directions in the horizontal plane; that is, it is omni directional. In the vertical plane, the radiation is directed low toward the horizon, thus providing excellent long distance propagation characteristics. LA (db) L Aa Omnidirectional Radiation Pattern L Ar 0.0 2.3 2.40 2.45 2.499 2.6 Antenna (GHz) Ground Ground Figure 7: Filter speciications Antenna Mirror Image Direct wave Relected Waves Figure 8: Perectly grounded quarter-wave vertical antenna with a quarter-wave mirror image This antenna is a quarter-wavelength long [2], laying vertically above a perect ground. It has the same characteristics as a hal-wave vertical dipole, because a perect ground will produce a mirror image o the quarter-wave, a result o the relected radio waves as shown in the igure. I Figure 9: Omni directional radiation pattern or a quarter-wave vertical antenna When used with a perect ground, the input impedance o the quarter-wave antenna is about 36 Ω, which makes an acceptable impedance match or a 50 Ω coaxial transmission line. Also, because one side o the vertical antenna is at ground level, the antenna is unbalanced, so an unbalanced coaxial cable is a perect eeder. The centre conductor o the coaxial cable is connected to the quarter-wave antenna, which is insulated rom ground, and the shield is connected to ground. I maximum power transer is to be delivered by a transmitter to its antenna, the antenna must be resonant to the requency o the radio wave energy transerred rom the transmitter. When the antenna is in resonance, the antenna current is maximum. The voltage and current displacement curves are shown in Figure 0. It must mentioned that both the voltage node and the current loop occur at ground level, which is eectively the centre o the antenna. ISSN: 790-57 29 ISBN: 978-960-474-39-7
Ground Antenna Mirror Image Figure 0: Current and voltage displacement curves or a quarter-wave antenna I V resistor R l between positive peaks o the carrier wave, but not so long that the capacitor voltage would not discharge at the maximum rate o change o the modulating wave, that is c << RlC<< (2) W where: W is the message bandwidth. An example o the carrier and its demodulated signal is shown in Figure 2, or a carrier requency o 20 khz and a message o 00 Hz. 4. Receiving system 4. Design o the detector According to reerence [5], the detector s design was implemented as ollows. Two assumptions were made: It was assumed that the diode was ideal, presenting resistance r to current low in the orward-biased region and ininite resistance in the reverse-biased region. It was urther assumed that the AM wave applied to the envelope detector was supplied by a voltage source o internal impedance R s. (a) R s AM wave Diode C R l V o(t) Figure : Envelope detector The charging time constant (r +R s ) C should be short compared with the carrier period / c that is: ( r + Rs) C<< () So that the capacitor C would charge rapidly and thereby ollow the applied voltage up to the positive peak when the diode is conducting. On the other hand, the discharging time constant R l C should be long enough to ensure that the capacitor discharges slowly through the load c (b) Figure 2: a) AM wave input b) Envelope detector output By substituting the speciications to equations () and (2): c << RlC << W ISSN: 790-57 292 ISBN: 978-960-474-39-7
and 408.6 0 2 << C R l ( r + R ) C << s c << 0 ( 7+ 50) << 9 C 2.45 0 4 where: rom datasheets r = 7 Ω and R s = 50 Ω. These equations were satisied or C =0 pf and R l = kω. As soon as the component values were calculated, the circuit o the detector was constructed. The device was surace mount. Again, the same substrate as or the ampliier was used. The circuit layout is shown in Figure 3 below: V in C by C by Gnd V cc L R L R 2 Diode C 50Ω line R l V out Gnd inductors were the same as the ones used or the ampliier. Resistors R and R 2 were used in order to bias the diode. R was set to be variable (0 kω) and R 2 = 000 Ω since a orward voltage o about 0.7 V was needed. With respect to Figure 6.5b it can be seen that the demodulated signal has a residue o the carrier requency which is appearing as noise imposed on the signal. That was impossible to be seen with the oscilloscope, however it existed. To eliminate this eect low pass iltering was applied. An active ilter was chosen so that ampliication would take place. 4.2 Choice o diode Since the requency o the carrier was at 2.45 GHz, a ast switching diode was needed. Also the lower the orward resistance the better the perormance would be. These characteristics were satisied by the Schottky Barrier diode. From MACOM catalogue the SOT-23(287) o MA4E2054 series diode was chosen. 4.3 Filter design and construction The ilter employed was a our pole ilter since a sharp cut-o response was required [6]. The speciications o the ilter were set arbitrarily: Butterworth response have a gain o 20 db cut-o requency at 0 khz attenuation o 80 db at 00 khz Figure 3: Envelope detector layout (not in scale) In the constructed circuits, as ar as the coupling was concerned, it was paid extra care to minimise it. The practical rule to ollow was that the micro strip line spacing had to be o the order o three substrate thicknesses or more. The circuits were attached to an aluminium base so that they would be easier to handle and a better grounding would be provided. The screws used to attach the micro strip board on the bottom o the housing had to be plastic, except o some circuits which needed grounding on the upper side like the oscillator and the diode detector. The bypass capacitors were 00 pf and the 00dB LA (db) 3dB L Ar ω c ω s L Aa Figure 4: Filter speciications ω (rad) ISSN: 790-57 293 ISBN: 978-960-474-39-7
4.4 Testing o the ampliier The ampliier employed or the implementation o the ilter was the LM 74 (o National Semiconductors). It is a cheap and robust IC. That is why it is widely used. The circuit components were soldered on a strip-board and testing took place. A 0.5 V amplitude sinusoidal voltage was applied at the input and the output waveorm was recorded. The whole requency range up to 00 khz was tested. The results were recorded and the ollowing graph was plotted: The results were satisactory. At the stop band the signal had an attenuation o 75 db. However the cut-o requency was not at 0 khz. The gain was almost lat up to 6.5 khz and then gradually it started to attenuate. At the requency o 0 khz it was equal to nine. By varying resistor R 8 the gain was altered and the maximum possible was K = 27. Above that limit the signal was distorted. signal used. I higher requencies were to be used the LM833 would be a better choice since its slew rate was 7 V/µs. Thereore the op-amp was not substituted since it had the advantage o an input impedance o 2 MΩ. The next stage involved the power ampliication. No urther (voltage) ampliication was needed since the power ampliier had an inherent gain o 50. 4.5 Power ampliier There was a large amount o audio power ampliiers available in the market. The LM380N 4-pin dual in line package was chosen, which required very ew external components to make a complete 2.5W power ampliier with a load o 8Ω. The ollowing coniguration was implemented. input Figure 6: Circuit coniguration Figure 5: Measured response o Low pass Filter In active devices the higher the gain, the larger the signal distortion. That was the reason the gain was kept low. The LM74 had a good perormance. Nevertheless, there were other ICs that had lower noise characteristics. The LM833 low noise dual op-amp was bought or trial purpose, to compare the noise perormances. Although the LM833 had 4.5 nv/hz /2 voltage noise, no actual dierence was observed. The slew rate o LM74 was 0.5 V/µs which was satisactory or the low requency modulating The component values are R V =2.2 Ω, R V2 = 2.2 MΩ, C = 3300 pf, C 2 = 470 µf (25 V). The components C and R V were inserted or tone control. These could be omitted i that option was not required. The IC exhibited a constant gain o 34 db (=50) which was checked by applying a voltage at its input. The loudspeaker used was chosen rom Farnell catalogue. Its was a cheap miniature loudspeaker o 0.3 W power. 5. Conclusions In this paper a complete transmitting and receiving system was constructed. The constructed circuit included a Voltage Controlled ISSN: 790-57 294 ISBN: 978-960-474-39-7
Oscillator (VCO), a modulator, three ampliiers, two ilters and a detector, at microwave requency, and an audio and power ampliier or the low requency signal. Two antennas were also constructed. The carrier requency was selected to be 2.45 GHz and the transmitted inormation had a bandwidth o a ew khz (voice). In order to achieve the required centre requency, the resonators were shortened by trimming conductor material orm the ends i.e. increasing the resonance requency. Reerences [] Libra, manual (Help option o the package). [2] F.R. Dungan, Electronic Communications Systems, Delmar, 997. [3] E.L. Holzman, R.S. Robertson, Solid-State Microwave Power Oscillator Design, Artech House, INC, 992. [4] G.L. Matthaei, L. Young, E.M.T. Jones, Microwave Filters, Impedance-Matching Networks and Coupling Structures, McGraw Hill, 964. [5] S. Haykin, Communication Systems, John Willey & Sons, 2 nd Edition, 983. [6] J.G. Wilson, The design and operation o a microwave link monitoring system, Measurement, Vol. 35, No., 2004, pp. 74-78. ISSN: 790-57 295 ISBN: 978-960-474-39-7