High-Power Dual-Interleaved ZVS Boost Converter with Interphase Transformer for Electric Vehicles

High-Power Dual-Interleaved ZVS Boost Converter with Interphase Transformer for Electric Vehicles G. Calderon-Lopez and A. J. Forsyth School of Electrical and Electronic Engineering The University of Manchester PO Box 88, Sackville Street Building Manchester, M60 1QD UK Abstract- A dual-interleaved boost converter is presented with zero-voltage switching for applications in electric vehicle power trains. The topology has one input inductor and an interphase transformer (IPT) between the two channels. By using an appropriate differential inductance value for the IPT, the converter achieves transitions with zero-voltage switching in the devices. The circuit operation is described followed by design details for a 40 kw prototype. The design is validated by experimental testing. I. INTRODUCTION Interfacing the energy sources and storage devices such as fuel cells, batteries and supercapacitors with the traction system on an electric vehicle requires several DC-to-DC conversion functions. Typically, fuel cell voltages are quite low and badly regulated, supercapacitor voltages must vary over a 2:1 range, whilst traction drives require a higher DC voltage, up to 600V. Multiphase interleaved boost converters represent an excellent option in the mid to high-power ranges for voltage conversion ratios in the range 1:4, and this is confirmed by the growing research interest in this topology for electric vehicles [1-5]. The switching frequency ripple cancellation at input and output and the lower rating of the individual devices are important advantages. The use of soft-switching techniques to reduce the commutation losses and increase the switching frequency is an established technique to gain size reductions without compromising a converter's efficiency, although the control requirements, component count and complexity of the overall system are usually increased. This paper deals with the operating principle of a simple, bi-directional, dual-interleaved boost converter with zero-voltage switching (ZVS) using an interphase transformer (IPT) and one input inductor for a 40 kw electric vehicle application. Compared to other soft-switched, interleaved boost converters reported in the literature [6-10], the total number of components is reduced because this topology does not employ auxiliary switches, the ZVS operation is achieved over a large range of duty-ratios and load conditions, and the switching frequency is constant. The circuit has some similarities with the solution in [11] where a three-phase topology was adopted and zero-voltage switching was achieved by designing each input inductor to carry a large ripple current. The solution proposed in this paper results in two magnetic components, one an inductor with predominantly DC magnetisation and the second an IPT which has purely AC magnetisation. A high flux core may then be used for the inductor whilst a ferrite material with low core losses may be used for the IPT, giving greater flexibility in the design optimisation. In this paper a distributed gap ferrite core is described for the IPT to minimise the AC copper losses. Furthermore the circuit inherently offers the possibility of bi-directional power flow. II. DESCRIPTION AND OPERATING PRINCIPLE OF THE CIRCUIT The diagram of the converter is presented in Fig. 1. The circuit comprises four bi-directional switches, each with a snubber capacitor in parallel to achieve zero voltage transitions during the turn-on and turn-off instants [12]. The snubber capacitors may consist partly or entirely of the device output capacitances. To understand the generation of the zero voltage switching, the steady-state waveforms are shown in Fig. 2, with two different values of IPT magnetising inductance, L diff. The waveforms correspond to leg A, and assume that transistors Q 1 and Q 2 operate with a 180 -phase shift and equal duty ratios, D, where D< 0.5. Also, the upper devices in each leg operate in anti-phase with the respective lower device, and a small dead time, t d, is included in each switching leg. Fig. 1. ZVS dual-interleaved boost converter with IPT and input inductor. 978-1-422-2812-0/09/$25.00 2009 IEEE 1078

Fig. 2 depicts the gate driving signals v g1 (t) and v g3 (t), the input inductor current, i in (t), the current in the first IPT winding, i L11 (t), and the collector-emitter voltage of Q 1, v ce1 (t). The input inductor current is continuous and divides equally between the two windings of the interphase transformer. In Fig. 2.(a), the magnetising inductance of the IPT is assumed to be very large, such that the differential current i diff (t) flowing between points A and B is small. Normal hard switching waveforms result. However in Fig. 2.(b), the magnetising inductance of the IPT is much smaller such that a substantial differential mode current flows between nodes A and B. Since the current in the first winding i L11 is equal to i in (t)/2 i diff (t), then for a sufficiently large differential current, i L11 (t) reverses transiently, as shown in Fig. 2.(b). The conducting devices in leg A are marked on the i L11 waveform. Each device in the leg is carrying a forward current at the turn off instant, which is commutated to the snubber capacitors, resulting in a controlled voltage transition before the current transfers to the incoming diode. Providing that the incoming transistor is signalled to turn on whilst the anti-parallel diode is in conduction, virtually lossless zero voltage switching is achieved. For duty-ratios equal to 0.5, or greater than 0.5, similar operation results. The same current waveforms are generated in leg B of the converter, but displaced in time by half a period, resulting in the cancellation of ripple currents at the input and output of the converter, and ensuring small overall input ripple current levels. The presence of devices Q 3 and Q 4 means that bidirectional power flow may be achieved, as for example would be required for a supercapacitor interface or battery charge/discharge regulator. III. BOUNDARY OF SOFT-SWITCHING OPERATION A straightforward way to identify approximately the boundary of the soft-switching operating mode of the converter is to consider the condition for achieving negative current in the IPT windings. The current i L11 in the first half of the IPT is sketched in Fig. 3 at the boundary of soft-switching operation for the conditions of D< 0.5 and D> 0.5. The current just falls to zero at the start and end of the cycle when Q 1 turns on. In practice, to achieve zero-voltage switching, a slightly negative current would be required at these instants, causing Q 3 to conduct and providing a current to charge/discharge the snubber capacitors. The current i L11 in the IPT is the sum of half the converter input current and the IPT differential current. The peak-topeak ripple current ΔI T in i L11 may therefore be written as ΔI T = ΔI in /2 + ΔI diff, (1) where ΔI in is the peak-to-peak ripple current in the input inductor and ΔI diff is the peak-to-peak differential ripple current. By considering the voltage waveform across the input inductor, and using the voltage conversion ratio expression v o /v in = 1/(1 D), the expressions for the peak-to-peak ripple currents may be shown to be: Fig. 2. Comparison of switching waveforms of leg A for D< 0.5: (a) Assuming a small i diff (t); (b) Assuming a large i diff (t) for ZVS operation. Δ I in = vindt 1 2D 2L 1 D in for 0 < D 0.5 vindt 2D 1 for 0.5 < D < 1 2L D in (2) 978-1-422-2812-0/09/$25.00 2009 IEEE 1079

L r decreasing Fig. 4. Boundaries for soft-switching operation in the converter with L r 4. Fig. 3. Ideal steady-state waveforms of i L11 at the ZVS boundary: (a) D< 0.5, (b) D> 0.5. vdt o for 0< D 0.5 Ldiff Δ I (3) diff = vo ( 1 D) T for 0.5< D <1 Ldiff where L in is the input inductor, L diff is the differential inductance of the IPT, and T is the switching period. From Fig. 3 the condition for soft-switching to occur may be expressed as: Δ IT I > L (4) 2 2 where I L is the average input inductor current. Substituting for ΔI T in (4) using (1), and eliminating ΔI in and ΔI diff using (2) and (3) the resultant condition for softswitching to occur is: 1 2 ( 1 D) D( D+ ) for D 0.5 2 Lr 2L in k = < (5) RT 2 1 2 ( 1 D) ( D 2 + ) for D > 0.5 Lr where L r is the inductance ratio, L r = L diff /L in, and R is the load resistance. Equation (5) was used to plot the soft-switching boundary for the converter in terms of the parameter k for a range of values of L r as shown in Fig. 4 and Fig. 5. Soft-switching will occur for operating points underneath the curves. The results indicate that soft-switching may be achieved over a greater range of duty-ratio values and with greater values of k when a smaller value is used for the inductance ratio L r. Also, with extremely small or extremely large values of duty ratio the converter will lose the ability to achieve soft-switching. L r increasing Fig. 5. Boundaries for soft-switching operation in the converter with L r 4. IV. PROTOTYPE DESIGN To demonstrate the operation of the converter a prototype was designed to supply a 40 kw, 500 V load, the operating frequency was chosen to be 30 khz. By selecting an inductance ratio L in /L diff = 0.7 and making L in = 50 μh and L diff = 35 μh, the resultant value of k= (2L)/(RT)= 0.48. From Fig. 4 with L r = 0.7 and k= 0.48 the soft-switching duty-ratio range is 0.19 D 0.6, corresponding to an input voltage range of 200 V v in 405 V for a fixed 500 V output. Semikron IGBT modules SKM400GB125D were used for the switching devices and the input inductor was wound on Metglas C-cores, the maximum inductor current being 167A. The differential current in the IPT has a maximum peak-topeak value of 238 A when the duty ratio is 0.5. A ferrite core was chosen for the IPT to limit the core losses and a distributed gap structure was used to minimise the eddy current losses in the windings. The construction of the IPT core is illustrated in Fig. 6 where it is seen that the two winding limbs of the IPT were formed using a stack of ferrite ring cores with small spacers between the cores to create the 978-1-422-2812-0/09/$25.00 2009 IEEE 1080

Fig. 8. Boundary of the ZVS operation. i Q1, 20 A/div, i L22, 20 A/div, 5 μs/div. v in = 403 V, v o = 500 V, P in = 39.8 kw, D= 0.21, f sw = 31 khz. Fig. 6. IPT with distributed gap core. distributed gap. The top and bottom of the core were formed using ferrite 'I' pieces. Each winding limb consisted of 15, 34 mm diameter ring cores. The spacers between the cores were approximately 0.4 mm thick. The IPT was wound using 0.4mm diameter litz wire. The losses in the IPT with the converter operating at 40 kw with a 0.5 duty-ratio were estimated to be 200 W, consisting of 80 W core loss and 120 W copper loss. The current sharing between the windings of the IPT was ensured by a peak current mode controller, as shown in Fig. 7. Individual clocks with a 180 phase shift initiate the conduction of the transistors. Two current transducers are used to sense the current levels in the windings of the IPT, and when these signals exceed the reference value, the transistors are turned off. To avoid sub-harmonic instabilities with D> 0.5, a slope compensating ramp per channel is used. The controller was implemented using the interleaved PWM integrated circuit UCC28220 from Texas Instruments. V. EXPERIMENTAL RESULTS Fig. 8 shows the waveforms of one of the IPT currents, i L22, and one of the IGBT currents, i Q1, for the converter operating at 40 kw at the minimum duty ratio for zero-voltage switching. The duty ratio was 0.21 and the input voltage was 403 V. The predicted value of the boundary duty-ratio in Fig. 4 is 0.19 and is slightly lower than the measured value due to circuit losses. The IPT current in Fig. 8 is unidirectional, but just falls to zero once per cycle. There is no anti-parallel diode conduction immediately before the Q 1 IGBT conducts. Fig. 9 shows the voltage and current waveforms for IGBT Q 1 with the converter operating at 40 kw. The input voltage was 300V and the duty-ratio was 0.4. The short period of antiparallel diode conduction immediately before the IGBT conducts confirms the zero voltage turn on waveforms. Expanded portions of the waveforms are shown in Fig. 10 and Fig. 11 to illustrate the turn on and turn off transients. A controlled discharge of v ce1 is seen in Fig. 10 immediately before the anti-parallel diode starts to conduct. Fig. 11 shows v ce1 rising rapidly at the turn off instant, however the only snubber capacitors present were the inherent output capacitance of the IGBTs. The addition of a small additional snubber capacitor would allow the rate of rise of v ce1 to be restricted and thereby limit the turn off losses. To illustrate the current sharing between the two IPT channels, Fig. 12 shows the IPT currents, i L11 and i L22, along with the input inductor current i in for the converter operating at 16 kw. A slight imbalance is evident between the two channels and this was attributed to small asymmetries in the Fig. 7. Block diagram of the peak current mode controller. Fig. 9. Steady-state waveforms with ZVS operation; Ch2: i Q1, 25 A/div, Ch4: v ce1, 100 V/div, 5 μs/div. v in = 300 V, v o = 500 V, P in = 40.8kW, P o = 39.6 kw, D= 0.40, f sw = 31 khz. 978-1-422-2812-0/09/$25.00 2009 IEEE 1081

additional snubber capacitors across the IGBTs would allow the device losses to be reduced by 10-15%. VI. CONCLUSIONS Fig. 10. Detail of the turn-on transient of Q 1. Ch2: i Q1, 10 A/div, Ch4: v ce1, 50 V/div, 500 ns/div. v in = 300 V, v o = 500 V, P in = 40.8 kw, P o = 39.6 kw, D= 0.40, f sw = 31 khz. This paper has presented a high power, dual-interleaved boost converter with interphase transformer for soft-switching operation. The topology employs the magnetising inductance of the IPT and the device output capacitances to achieve commutations with zero voltage. The boundaries for the ZVS operation were shown, and the details for building the interphase transformer with a distributed air gap were given. The PWM control of the converter and current balancing between the channels was achieved by a peak current mode controller, and the experimental results prove the operation of a prototype for 500 V output, 40 kw at 31 khz, achieving high efficiencies. ACKNOWLEDGMENT G. Calderon-Lopez thanks to the Mexican National Council of Science and Technology (CONACYT) for sponsorship of his PhD (grant 187133), the National Polytechnic Institute of Mexico (IPN) and the Public Education Ministry (SEP). The authors are also grateful to the UK Technology Strategy Board for funding this research. REFERENCES Fig. 11. Detail of the turn-off instant of Q 1. Ch2: i Q1, 20 A/div, Ch4: v ce1, 50 V/div, 200 ns/div. v in = 300 V, v o = 500 V, P in = 40.8 kw, P o = 39.6 kw, D= 0.40, f sw = 31 khz. circuit. The input current waveform confirms that the input current ripple frequency is twice the switching frequency, 62 khz. The efficiency of the converter when operating at 40 kw was measured to be in the region of 97%. The approximate distribution of the losses was 800 W in the IGBTs and diodes, 200 W in the IPT and 50 W in the input inductor. The use of Fig. 12. Current sharing in the windings of the IPT. Ch2: i L11, Ch3: i in, and Ch4: i L22, all at 20 A/div, 10 μs/div. v in = 300 V, v o = 419 V, P in = 16.6 kw, P o = 16.2 kw, D= 0.28, f sw = 31 khz. [1] A. Fratta, P. Guglielmi, F. Villata, and A. A.-V. Vagati, A., "Efficiency and cost-effectiveness of ac drives for electric vehicles improved by a novel, boost dc-dc conversion structure," Power Electronics in Transportation, 1998, pp. 11-19, 1998. [2] S. Chandrasekaran and L. U. Gokdere, "Integrated magnetics for interleaved dc-dc boost converter for fuel cell powered vehicles," IEEE Power Electronics Specialists Conference, pp. 356-361, 2004. [3] M. Gerber, J. A. Ferreira, I. W. Hofsajer, and N. Seliger, "Optimal interleaving of dc/dc converters in automotive applications," 10th International Power Electronics and Motion Control Conference, pp. 1-10, 2003. [4] H. Xu, X. Wen, E. Qiao, X. Guo, and L. Kong, "High power interleaved boost converter in fuel cell hybrid electric vehicle," IEEE International Electric Machines and Drives Conference, pp. 1814-1819, 2005. [5] G. Calderon-Lopez, A. J. Forsyth, and D. R. Nuttall, "Design and performance evaluation of a 10-kW interleaved boost converter for a fuel cell electric vehicle," 5th International Power Electronics and Motion Control Conference, vol. 2, pp. 1328-1332, 2006. [6] N.-J. Park and D.-S. Hyun, "N interleaved boost converter with a novel ZVT cell using a single resonant inductor for high power applications," IEEE Industry Applications Conference, vol. 5, pp. 2157-2161, 2006. [7] E. Sanchis-Kilders, A. Ferreres, E. Maset, J. D. Ejea, V. Esteve, J. Jordan, R. Garcia, and A. Garrigos, "High power passive soft switched interleaved boost converters," IEEE Power Electronics Specialists Conference, pp. 426-432, 2004. 978-1-422-2812-0/09/$25.00 2009 IEEE 1082

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