47 CHAPTER 3 DC-DC CONVERTER TOPOLOGIES 3.1 INTRODUCTION In recent decades, much research efforts are directed towards finding an isolated DC-DC converter with high volumetric power density, low electro magnetic interference and low cost for effectively processing the energy. In order to process the energy efficiently, different types of DC-DC converters are proposed. This chapter consists of different converter topologies with its advantages and limitations. 3.2 HARD SWITCHING CONVERTERS Hard switching converters obey the conventional switching phenomenon. While the switch is turned ON, the voltage across the switch decrease and the current through the switch tend to increases. This results in switching losses. Similarly, when the switch is turned OFF, the current through the switch tends to decrease and the voltage across its terminals increases. This too results in switching losses. Hard switching PWM converters such as two switch forward converters received a lot of interest and appreciations because of its robustness. For half bridge and full bridge converters, the primary switches are connected in a totem pole structure. Whenever the two switches are turned ON at the same time due to electromagnetic noise or radiation, it will be a destructive failure.
48 This problem is solved in two switch forward converter which is very critical for aerospace power supplies exposed to high energy radiation. The schematic of two switch forward converter is shown in Figure 3.1. Figure 3.1 Two switch forward converter The major disadvantages of two switch forward converters are hard switching and the need of large filter inductor. Hard switching leads to high switching losses for high frequency operation. In addition, the voltage on the output inductor is much higher in two switch forward converters than in half bridge converters. Two switch forward converters are not desirable for meeting higher efficiency and high power density requirements because of these penalties. 3.3 SOFT SWITCHING CONVERTERS Conventional PWM converters operate on hard switching phenomenon where voltage and current pulses, during their transition from high to low values or low to high values, interact with each other and cause power losses called switching losses and generate a substantial amount of electromagnetic interference.
49 Switching losses arise because of output capacitance of transistor, capacitance of diode and the diode reverse recovery. It is observed that switching losses are proportional to switching frequency. So, higher switching losses lead to the limitation in switching frequency. Because of the presence of wide spectral range of harmonics in PWM waveform, a high Electro Magnetic Interference (EMI) occurs. EMI also results from high current spikes caused by diode recovery. Switching losses and EMI can be reduced by using soft switching techniques at the expense of stress on the device. If the semiconductor device is made to turn OFF or turn ON when current or voltage is zero, then the product of voltage and current during transition is zero which leads to zero power loss. Thus switching losses are eliminated and the device can be made to operate at high switching frequencies. Size and weight of the device also reduces because of non-requirement of heat sink. Soft switching techniques can also be a valuable option to enhance the converter s efficiency. The soft switching techniques are widely categorized into two types namely Zero Voltage Switching (ZVS) and Zero Current Switching (ZCS). The technique in which the MOSFET turns ON at zero voltage is called Zero Voltage Switching (ZVS). During turn ON, the voltage is made as zero across the switch and current begins to rise after the voltage becomes zero. This is called ZVS. Conventional switching and Zero Voltage Switching are shown in Figure 3.2 and 3.3 respectively. The technique in which the semiconductor switch turns OFF at zero current is called Zero Current Switching (ZCS). Initially the device is conducting. So the current passing through the device is not zero and the voltage across the device is zero. In the ZCS condition, current is made as
50 zero. Thus there is no power loss during turn OFF of the device. Two soft switching techniques are inherently available for half bridge converters without any additional components. Figure 3.2 Characteristics of conventional switching Figure 3.3 Characteristics of zero voltage switching 3.4 PHASE SHIFT FULL BRIDGE PWM CONVERTER The phase shift full bridge PWM converter is widely used for DC-DC conversion. The topology is shown in Figure 3.4 and bulky capacitors are used to provide energy during the holdup time.
51 Figure 3.4 Phase shift full bridge PWM converter The bulky holdup time of the capacitor becomes the bottleneck to increase power density because the size of the capacitor is determined by the energy required during this time and the conducting current. Enlarging the operation range of the DC-DC stage could reduce the holdup time of the capacitors. In this way more energy stored in the capacitor could be utilized. However, conventional PWM converters have to sacrifice normal operation efficiency to extend their operation range. It is difficult to design a PWM converter for wide range of current with high efficiency. The smaller transformer turns ratio leads to high primary side current. So the conduction and switching losses increase. Consequently, efficiency suffers at normal operation conditions. For phase shift full bridge converters, the major problems are the high circulating current during normal operation, hard switching on the secondary side and efficiency at light loads. During each switching cycle there are freewheeling periods and high current circulates in the converter. To satisfy the holdup requirement, the duty cycle is selected relatively small. Thus, it leads to relatively high circulating current and a large amount
52 of conduction loss. More conduction loss reduces the efficiency. On the other hand, although the soft switching is achieved at the primary side, hard switching problems still remain for the secondary side devices. Switching loss and voltage stress of secondary side devices are severe issues. At light load conditions ZVS may be lost. Thus the efficiency under light load is another concern. 3.5 DIRECT COUPLING CONVERTER The well known Pulse Width Modulation (PWM) technique is widely used by which the width of a fixed amplitude pulse is modulated to obtain converter regulation. The other popular modulation technique is frequency modulation by which the frequency of a pulse is modulated to obtain converter regulation. In direct coupling converter the amplitude of a pulse train is modulated in order to obtain regulation. Such amplitude control is obtained by controlling the duty cycle of switches. The advantage of modulating the amplitude is that the output is directly coupled and controlled without the need of an output filter. This enables fast response, lower component count, lower cost and size. A significant inherent advantage of this converter is its soft switching nature. It enables zero voltage switching of the switching devices and greatly reduces switching losses. Efficiency over 90% is achieved. This results in compact converter design without the need of heat sinks. Figure 3.5 shows the direct coupling converter. It is assumed that there is an antiparallel body diode embedded in each of the two MOSFETs S 1 and S 2 which enable reverse current conduction through the switch. At the moment, when the current flowing through inductor L 1 is positive, switch S 1 which is under OFF state will now turn ON the antiparallel diode of S 2 in order to sustain the current through L 1.
53 Figure 3.5 Direct coupling converter This will reverse the voltage across L 1, since the capacitor C 1 has a value high enough, so that the voltage change across it is concerned as insignificant. The output diode D 1 will be reverse biased and the inductor current will decrease and tend towards a negative value. However, as long as the antiparallel diode of S 2 is turned ON, there is a zero voltage condition for S 2 which is then turned ON and zero voltage switching is obtained. When S 2 is turned OFF and current flow through the inductor L 1 is negative, S 1 can be programmed to turn ON. When current is directed through its antiparallel diode, ZVS is obtained as same as that of S 2. When the load current is increased, the average DC current level in the inductor L 1 will increase towards the negative side. Hence, there is no chance to form a loop current flowing through the body diode of S 2 and in this case only S 1 can have ZVS. Unlike conventional converters, this direct coupling converter modulates the output waveform amplitude. Inherent zero voltage switching accompanies this converter and makes this converter highly efficient. Thus compact size can be achieved. The operating load range of this converter is exceptionally high from full load to no load.
54 3.6 RESONANT CONVERTERS Resonant converter can achieve very low switching loss and enables resonant topologies to operate at high switching frequency. In resonant topologies, Series resonant converter, Parallel resonant converter, Series parallel resonant converter and LCC resonant converter are the most popular topologies. The characteristics and limitations of these topologies are presented in the following section. 3.6.1 Series Resonant Converter The circuit diagram of a half bridge series resonant converter is shown in Figure 3.6. The resonant inductor L r and resonant capacitor C r are connected in series. They form a series resonant tank. The resonant tank will then be in series with the load. From this configuration, the resonant tank and the load act as a voltage divider. By changing the frequency of input voltage, the impedance of resonant tank will change. Figure 3.6 Half bridge series resonant converter This impedance will divide the input voltage with load. Since it is a voltage divider, the DC gain of series resonant converter is always less than one. At resonant frequency, the impedance of series resonant tank will be very small and all the input voltage will drop on the load. So, for a series resonant
55 converter, the maximum gain happens at resonant frequency. Operating region is on the right side of resonant frequency f r. When switching frequency is lesser than the resonant frequency, the converter will work under zero current switching condition. In fact, the rule is that when the DC gain slope is negative, the converter is working under zero voltage switching condition and when the DC gain slope is positive, the converter will work under zero current switching condition. For power MOSFET, zero voltage switching is preferred. At light load, the switching frequency needs to be increased to keep the regulated output voltage. This is a big problem for series resonant converter. To regulate the output voltage at light load, some other control method has to be added. With the above analysis, it can be concluded that the series resonant converter has the major problems of light load regulation, high circulating energy and turn OFF current at high input voltage condition. 3.6.2 Parallel Resonant Converter The schematic of parallel resonant converter is shown in Figure 3.7. For parallel resonant converter, the resonant tank is still in series but the load is in parallel with the resonant capacitor. Figure 3.7 Half bridge parallel resonant converter
56 More accurately, this converter should be called as series resonant converter with parallel load. Since transformer primary side is a capacitor, an inductor is added on the secondary side to match the impedance. At light load, there is no need to change the frequency to keep the regulated output voltage. So light load regulation problem does not exist in parallel resonant converter. At high input voltage, the converter is working at higher frequency far away from resonant frequency. With the above analysis, it is concluded that parallel resonant converter has the major problems of high circulating energy and high turn OFF current at high input voltage condition. 3.6.3 Series Parallel Resonant Converter The resonant tank of series parallel resonant converter can be looked as the combination of series and parallel resonant converter. The resonant tank consists of three resonant components L r, C sr and C pr. Output filter inductor is added on secondary side to match the impedance. It combines the good characteristics of series and parallel converter. With load in series with series tank L r and C sr the circulating energy is smaller compared with parallel converter. The schematic of series parallel resonant converter is shown in Figure 3.8. Figure 3.8 Half bridge series parallel resonant converter
57 With the parallel capacitor C pr, series parallel converter can regulate the output voltage at no load condition. Similar to series and parallel, the operating region of series parallel resonant converter is also designed on the right hand side of resonant frequency to achieve zero voltage switching. The input current is much smaller than parallel converter and little larger than series converter. This means that for series parallel converter, the circulating energy is reduced compared with parallel converter. In this series parallel resonant converter circulating energy is small and it is not sensitive to load changes. Unfortunately it has a big penalty of a wide input range design. With this, the conduction loss and switching loss will increase at high input voltage. 3.6.4 LLC Resonant Converter LLC resonant converter has two resonant frequencies. Low resonant frequency is determined by series resonant tank L r, C r and high resonant frequency is determined by C r and equivalent inductance of L r and L m in series. For resonant converter, it is normally true that the converter could reach high efficiency at resonant frequency. The LLC resonant converter is shown in Figure 3.9. Figure 3.9 Half bridge LLC resonant converter
58 For LLC resonant converter although it has two resonant frequencies, unfortunately, the lower resonant frequency is in ZCS region. Now the higher resonant frequency is in the ZVS region which means that the converter could be designed to operate around this frequency. It is designed to operate at a switching frequency higher than resonant frequency of the series resonant tank of L r and C r. 3.7 MODIFIED ASYMMETRIC DC-DC CONVERTER The advantage of complementary driven converter topologies and the asymmetrical half bridge is their inherent ZVS capability. Asymmetrical half bridge is suitable for high frequency operation and therefore it is used for high power density applications. However ZVS capable topology is suitable for a particular application that depends on how well its properties match the application requirements and how easy it is to realize these requirements with minimal expense. It is to be noted that the word expense here connotes not only the cost of components, but also efficiency, volumetric efficiency and other parameters which affect the competitiveness of commercial products. The well known asymmetrical half bridge topology and its modification are shown in Figure 3.10 and 3.11 respectively. Figure 3.10 Conventional asymmetric DC-DC converter
59 Figure 3.11 Modified asymmetric DC-DC converter below. The advantages of modified asymmetric DC-DC converter are listed The voltage across the switches in the half bridge is V s that allows the use of more efficient and relatively less expensive lower voltage MOSFETs. Attainment of ZVS is easier in the half bridge because the load current reflected to the primary flows in both directions. Lower voltage across the transformer primary allows a smaller number of primary turns. This results in lower leakage inductance and makes construction of transformer easy. Amidst of all the above said advantages, it has the following disadvantages too. The distribution of voltage stress between the output rectifiers is more uneven. The tapping in the transformer secondary may cause extra loss at high frequencies.
60 Presence of DC component in the magnetizing current requires a gapped transformer core. In a modified asymmetric half bridge converter, the power train permits extension of the allowable duty cycle to D > 0.5 while changing the shape of the converters static transfer function. This change permits optimization of the stress distribution among the components. Comparison of asymmetric half bridge converter topologies shows that for relatively high input voltage applications, it offers important advantages of lower voltage across switches and easier attainment of ZVS. However, these advantages come with some drawbacks such as substantially uneven stress on the output rectifiers. The given modified asymmetric half bridge converter is a deliberate choice of uneven turns ratios in the two transformer converter circuit and mitigates this drawback while retaining the original advantageous features of the topology. As a result, lower voltage and more efficient rectifiers can be utilized. 3.8 FLYBACK CONVERTER Flyback derived converters are attractive because of their simple capacitive output filter when compared with the other converters used in cost sensitive applications. Nevertheless, hard switching operation of the power switch results in high switching loss, high EMI noises and high switch voltage stresses. Various kinds of soft switching techniques were proposed for flyback converters. Among them, the resonant converter, the active clamp circuit and the asymmetrical half bridge converter are probably the most well known converters. Resonant converter can reduce the switching losses and EMI
61 noise. However, the voltage and current stresses increase and result in high conduction losses. Incorporation of active clamp network provides the benefit of ZVS operation of the power switches in the active clamp flyback converter, but the high voltage stress on the power switch is the drawback. The asymmetrical half bridge flyback converter which can achieve ZVS operation of the power switches is gaining popularity. The switch voltage stresses not more than the input voltage can be achieved. The simplified circuit diagram of the asymmetrical half bridge flyback converter is shown in Figure 3.12. The inductor L m denotes the combination of the leakage inductance of T x and the external inductor. Figure 3.12 Asymmetric half bridge flyback converter The ZVS conditions for the power switches S 1 and S 2 are quite different. One of them is just a linear charging process. The ZVS operation of the power switch S 2 can be maintained simply when there is a sufficiently long dead time between S 1 and S 2. However, the ZVS operation of the other switch S 1 is achieved only when the energy stored in the resonant inductor is greater than that of the output capacitors of these power switches.
62 Not only the magnetizing inductor of the transformer and the resonant inductor, but also the blocking capacitor also stores energy when the output rectifier is OFF. When the duty cycle increases, the energy stored in the blocking capacitor will also increase. 3.9 SUMMARY The asymmetrical half bridge converter is desired to operate with 50% duty cycle and achieve maximum efficiency. The switch voltage stress will never be more than the input voltage. By analyzing the features of existing converters, this work focuses on half bridge isolated DC-DC converter because of its advantages like, Low switching loss due to zero voltage switching capability of primary side switches. Low voltage stress in secondary side rectifier due to its zero current switching. Considerable reduction in the number of bulky capacitors due to high voltage gain.