Development of a Dual-Frequency, Dual-Polarization, Flexible and Deployable Antenna Array for Weather Applications Dimitrios E. Anagnostou, Member, IEEE, Ramanan Bairavasubramanian, Student Member, IEEE, Gerald DeJean, Student Member, IEEE, Guoan Wang Student Member, IEEE, Nickolas Kingsley, Student Member IEEE, Manos Tentzeris, Senior Member, IEEE and John Papapolymerou, Senior Member, IEEE Abstract The development of a dual-frequency, dual polarization, aperture-fed, multiple-layered microstrip antenna array on flexible organic material for System-on-a-Package (SoP) RF front ends is described in this work. The integration of RF MEMS phase-shifters with the array enables accurate beamscanning over the earth s surface. The flexible light-weight organic substrate minimizes the production cost while it enables the structure to be folded or rolled-up thus saving space and weight; elements determinant in a satellite mission. T Index Terms Antenna, Array, MEMS, I. INTRODUCTION HE global water cycle is critical for the functioning of the Earth s system, as it integrates the physical, chemical and biological processes that sustain ecosystems and influence climate, weather, natural hazards and related global changes. Water evaporation, precipitation and water vapor feedbacks alter the surface and atmospheric heating and cooling rates, which lead to adjustments in atmospheric circulation and precipitation patterns processes that are not adequately taken into account by currently used climate models. Sufficient understanding of such phenomena and processes is a key source for accurate prediction of the Earth s climate through the development of models that respond to the water cycle s variability and changes that take place. Most important, from the forces acting on the planet s climate, it is estimated that 40% are due to human actions and 60% are due to feedback effects on precipitation and temperature changes [1]. For these reasons, water cycle research is a high-priority area of research. Weather monitoring can be achieved from geostationary satellites. The major issue for all space missions is the minimization of equipment s weight and cost, including the antennas. To monitor precipitation patterns, NASA requires dual-frequency (14/35 GHz for rain and 19/37 GHz for snow), dual-polarization radar and radiometers. Moreover, it is desirable that the systems simultaneously measure the precipitation over the same area, which implies that the antenna radiation pattern should be the same for both frequencies and polarizations. High-gain antennas used in most space missions are traditionally implemented as parabolic reflectors that are bulky, heavy, difficult to deploy, very expensive and have a limited scanning capability. To circumvent these problems several researchers focused on printed [2] or even inflatable [3] reflectarray antennas that consist of an illuminating feed and a thin reflecting surface, and can offer a low-mass, lowcost, high-packaging efficiency in future high-gain spacecraft antennas. Current limitations of this technology include low antenna efficiency, difficulty in maintaining the uniform membrane spacing and surface flatness, increased sidelobe levels and single-frequency operation. While past efforts have developed deployable antenna structures, they have not been dual-frequency and dualpolarization, and past antennas that met those goals were nondeployable or were based on large weight/volume and highcost technology. NASA s latest efforts in collaboration with the Georgia Institute of Technology are in the development of advanced, low-cost, low-mass, deployable antennas with large surface that can be rolled-up or folded for launch and then deployed in space. Electronic scanning and shaping of the radiation patterns of the two beams at the two frequencies is desirable in order to cover the same area simultaneously. In this work, a dual-frequency, dual-polarized antenna array on a flexible Liquid Crystal Polymer () organic substrate is developed. can be processed around 300 o C, has excellent electrical properties, very low moisture absorption, can be combined in multiple layers, and can be folded or rolled. Fig. 1 shows how in this multi-layer substrate, the complete feed network is integrated with the dual-polarized microstrip patch antennas for each frequency band. Additionally, electromagnetic band gap (EBG) structures embedded in the substrate can reduce the coupling between the two antenna sets and the feed network. Lastly, RF MEMS phase shifters and switches are integrated onto the feed network to control the RF signal path, to select between the two available polarizations and to permit wide beam scanning.
II. ANTENNA AND ARRAY DESIGN AND FEED MECHANISM The substrate used in this work is. Some of the advantages of this organic substrate include low dielectric loss (tanδ ~ 0.004), constant dielectric permittivity at the frequencies of interest (εr ~ 3.15), low moisture absorption (<0.02%), light weight, mechanical stiffness, thermal stability (CTE = 0-30 ppm/ C), chemical resistance, ease of mass fabrication and great flexibility which allows for the material to be rolled up, which is ideal for circuits and structures that need to be deployed in space. In order to implement the dual-frequency/dual-polarization array, we adopt a two-layered sub-array structure consisting of two 2 2 uniformly spaced square microstrip antennas. The 14 GHz array is square, physically larger, and is placed at the interface of the air with. The 35 GHz array is smaller and is embedded underneath, between two layers of of equal thickness t=250 µm. The array design and feed network are shown in Fig. 2. Even though parasitic coupling between the two sub-arrays of this configuration was insignificant, it can be further reduced if one sub-array has a cross square configuration rotated by 45 degrees with respect to the shape of the other. Full-wave simulation results validated this. Additionally, there were very small differences in the radiation patterns and return loss of the 35 GHz array with and without the presence of the 14 GHz patches and vice versa. The element separation distances were adjusted in order to achieve similar patterns at both frequencies (identical scanning trace) and to confine the cross polarization radiation and the sidelobes below 25 db, thus allowing for a scanning range larger than 30 degrees. As this subarray employs only canonical shapes, it can be cascaded in both longitudinal and transverse directions. A theoretical investigation illustrated that 12 subarrays of this type can provide a beamwidth smaller than 3 to a specific direction. In order to feed the antenna elements and to extend the array size with additional antennas in both directions, a separate feed layer is needed and is placed underneath the antenna layers. The antennas are fed by aperture coupling via slots in the ground plane. This method has low parasitic radiation from the microstrip feed network below the slots. A cross-section of the structure along with the feed design is shown in Fig. 2. A picture of the different layers of the fabricated antennas, apertures and feed network is shown in Fig. 3. In Fig. 4, a different type of feeding is shown, which can enhance component integration capability in large arrays by providing more space for component placement. The available component space on each 2 2 sub-array can be more than 2cm x 0.8cm, which fits in series a phase shifter along with a bandpass filter. This way, unwanted signals are filtered out. A schematic of this integration is shown in Fig. 5. A critical step in the fabrication process is the bonding of the substrates, which takes place using a 1-mil low-melt temperature (type-i ) bond layer in-between the two 4-mil high-melt temperature (type-ii ) core layers. The bond 12 mm 6 mm 14 GHz patches 35 GHz patches (embedded) 7 mm Ground 2 mm Layer 3 Layer 2 Layer 1 Fig 1 Topology of the dual-frequency, dual-polarization flexible 2 2 antenna sub-array. The 35 GHz elements are embedded between the two layers. The substrate layers have different color for illustrative purposes only. The dimensions of the structure are also shown. a) b) Fig. 2. a) Cross-section of the array layers, b) Final Array layout. All layers are visible so that the antennas, the slots and the feed network structure can be observed. Fig. 3. Picture of the fabricated antenna array layers on. The apertures, the 35 GHz array, the 14 GHz array and the feed network are shown.
Fig. 4. Schematic showing the available space on a 2x2 single-frequency subarray using a corporate aperture coupling feed. Fig. 6. a) Effect of the element factor (EF) of microstrip antenna elements on the radiation pattern of the 26x2 array. Fig. 5. Beam-steering capability of the 26x2 array, using one phase-shifter every each 2x2 sub-array. layer melts at a lower temperature than the core layers and its flow coupled with the tool pressure applied between the core layers results in multilayer structures. The cost of fabrication is low due to s low processing cost, while low deployment costs result from the capability to roll/unroll the substrate and thus the structure. The bonding process was optimized for temperature and tool pressure to prevent shrinkage, formation of bubbles, and melting of the core layers. Alignment of the layers, which is a critical procedure for this design, was performed using laser-drilled holes with positional accuracy of 25 µm or better. III. ARRAY BEAM SCANNING Radiometry and radar antenna systems need to be able to scan their radiation pattern over the planet surface to obtain the data required by the NASA ESE. Historically, GaAs MMIC phase shifters are used, but they have high insertion loss, which degrades system sensitivity. In addition, power requirements of GaAs MMICs can be a concern in the design of airborne or space borne systems. In this work, MEMS phase-shifters on with low loss and negligible DC power consumption are used. Additionally, RF MEMS routing switches integrated on the same layer with the feed network that can be controlled digitally/electronically via a microprocessor enable the polarization selection (E-pol vs. H- pol). The switch and phase shifter design and measurements are described in [4] thus they are not mentioned here. We focus instead on the beam steering that can be achieved with the entire array structure. It is anticipated that the 4-bit phase shifter per 2x2 subarrays will allow a minimum scanning range of 35 degrees for the 2x26 antenna array. The array s pattern main cut is shown in Fig. 6 for a uniform excitation. The main lobe has a 3dB beamwidth of 3.5, which constrains the main beam s steps to those smaller or equal to this angle. This can also define the number of bits needed per phase shifter, which in this case was found to be 4. A larger number of bits would provide smoother beam steering, while a smaller one would provide a more abrupt beam steering which cannot be used unless a 10dB beamwidth is acceptable. As shown in Fig. 7, by using a 4-bit phase-shifter, smooth steps for beam rotation up to an angle of 18 degrees are achieved, without the formation of significant sidelobes. The noticed decrease in the amplitude of the pattern is due to the element factor of the antenna elements that was taken into account. IV. RESULTS For measurement purposes, the dual-frequency array was excited at one frequency, while the other was treated as parasitic. The array s insertion loss and patterns were measured at both frequencies and are shown in Figs. 8 and 9 respectively. The results show a frequency shift of 200 MHz for the 14 GHz and of 250MHz for the 35 GHz array. The 14 GHz array s return loss is slightly smaller than the simulated, possibly due to mismatch at the input terminals between the connector and the feeding structure. The measured impedance bandwidths at both frequencies are in good agreement with the simulated. These results are summarized in Tables I and II. Additionally, simulated and measured 2D radiation patterns are shown in Fig. 9(a, b) and Fig. 9(c,d) (for the E- and H- plane at 14 and 35 GHz respectively), and are summarized in Tables III and IV. It is noticed that the E and H-plane beamwidths and patterns are consistent with the simulations for the 14 GHz array. The center-to-center distance can be further increased in order to reduce the E-plane beamwidth to a value close to the H-plane beamwidth if desired, but this will result in the formation of more significant of side-lobes. A photo of the antenna array under measurement setup, in NASA
a) b) Fig. 7. Beam-steering capability of the 26x2 array, using one phase-shifter every each 2x2 sub-array. 0 5 10 S 11 [db] 15 20 25 a) b) S 11 [db] 30 Simulated Measured 35 12 12.5 13 13.5 14 14.5 15 15.5 16 Frequency [GHz] 0 5 10 15 20 25 30 35 Simulated Measured 33 34 35 36 Frequency [GHz] Fig. 8(a, b). Simulated and measured return loss of the array at a) 14 GHz, and b) 35 GHz. Glenn Research Center, OH, is shown in Fig. 10. With this the measurement setup, it was not possible to measure the backside radiation accurately, due to the interference of the mounting post. c) d) Fig. 9(a-d). a) E-plane radiation pattern of the 14 GHz array. b) H-plane radiation pattern of the 14 GHz array. c) E-plane radiation pattern of the 35 GHz array. d) H-plane radiation pattern of the 35 GHz array. Table I Return Loss Characteristics of the 14 GHz Array Resonant Frequency 14 GHz 13.72 GHz Return Loss 30.7dB 16.5 db Bandwidth 140 MHz 160 MHz Table II Return Loss Characteristics of the 35 GHz Array Characteristic Simul. 35 GHz Measured 35 GHz Resonant Frequency 34.87 GHz 34.32 GHz Return Loss 32.5dB 39.6 db Bandwidth 1560 MHz 1530 MHz Table III Radiation pattern characteristics of the 14 GHz array E-Plane Θ bw (-3 db) 65 0 67º H-Plane - Θ bw (-3 db) 58 0 58º Max.E-plane cross-pol. level -31 db -16 db Max.H-plane cross-pol. level -33 db -25 db Table IV Radiation pattern characteristics of the 35 GHz array E-Plane Θ bw (-3 db) 65 0 66º H-Plane - Θ bw (-3 db) 59 0 59º Max.E-plane cross-pol level -15 db -13 db Max.H-plane cross-pol. level -16 db -15 db
Fig. 10. A photo of the multilayered array under measurement setup. Only the 14 GHz patches can be seen as the other elements of the structure are embedded between the dielectric substrates. V. CONCLUSIONS A dual-frequency, dual-polarization microstrip antenna array for System-On-a-Package (SOP) RF front ends was presented for the first time on a flexible multilayer substrate. The design was simulated using commercial full-wave simulation software. The array was fabricated and the measured performance with respect to its scattering parameters and radiation patterns was analyzed. The measurements showed return loss of better than 20dB, with good cross-polarization characteristics. These designs can be extended to arrays of larger dimensions. The results shown here pave the way for the development of low cost, lightweight, and low power RF front ends and antennas on an all-package solution for future communication and remote sensing systems. REFERENCES [1] Environmental Protection Agency, US Climate Action Report 2002, available at: http://yosemite.epa.gov [2] D.M. Pozar and T.A. Metzler, Analysis of a reflectarray antenna using microstrip patches of variable size, Electronics Letters, pp. 657-658, Apr. 1993 [3] J. Huang and A. Feria, Inflatable microstrip reflectarray antennas at X- and Ka-band frequencies, IEEE AP-S Symposium, Vol. 3, pp. 1670-1673, June 1999 [4] N. Kingsley and J. Papapolymerou, Multibit MEMS Phase Shifter on Flexible, Organic Substrate for Microwave Antenna Systems, 15th IST Mobile and Wireless Communication Summit, 4-8 June, 2006, Mykonos, Greece.