Link Efficiency-Led Design of Mid-Range Inductive Power Transfer Systems Christopher H. Kwan, George Kkelis, Samer Aldhaher, James Lawson, David C. Yates, Patrick C.-K. Luk, and Paul D. Mitcheson Department of Electrical and Electronic Engineering, Imperial College London, United Kingdom Power Engineering Centre, Cranfield University, United Kingdom Email: christopher.kwan9@imperial.ac.uk Abstract For mid-range inductive power transfer (IPT) systems, improving link efficiency entails operating in the multi- MHz region in order to increase coil Q factors. However, designing end-to-end systems at such frequencies poses challenges associated with the efficiency of the power electronics. This paper presents a set of design principles with the aim of achieving maximal DC-to-load efficiency of such systems. se include the selection of semiconductor devices and power converter topologies that are suitable for high frequencies. Through these design methods, a 6.78 MHz ISM-band IPT system has been implemented, transferring W of power across cm with a DC-to-load efficiency of ~7 %. Index Terms DC-to-load efficiency, converters, mid-range wireless power I. INTRODUCTION In this paper, design principles for maximising the efficiency of a mid-range inductive power transfer (IPT) system are detailed. se principles address the challenges of operating at multi-mhz frequencies due to the potential for significant switching losses of the power electronics in the MHz region. methods include choosing appropriate types of semiconductor devices and selecting suitable converter topologies for the power electronics. Fig. shows a block diagram of a typical IPT system. Section II explains the rationale behind operating in the MHz range in order to improve link efficiency with air-core coils. Section III describes the choice of semiconductors available for high frequency power electronics. Section IV explains the use of the Class-E inverter in semi-resonant operation mode, the Class-E inverter with a saturable reactor, and the Class-EF or Class-E/F inverter, in order to drive the transmitter coil. Section V presents Class-D and Class-E rectifiers as efficient means of rectifying the high frequency AC voltage of the receiver coil. Section VI introduces load emulation in order for the IPT system to operate at maximal efficiency regardless of the actual impedance of the connected load. Section VII outlines the guidelines and regulations on exposure of humans to electromagnetic fields, which can also influence the design and usage scenarios of IPT systems. Section VIII highlights results from an implemented 6.78 MHz mid-range IPT system which was designed with these principles in mind. 6.78 MHz was selected as the operating frequency because it is the first ISM band in the MHz region; an example of an IPT application DC Power Supply Inverter Rectifier Load Emulation Fig.. Block diagram of an inductive power transfer system Load at this frequency is the charging of personal devices (e.g. mobile phones) where portability and being lightweight are highly desirable characteristics of the wireless charging system. Section IX concludes by summarising the methods described in this paper for designing mid-range IPT systems with the aim of maximising link efficiency. II. MAGNETIC DESIGN Coils with ferrite cores can be heavy and thus not very portable; in [], a khz system transferring.5 kw over 7 mm uses an H-shaped ferrite core which weighs.9 kg. Also, their directed magnetic flux leads to a restricted freedom of movement for both the transmitter and receiver sides due to the need for accurate coil alignment. refore, there are many situations in which air-core coils, with their wide flux coverage, are more suitable for wireless power transfer applications. In order to achieve high-q factors with an air core, MHz frequencies are necessary to maximise the coil Q factors. With the coils acting as a weakly-coupled air-core transformer, efficiency can deteriorate rapidly with distance. To maximise link efficiency, receiver resonance should be used to cancel the secondary leakage inductance, and the optimal load should be connected to the receiver s resonant tank. For a parallel resonant secondary, the optimal load is given by (), where k is the coupling factor, Q TX is the Q factor of the transmitter coil, Q RX is the Q factor of the receiver coil, ω is the frequency of operation and C RX is the secondary tank capacitance chosen to resonate with the receiver coil []. optimal load for a series resonant secondary is given by (), where L RX is the inductance of the receiver coil []. R opt,par = ωc RX ( ) Q RX + k Q TX Q RX ()
R opt,ser = ωl RX ( + k Q TX Q RX Q RX As a result of using either form of secondary tank resonance, the maximum link efficiency can be evaluated using () []. η link = ) () k Q TX Q RX ( + + k Q TX Q RX ) () From (), it can be seen that the Q factors of both the transmitter and receiver coils influence the maximum link efficiency. refore, to achieve improvements in the link efficiency, the Q factor of the coils should be maximised. This can be done by increasing the frequency of operation of the IPT system, but only up to a certain point after which the coils far-field radiation begins to dominate (causing the coils Q factor to drop) []. However, it does not necessarily follow that the overall efficiency of the system will be increased as well, as switching losses of the power electronics in both the inverter and rectifier circuits rise with frequency. refore, efficient high-frequency soft-switching power electronics are desired. se circuits, which will be described in more detail in Sections IV and V, rely also on fast devices which will be described in the next section. advantages of air-core coils can also be seen in longer range applications (up to m between TX and RX coils) where a network of sensors with mlliwatt-level consumption could be remotely powered [5]. In these situations, the superior tolerance to angular offsets and transverse displacements compared to coils with ferrite cores is essential to being able to supply power to multiple sensor nodes concurrently. Furthermore, the reduction in size and weight of the coils due to the absence of the ferrite core means that the wireless sensors can be kept small and lightweight. III. SEMICONDUCTORS Due to the high frequency, high voltage and high current requirements of the power electronics, the task of selecting appropriate semiconductor devices is not trivial. As midrange high power IPT systems operate near the limits of the capabilities of traditional Si devices, wide-bandgap semiconductors such as SiC and GaN ought to be considered due to their superior characteristics as power devices, e.g. faster switching rates and higher breakdown field strengths. Alternatively, specialist high-speed RF MOSFETs such as those from IXYS RF can be incorporated into designs. For example, the IXYS RF IXZDFN has a total rise and fall time of 7.5 ns [6], whilst the combined rise and fall times of highvoltage MOSFETs from International Rectifier are typically at least ns [7]. An IXYS RF power MOSFET combined with gate driver (IXZDFN) was used in the Class-E inverter of the mid-range IPT system that was designed and implemented (see Section IV). Cree SiC Schottky diodes (CD7 and CD6) were used in the Class-D and Class-E rectifiers (see Section V). packaging of the devices used in the power electronics can influence the performance of the inverter and rectifier circuits. Cree CD7 SiC Schottky diode in the rectifier comes in a TO-7- package [8], which has long, narrow leads, adding stray inductances to the circuit. Contrastingly, the IXYS RF IXZDFN module for the inverter has a surface-mount low-inductance package which means that parasitic effects, which could potentially reduce the switching speeds, can be minimised. It also has a low intrinsic gate resistance, leading to a decrease in rise and fall times, and enabling faster switching of the devices. IV. INVERTER Conventional hard-switching inverters are not suitable for IPT systems when operating in the MHz region. Since the switching time of the devices becomes comparable to the period of the driving signal, the result is that they can be inefficient at higher frequencies. Soft-switching inverters, such as Class- D and Class-E inverters, address this issue by employing zero-voltage switching to minimise power dissipation in the MOSFET during switching. This achieved by preventing concurrent high voltage across and current through the MOSFET. A disadvantage of Class-D inverters, which are popular with low-power systems adhering to Qi or AWP standards, is that they have lower output power compared to Class-E inverters for the same voltage and output load. Another issue is that they require a floating gate drive due to the presence of a high-side switching device. However, in contrast to Class-E inverters, Class-D inverters are able to operate over a larger load range with zero-voltage switching if the switching frequency is below the resonant frequency of the output load network. Fig. depicts the Class-E inverter in semi-resonant operation mode [9] used to drive the transmitter coil. In this topology, the transmitter resonant tank is tuned to a slightly higher frequency than the secondary resonant tank to keep the primary tank impedance inductive, a requirement for Class-E operation. parallel combination of the capacitor C res (in Fig. ) and the transmitter coil forms an impedance transformer, which causes the load impedance to appear larger, leading to an increase in driver efficiency. Fig. shows the simulated drain-source voltage of the MOSFET of the Class-E semi-resonant inverter. Class-E inverters may also include a saturable reactor [] to tune for optimum switching operation when a change in the load occurs (see Fig. ). A saturable reactor is essentially an AC-to-AC transformer that consists of a primary and a secondary winding, both wound on a single magnetic core. It operates by applying a low DC current in one winding, which causes the magnetic core s permeability to decrease, and therefore effectively changing the impedance of the second winding. tuning procedure relies on varying the switching frequency and the effective reactance of capacitor C in Fig. via the saturable reactor. Although Class-E inverters can achieve zero voltage and current switching operation, their voltage and current stresses can be large compared to other inverters. It has been reported in [], [] that adding a series LC resonant network in parallel
Fig.. Semi-resonant Class-E inverter from [9] Fig. 5. Circuit diagram of the Class-EF or Class-EF inverter. Inductor L represents inductance of transmitting coil. R L represents 6 reflected load seen by inverter in addition to coil ESR. [] 6 6 TABLE I COMPARISON BETWEEN DIFFERENT RESONANT INVERTER CLASSES Class-E.5768 Class-EF.556........ Class-E/F...76 time (ms) time (ms) (a) k=.5 optimum operation (b) k=. coils further apart Fig.. Simulated drain-source voltage for semi-resonant Class-E inverter against time in µs [9] network has been referred to as the Class-EF inverter when the V added resonant LC network is tuned to the second harmonic, or the Class-E/F inverter when the added resonant LC network RFC is tuned to the third harmonic. Fig. 5 shows the circuit diagram r C C of the Class-EF or -E/F inverter; inductor L and capacitor C form the added resonant network and their values are set C DC r r r LP suchr Q LS that their C CP r resonant frequency is either twice or three times CS f s Mthe switching frequency []. Fig. 6 compares the waveforms of the Class-EF and Class-E/F R L inverter with the Class-E inverter. V GS C L Sat C P C L P Class-EF LS S inverter results in lower voltage stresses whereas the Class-E/F inverter results in lower current stresses through the MOSFET. Table I shows a comparison of the normalised output power of the Half-Bridge Class-D ZVS, the Class-D ZCS, the Class-E, Fig.. Class-E inverter with a saturable reactor [] the Class-EF and the Class E/F inverters. 6 Inverter Class Normalised Output ( ) PoR L Power Vi Half-Bridge Class-D ZVS.6 Class-D ZCS.98 Al of nh/turn. control windings with the MOSFET can reduce its voltage or current stresses and therefore improve the efficiency of the inverter. added LC network is tuned to either the second or third harmonic of the µh for each toroid. DC control switching frequency. Adding resonant networks in inverters is a common technique used in Class-F and Class-F - inverters to shape the MOSFET drain voltage and current waveform. This hybrid configuation of Class-E switching with a resonant V. RECTIFIER VALUES AND RANGES OF SEVERAL PARAMETERS OF THE CLASS E INVERTER AND THE INDUCTIVE LINK MEASURED AT 8 KHZ As with hard-switching inverters, rectifiers can suffer from significant diode reverse recovery losses in the MHz region if they are hard-switched. use of soft-switching rectifiers avoids the requirement of a hard recovery and almost eliminating the associated losses with re-establishing reverse blocking function. Class-E rectifiers are soft-switching topologies; one Component/Parameter Value ESR Value L P 5.76 µh r LP.7 Ω L S 6.69 µh r LS.8 Ω C S (Polypropylene) 5.9 nf r CS.5 Ω R - kω - -
v DS VIN v DS VIN v DS VIN π π π π π π i S IIN i S IIN i S IIN π π π π π π (a) Class-E (b) Class-EF (c) Class-E/F Fig. 6. Comparison of normalised voltage and current waveforms of different inverters [] v in i in v Lr + L r v Cr + i Cr C r D r i Dr C st i Cst R dc Fig. 7. A voltage-driven low dv/dt class-e rectifier [] I dc + V dc such circuit topology with low dv/dt that is voltage-driven is shown in Fig. 7 []. This specific Class-E topology contains an inductor L r in series with a parallel connection of a capacitor C r and a diode D r. inductor L r is in resonance with the capacitor C r at the operating frequency of the system. refore, the L r -C r - D r connection provides half-wave rectification. Output filtering is performed by the first-order low-pass filter consisting of stabilising capacitor C st and the output load R dc. Any leakage inductance from the secondary coil can be absorbed into L r. In addition, the diode-capacitor parallel combinations means that the diode s junction capacitance can be absorbed into C r. When designing a Class-E rectifier, the primary objective is to ensure that the rectifier s impedance, R in is equal to the optimal load R opt. Since the rectifier depicted in Fig. 7 is a voltage-driven Class-E rectifier for a parallel-tuned secondary, the optimal load is given by R opt,par in (). impedance R in relates to the output load R dc of the Class- E rectifier by (), where M is the AC-to-DC gain [5]. R dc = M R in () Fig. 8 shows the waveforms of the Class-E rectifier. top plot is the rectifier s diode voltages, whilst the bottom plot shows the current through the diode D r (dotted) and the capacitor C r (solid). Another type of Class-E rectifier is shown in Fig. 9 []. This half-wave low dv/dt Class-E rectifier is current-driven, so it is suitable for a series tuned receiver. This rectifier consists of a capacitor-diode network connected to a second-order output filter. This filter is made up of an inductor L f, a capacitor C f and the output load R dc, and provides load independent filtering. For this rectifier, C d acts as a snubber capacitor and therefore ensures zero voltage at turn-on and turn-off and zero rate of voltage change at turn off. relationship between the impedance R in of this current-driven rectifier and its output load R dc is given by (5), where K I is the AC-to-DC current gain. R dc = R in K I Although conventional hard-switching rectifiers can be inefficient in the multi-mhz region, Class-D rectifiers with SiC Schottky diodes have shown to be usable in the MHz region. An example of such a topology is shown in Fig., which is a half-wave Class-D rectifier consisting of two diodes []. It is also current driven, making it appropriate for series resonant receivers. impedance of this current-driven rectifier (5)
Diode Voltage [V] Diode (solid) and Capacitor (dotted) Current [A] 5,,5 9.75 9.8 9.85 9.9 9.95 Time [s] 6 s-d and Class-E Half-Wave Rectifiers ower IPT Applications VI. RECEIVER LOAD EMULATION As shown in (), (5) and (6), the Class-D and Class-E rectifiers have an resistance which depends on the load attached to its output. refore, in order to have maximum link efficiency, a load emulation circuit should be attached to the output of the rectifier, so that the rectifier (and the receiver resonant tank) is always presented with the optimal load, given by () for a parallel tuned secondary and () for a series tuned secondary. Otherwise, different levels of current draw in the load would detune the magnetic link, resulting in a drop in efficiency and received power. Such a circuit may take the form of a DC-to-DC converter, such as the Buck converter, which can be controlled to achieve.5 optimal loading. This is done by measuring the voltage at the output of the rectifier and dividing by the desired load in order. Yates, Paul D. Mitcheson to obtain the current demand. This current demand is then l and Electronic Engineering compared with the actual measured current in order to give an ollege London error signal with which the duty cycle of the Buck converter s@imperial.ac.uk 9.75 9.8 9.85 9.9 9.95 can be adjusted to emulate the desired load. Time [s] 6 In addition, from () and (), it can be seen that optimal load varies with the coupling factor, which in turn changes with distance between the transmitter and receiver coils. Hence, the s, the Fig. current 8. Simulated flowing voltage-driven through Class-E D allowing rectifier waveforms: an ac current [Top] Voltage to flow load emulation circuit should be controlled in such a way that s. in the across receiving D r; [Bottom] coil. Current through D r (solid) and C r (dotted) [] the optimal load is always being emulated even as distance r C o f and I dc cause a square wave voltage across the blockingchanges to ensure maximal link efficiency. L f therefore functions as the current source applying pow r diodes. square wave voltage could i Lf be a potential source In case of longer range systems (up to m between the rectifier under test [7]. c of switching losses in cases where the reverse recovery i Cd i D + + i I timetx and RX coils) such as those for wireless sensor network dc s of the semiconductor is not insignificant Cf compared to theapplications, the coupling coefficient is extremely low, which i in C d D v D C f R dc V means that the optimal load varies little with distance and period of the current. Despite the potential switching dc r, is dominated by the complex conjugate of the receiver s losses, the topology has a high output power capability as the resonant tank impedance. In these scenarios, C Rx L receiver power is diodes are stressed to the current during conduction and Rx typically very small (in the tens of mw i in range), so an open- type ofvload in emulation circuit + is preferred; this type stressed to the Fig. output. Class-E voltage Low when dv/dt Half reverse Wavebiased, Rectifiergiving goodloop Fig. 9. Current Driven Class-E Low dv/dt Half Wave Rectifier [] semiconductor utilisation. of circuit eliminates the power consumption of any active e control circuitry required for a closed-loop v in Rectif system. ierexamples under test of n As the capacitor-diode network D is shunting the current suitable ultra-low power open-loop load emulators include the a source and the second order outputi D filter, the current flowing buck-boost and the flyback converter operating in discontinuous t through the diodei and+ capacitor is the + D v D i ac current I dc + conduction mode [6]. Cf superimposedi in Fig.. Rectifier Test Rig s on the output D dcv D current. refore C the soft switching VII. ELECTROMAGNETIC FIELD LIMITS AND f R dc V dc e property of the topology increases conduction losses as the REGULATIONS real power delivered from the inverter to the rest of t f diode current exceeds the resonant tank current, when the Incircuit addition (average to the challenge power) of operating can be efficiently calculatedinbythemultiplyi d tank current is negative (Fig. ). Furthermore, the peak diodemhz region, the design of IPT systems must also consider the rms of the square voltage, the ac current throu Fig.. e Fig. Current. Class-D Driven Class-D Half Wave Half Wave Rectifier Rectifier [] limits on electromagnetic (EM) field levels that are in place voltage during reverse bias is larger than the output voltage the resonant tank ( current) and the phase difference, in order to protect humans from the adverse health effects of and consequently, device utilisation is poorer than in the Class- measured by the Power Analysis utility of the oscilloscop s D circuit. Nevertheless, impedance soft of the switching topology allows is functionally the utilisation resis-exposurtive, to EM fields. One of these thermal effects, which is affected only by its output load, and the relationship is given of are caused Using by thetissue power heating delivered throughtoenergy the dc absorption load (output from powe s large by i.e. diodes, (6). the fundamental slower than the frequency frequency component of operation, of the as reverse square EM fields the efficiency in the tissue. of theother receiving is non-thermal end of effects, the IPTcaused system can d voltage recovery across effects the are current largely source eliminated. is in phase with the by the calculated. stimulation This of efficiency muscles, nerves also includes and sensory losses organs. in the Rx c r current component and hence, stress the rectifier and the has a resistive impedance of impedance. R dc = π R the circuit (6) Working and at thus, 6.78part MHzof means thethat inductive both thermal link and efficiency. non-thermal Furthermo Furthermore, depend on the it is duty frequency cycle, which independent is affected if the by parasitic R dc, C capacitances d andeffects R need in can to be be determined taken account from of when the designing values of such IPT power a the frequency of advantages the of diodesof operation. have the negligible Class-D rectifier, Class-E impedance which rectifier presented compared include a has tosystems. current. When the inverter voltage and curre the simpler ) an impedance design impedance of process, the of resonant lower a series connection tank cost capacitor. implementation, between a dc higher capacitor load is A European Union (EU) Directive [7] was adopted on 6 are in phase the impedance at the Rx end is resistive a the tolerance (C only in ) and component to DC load a resistor (R that in ) affects variation [], [5]. theand When better the ac rectifier resistance semiconductor is added (R in ) June by the European Parliament and the Council of utilisation, are more noticeable in higher voltage operation, Europe, represented which sets byout thelimits onresistance the exposure of the of workers rectifier. to which to an IPT must system equal to R R ac,ser in must when be evaluated the rectifier for maximal is integrated where semiconductor parasitic effects are minimised. linkem fields. This Directive is to be transposed into UK law by e to B. Experimental Results efficiency an IPT and system. C in must following be taken expression into consideration relates the when two n resistances tuning the receiving []. coil. Designers have a degree of freedom experiments investigated the efficiency of the select e in selecting a duty cycle R value. other components are topologies under several resistance designs. Cree S s dc = π R in () hence evaluated in [] as: schottky diodes (CD6) were used for high power op B. Class-E Topology R dc = R ation at 6.78 MHz. in KI () evaluation of R ac,ser was made using the informati S 978--799-66-/5/$. current driven Class-E 5 low dv/dt IEEE rectifier (Fig. ) is composed of a capacitor-diode network connected to a second provided in [] and []. Using () to () the components of t order filter. second order filter C d = Q rectincludes a filter inductor rectifiers were set to: R dc = 7 Ω for the Class-D, and R dc () o (L f ), a filter capacitor (C f ) andω the R dc load (R dc ). L f ensures Ω and C d = 7 pf for the Class-E rectifier. Those valu
July 6. Prior to the passing of this Directive, there have been no statutory limits on EM fields for both workers and the general public in the UK. safety limits described in the directive are based on the 998 and ICNIRP limits [8], [9]. Directive defines both exposure limit values (ELVs) and action levels (ALs). ELVs (which ICNIRP calls basic restrictions) are quantities that are directly related to established health effects (i.e. tissue heating and nerve stimulation). se quantities, which are generally difficult to measure, must not be exceeded. particular ELV which relates to thermal effects is the Specific Absorption Rate (SAR), whilst the ELV for non-thermal effects is the internal electric field induced in the body. At 6.78 MHz, the SAR limit is. W kg (whole body, averaged over 6-minute period and g of tissue) and the induced internal electric field limit is 576 V m (peak). Because ELVs are difficult to measure directly, the Directive also defines ALs (referred to as reference levels by ICNIRP). se external quantities, which can be measured, are external electric field and external magnetic field. At 6.78 MHz, the magnetic field ALs are µt for non-thermal effects (induced internal electric field) and. µt for thermal effects (SAR). Compliance with these ALs ensures compliance with the respective ELVs. However, if the ALs are exceeded, it does not necessarily follow that the ELVs will be exceeded as well. In these cases, further tests are needed to prove compliance with the ELVs such as performing D EM simulations. In reality, depending on the power requirements of the application, it may not be possible to deliver enough power to a load whilst at the same time keeping magnetic field levels within the EU Directive AL limits. Hence, it would be necessary to define an exclusion zone, outside of which it would be safe for humans to be physically present. Nevertheless, these EM field limits and regulations imply that it is important to design IPT systems with high link efficiencies, so the required level of power can be delivered to the receiver load with minimal magnetic field. VIII. EXPERIMENTAL RESULTS A 6.78 MHz mid-range IPT system capable of transferring W of power across a distance of cm with a DC-to-load efficiency of ~7 % has been implemented and demonstrated. This system was designed with the principles introduced in this paper. experimental setup is shown in Fig.. transmitter coil is a cm -turn air-core coil made from copper piping. A Class-E inverter in semi-resonant operation was selected to drive the transmitter coil, with the IXYS RF IXZDFN combined gate driver and MOSFET module used as the switching device. receiver coil is a cm 5-turn air-core coil, also made from copper piping. chosen rectifier topology is the voltage-driven Class-E low dv/dt rectifier, utilising the Cree CD7 SiC Schottky diode. se methods have resulted in a reduction in losses and an improvement in efficiency of the power electronics. Consequently, this mid-range IPT system is able to operate Fig.. Experimental setup of 6.78 MHz mid-range IPT system feasibly in the MHz region, leading to an increase in coil Q- factors, link efficiency and overall DC-to-load efficiency. IX. CONCLUSION design of a lightweight and portable IPT system calls for the use of air-core coils in favour of coils with ferrite cores. weak coupling of air-core coils suggests that the operating frequency should be increased to the multi-mhz region to maximise link efficiency. However, the efficiency of the power electronics will tend to decrease with frequency, unless suitable high frequency power converters are utilised. Inverters that are appropriate for multi-mhz frequencies include the Class- E semi-resonant inverter, the Class-E inverter with a saturable reactor, and the Class-EF or Class-E/F inverter. Class-D or Class-E rectifiers may be used to rectify the high frequency coil voltage. Different types of semiconductor devices need to be considered, including GaN, SiC and specialist high-speed RF Si devices. Receiver load emulation may be needed to maintain maximum link efficiency by emulating the optimal load seen by the rectifier. regulations on human exposure to EM fields may also influence IPT system design and usage scenario, and ultimately encourage the design of a highly efficient system. By following the link efficiency-led design principles in this paper, a mid-range IPT system can be designed with maximum link efficiency and overall DC-to-load efficiency. ACKNOWLEDGMENT authors would like to acknowledge the Department of Electrical and Electronic Engineering, Imperial College London for financial support. REFERENCES [] M. Chigira, Y. Nagatsuka, Y. Kaneko, S. Abe, T. Yasuda, and A. Suzuki, Small-size light-weight transformer with new core structure for contactless electric vehicle power transfer system, in Energy Conversion Congress and Exposition (ECCE), IEEE, Sept, pp. 6 66. [] K. van Schuylenbergh and R. 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