Designing A Medium-Power Resonant LLC Converter Using The NCP395 Prepared by: Roman Stuler This document describes the design procedure needed to implement a medium-power LLC resonant AC/DC converter using the NCP395 resonant mode controller. A 40W converter has been selected as an illustrative example. The specifications of the converter are as follows: - Input voltage range from 90 to 65VAC - 40W continuous output power at 4V output voltage - Total efficiency greater than 90% at full load and 30VAC input voltage - Low power consumption under no-load conditions - Built-in overcurrent protection The LLC series resonant converter topology using the NCP395 is a very attractive solution for this design because it offers these features: Brown out protection pin This pin continuously monitors the converter input voltage, ensuring that the converter will operate only in the desired input voltage range. This feature is very useful for the LLC topology, which is usually optimized to operate within a narrow range of input voltage. Fast fault input This input can be used as a shut down input in applications such as an LCD television SMPS. Because this input lacks a soft-start feature, it can also be used to implement skip cycle mode during light load conditions, thus significantly decreasing standby power consumption. Timer-based fault protection (slow fault input) When this input is asserted the NCP395 shuts down after a programmed delay. This pin integrates successive inputs, so a transient overload condition will not stop converter operation. Internal transconductance opamp This internal amplifier can be used to create effective overload protection circuitry. The power supply can then operate in either CV or CC mode, making it ideal for applications such as battery chargers. Common collector optocoupler connection With this circuit arrangement, additional feedback loops can be included and hence contribute to the output voltage regulation. Examples include an overcurrent sensing circuit, an overtemperature sensor, etc. Please refer to the NCP395A/B datasheet for detailed description of the above features, as well as information on additional functions. Demo board schematic description The schematic of the demo board is shown in Figure. The complete design contains three blocks: the PFC front end (required for the specified power level), the LLC converter stage, and an auxiliary buck converter to power the PFC and LLC controllers.
Input Filter Stage Capacitors C -C 5, together with inductors L and L and varistor VDR, form the input EMI filter that suppresses noise conducted to the mains. Diodes D4A & B and D5A & B rectify the line voltage, and capacitors C 6 and C 7 filter the HF ripple current generated by the PFC stage. PFC Stage The NCP653A Fixed-Frequency Continuous Conduction Mode PFC controller is used to control the PFC stage. The classical PFC boost topology includes inductor L 3, MOSFET switch M, SiC diode D, bulk capacitors E, E and inrush current bypass diode D. The current in this stage is monitored by current sense network R, R 8 //R 9. The output voltage of the PFC stage is regulated to 400VDC nominal via feedback loop components R -R 4 and C 9. A voltage divider formed by R 5 -R 7 and capacitors C 3 and C 4 generates a portion of the rectified input voltage to provide over-power protection for the PFC switch. Capacitor C filters the control voltage and sets the PFC feedback response time. The components on pin 5 of the NCP653A set the PFC stage to CCM and also dictate the over-power protection level. Please refer to application note AND884/D for a detailed explanation of how to design a PFC stage using the NCP653A controller. Auxiliary Buck Converter An NCP0ST065T3 monolithic offline switching regulator forms the heart of the auxiliary buck converter. This converter ensures proper operation of the other stages over the entire range of operating conditions, especially when a short-circuit is applied to the LLC stage output. The NCP0ST065T3 serves as a high side switch in this configuration. Diode D 3 is the freewheeling diode. The buck converter is supplied from the rectified mains via diode D 5 and electrolytic capacitor E 3. Feedback is via diode D 6, optocoupler IC 4 and capacitor C 8. The supply voltage of the NCP0 is maintained on capacitor E 4 using diode D 4, resistor R 3 and the IC s internal Dynamic Self Supply (DSS) architecture. The DSS block is inactive during steady state operation to reduce the power consumption. The output voltage of this converter is regulated to 6V, providing a V margin above the PFC controller max V ccon level of 5V. Please refer to application note AND89/D for further information on the NCP0xx family of products. LLC Stage As mentioned above, the NCP395 resonant mode SMPS controller is used to implement the LLC converter stage. The power stage of this converter is formed by bulk capacitors E, E, MOSFETs M, M 3, resonant inductor L 5, transformer TR and resonant capacitor C 3. A center-tapped winding is used on the secondary side to increase the converter efficiency. Electrolytic capacitors E 6 -E together with inductor L 6 form the output filter.
Regulation of the output is by means of IC 6, a TL43 programmable shunt regulator diode. R 46, R 4 and R 43 form a divider that sets up the desired output voltage, and R 44 supplies the TL43 biasing current. Frequency compensation of the TL43 is implemented by the series combination of capacitor C 6 and resistor R 4. R 40 limits the maximum current through optocoupler IC 5. The primary current of the LLC stage is sensed by current sense transformer TR together with diodes D 8 -D, resistors R 35, R 36 and capacitor C. If required, the current sense transformer can be replaced by an alternative sensing circuit formed by resistors R 38, R 39 diodes D 7, D 3 and capacitor C 4. This circuit is also included in the demo board layout. The minimum and maximum allowable operating frequencies of the NCP395 are set by resistors R 8 and R 9 respectively. The dead time between outputs A and B is set by the value of R 0. C 5 sets up the soft start duration, and the combination of C 6 and resistor R controls the fault timer duration. The bulk voltage is monitored for brownout detection by the voltage divider formed by R 4, R 5, R 6, R and capacitor C 8. level at which skip mode starts is set by the voltage divider R 6 and R 7. The Fast Fault input is filtered by capacitor C 0. Resistor R 30 sets the voltage gain of the operational transconductance amplifier (OTA), and capacitor C is used for the current feedback loop compensation. The OTA output voltage is limited to a maximum of 7.5V by resistor R 47 and zener diode D 6. This limit is necessary because a higher voltage could ignite the skip mode during start-up or under an overload condition. The output current level at which skip cycle mode occurs during overload conditions is set by voltage divider R 4 and R 5. A separate driver module, mounted vertically on the converter main board, drives the half bridge switches M and M 3. This arrangement allows the designer to test different driver topologies quickly and easily. Two different drive circuits utilizing either a transformer or a high-voltage driver are available. Their schematics are shown in figures and 3 respectively. The NCP58 integrated high voltage driver circuit is especially suitable for cost-sensitive consumer applications. C 9 provides decoupling between the ground and V cc pin of the controller. Regulation voltage is produced on the resistor R 33 by optocoupler IC 5. Resistor R 8 limits the current that can flow to zener diode D, which protects the feedback input pin. The output power 3
Figure.: MOSFET drive circuit with a transformer No increased component count: the component count is virtually the same as the classical half bridge topology. The disadvantages of the LLC converter are as follows: Figure 3.: MOSFET drive circuit with the NCP58integrated driver The design steps to determine the component values for the resonant tank circuit are described in detail below. LLC converter topology overview The LLC Resonant converter is an attractive alternative to the traditional Half Bridge (HB) topology for several reasons. Advantages include: ZVS capability over the entire load range: Switching takes place under conditions of zero drain voltage. Turn-on losses are thus nearly zero and EMI signature is improved compared to the HB, which operates under hardswitching conditions. Higher peak and RMS values of the primary and secondary currents compared to the classical HB topology. Consequently the LLC topology is not suitable for very high output current levels. Narrow input voltage operating range: The LLC converter has to be optimized for use over a narrow input voltage range in order to take full advantage of its benefits. Variable operating frequency: the operating frequency of the LLC converter has to vary in order to maintain output regulation, complicating the downstream filter design. LLC resonant converter operation description The topology of the LLC resonant converter power stage is shown in figure 4. Low turnoff current: Switches are turned off under low current conditions, and so the turn-off losses are also lowered compared to the HB topology Zero current turnoff of the secondary diodes: when the converter operates under full load, the output rectifiers are turned off under zero-current conditions, reducing the EMI signature. Figure 4.: Power stage of the LLC resonant converter There are three resonant components in this topology: 4
L R the resonant inductance L M the magnetizing inductance of the transformer C R - the resonant capacitor. We can define two different resonant frequencies using Thompson equations and : f r (eq.) π Lr Cr f r π ( Lr + Lm) Cr (eq.) n is the transformer turns ratio η is the expected efficiency Using the equivalent loading resistance R ac we can obtain the gain transfer characteristic of the LLC converter for a given value of load resistance and resonant tank components (i.e., for a particular Q factor of the resonant circuit). The fastest way to do this is by using SPICE simulation. The results of such a simulation can be seen in figure 6. This topology behaves like a frequency dependent voltage divider whose equivalent circuit is shown in figure 5. Figure 6.: Typical gain characteristic of the LLC resonant converter Figure 5.: Equivalent circuit of the LLC resonant converter The gain transfer function of the divider can be found using fundamental analysis [4], [5]. The main simplifying assumption of this analysis is that only the fundamental frequency is passed through the resonant tank. As a result of this simplification, the real loading resistance R L can be converted to an equivalent loading resistance R ac using equation 3. R ac 8 RL π n η (eq.3) Where: R L is the real loading resistance The load conditions will change in a real application so it is very useful to show in a single graph the performance under different load conditions. We can use parametric analysis see figure 7. Figure 7.: Gain characteristics for different load conditions Three operating areas can be identified from Figure 7: 5