Application Note, V1.1, Apr. 2002 CoolMOS TM AN-CoolMOS-08 Power Management & Supply Never stop thinking.
Revision History: 2002-04 V1.1 Previous Version: V1.0 Page Subjects (major changes since last revision) Document s layout has been changed: 2002-Sep. For questions on technology, delivery and prices please contact the Infineon Technologies Offices in Germany or the Infineon Technologies Companies and Representatives worldwide: see our webpage at http://www.infineon.com. CoolMOS TM, CoolSET TM are a trademarks of Infineon Technologies AG. We Listen to Your Comments Any information within this document that you feel is wrong, unclear or missing at all? Your feedback will help us to continuously improve the quality of this document. Please send your proposal (including a reference to this document) to: mcdocu.comments@infineon.com Edition 2002-04 Published by Infineon Technologies AG, St.-Martin-Strasse 53, 81669 München, Germany Infineon Technologies AG 2002. All Rights Reserved. Attention please! The information herein is given to describe certain components and shall not be considered as warranted characteristics. Terms of delivery and rights to technical change reserved. We hereby disclaim any and all warranties, including but not limited to warranties of non-infringement, regarding circuits, descriptions and charts stated herein. Infineon Technologies is an approved CECC manufacturer. Information For further information on technology, delivery terms and conditions and prices please contact your nearest Infineon Technologies Office in Germany or our Infineon Technologies Representatives worldwide. Warnings Due to technical requirements components may contain dangerous substances. For information on the types in question please contact your nearest Infineon Technologies Office. Infineon Technologies Components may only be used in life-support devices or systems with the express written approval of Infineon Technologies, if a failure of such components can reasonably be expected to cause the failure of that life-support device or system, or to affect the safety or effectiveness of that device or system. Life-support devices or systems are intended to be implanted in the human body, or to support and/or maintain and sustain and/or protect human life. If they fail, it is reasonable to assume that the health of the user or other persons may be endangered.
Table of Contents Page 1................................... 4 1.1 Flyback Converter............................................ 4 1.2 Boost Converter (PFC)........................................ 5 1.3 Single Transistor Forward Converter.............................. 7 1.4 Half Bridge Forward Converter.................................. 9 1.5 Two Transistor Forward Converter.............................. 11 1.6 Full "H" Bridge Converter...................................... 13 1.7 Full Bridge ZVT Converter..................................... 15 2 List of related application notes............................... 18 Application Note 3 V1.1, 2002-04
AN-CoolMOS-08 This application note gives a briefly overview of major state of the art SMPS topologies. 1 A variety of converter topologies are used in switch mode power supplies employing pulse width modulation to regulate an output voltage. Table 1 shows the basic topologies in widespread use. Note that though the classic description of these topologies specifies only hard PWM as a switching mode, there are resonant variations of many of these with similar characteristics. Also there are specialized converters such as the Cúk and SEPIC using multiple reactive components for energy transfer, but with operating characteristics for the power switch which are similar to the basic topologies described below, though often with increased V DS requirements. Table 1 SMPS Topologies & Transistor Selection Topology CoolMOS Generation Voltage Rating PFC Boost Converter S5, C2, C3 500 V, 600 V Flyback Converter S5, C2, C3 600 V, 800 V Forward Converter Single Transistor C3 800 V Half-Bridge Converter -Symmetrical PWM -Symmetrical Resonant -Asymmetrical (2 Transistor Forward) Center Tap Single Push-Pull Converter not recommended S5, C2, C3 S5, C2, C3 C3 500 V, 600 V 500 V, 600 V 500 V, 600 V 800 V Full Bridge PWM not recommended 500 V, 600 V Full Bridge ZVT-Phase Controlled C2, C3 500 V, 600 V Application Note 4 V1.1, 2002-04
1.1 Flyback Converter The Flyback converter is one of the simplest and most economical SMPS power supply topology, suited best to lower power levels, because the triangular current waveforms incur high peak losses in the primary side switch, and relatively high output ripple current and ripple voltage on the output side. The flyback transformer is designed as an energy storage and transfer inductor, sized to store the energy required at the peak of the primary current during the first switching state. This maximum energy storage is irrespective of the input line voltage; variations in line voltage merely change the duty cycle required to charge the flyback transformer to the programmed current level. The transformer turns ratio is selected based on the allowable reflected flyback voltage as well as the desired output voltage. During the second switching state, the power switch must block the bus voltage "+V In " plus the reflected reset voltage determined by the regulated V Out and the transformer turns ratio. Uncoupled inductance from the primary to secondary (leakage inductance) will also store energy, and since this energy is not clamped by the output winding, it will cause an avalanche on the primary unless clamped by an RCD snubber network. Figure 1 Flyback Converter Energy transfer occurs by charging a current into the flyback inductor/transformer primary by turning on the power transistor. When the transistor turns off, the inductor reset on the secondary side conducts through CR1 to the output capacitor. Leakage inductance on the primary must be clamped on primary side. Application Note 5 V1.1, 2002-04
1.2 Boost Converter (PFC) The boost converter as shown (Figure 2) is not an isolated output SMPS converter. It is used to raise an unregulated input voltage to a higher level, and is commonly employed in active Power Factor Correction circuits. The boost inductor is the primary energy storage and transfer element, storing energy when the switching transistor Q1 is turned on, and delivering it to an output capacitor through CR1 when the switching transistor turns off. If the high frequency power loop formed by Q1, CR1, and C Out is reasonably small, with low stray inductance, CR1 will clamp the drain voltage of Q1, and avalanche conduction is unlikely. If there is considerable stray inductance in this loop, then at high di/dt for Q1 turn off there may be a possibility of brief avalanche events. Figure 2 Boost Converter Energy transfer occurs by charging a current into the boost inductor by turning on the power transistor. When the transistor turns off, the inductor reset transfers energy to the output capacitor through the boost diode. Output voltage is higher than input, because the inductor reference is at the input voltage. Application Note 6 V1.1, 2002-04
1.3 Single Transistor Forward Converter The single transistor forward converter (Figure 3) offers some significant performance advantages over the flyback SMPS converter, but at the cost of many additional components. Instead of combining energy storage and voltage isolation/conversion in one magnetic component, a separate transformer and output filter inductor are used, permitting more favorable trapezoidal current waveforms and lower output current and voltage ripple, thus reducing noise and decreasing stress on semiconductors and capacitors. In a conventional single transistor forward converter, the transformer reset occurs after the power transfer cycle, and requires that the input transistor block a minimum of twice the input voltage. In practice, the coupling between the reset clamp winding and the primary power winding may not be ideal, and leakage inductance on the primary winding can store energy which can cause avalanche voltage overshoots unless clamped by an RCD snubber network across the power transistor Q1. For a maximum rectified bus voltage of 360 V, a V DS rating of 800 V is required for the power transistor Q1, unless alterations to the maximum duty cycle, and special clamp winding arrangements are made to lower the reflected voltage on Q1. Application Note 7 V1.1, 2002-04
Figure 3 Single Transistor Forward Converter Energy transfer occurs across the isolation transformer, when the power transistor Q1 turns on the primary voltage is reflected across the output windings, and rectified by CR1, charging the output inductor. When the primary switch turns off, the bifilar primary clamp winding conducts through the clamp diode, clamping the drain voltage of Q1 at twice the input voltage, and returning the energy from the magnetizing inductance of the transformer to the primary power bus (C In ). The driven side of the output inductor is clamped at 0.7 volts below ground by the recirculating diode, and the output inductor and output capacitor store energy and integrate the duty cycle so that the output voltage is proportional to the product of the rectified output voltage and duty cycle. If the primary winding is not bifilar (wound at the same time) with the clamp winding, there will be a substantial unclamped leakage inductance on the primary. This leakage inductance and any additional stray inductance stores energy which will not be clamped by clamp winding, and must be dissipated on the primary side by the power transistor in avalanche, or by additional protective snubber networks. Application Note 8 V1.1, 2002-04
1.4 Half Bridge Forward Converter This converter design offers the possibility of reducing the size of the transformer by nearly 1/2 compared with the single transistor forward converter, because it's single ended push-pull configuration uses the transformer flux in both directions. It doesn't require a clamp winding, but does require two output windings, to support both polarities of output drive from the transformer. By replacing the small flux balance cap with a resonant network, it is possible to easily make a resonant mode converter, with very low switching losses because the voltage turn-on and turn-off occurs at very low current. The body diodes of the switching transistors Q1 and Q2 provide clamping of turn-off transients due to leakage inductance, so avalanche is not normally an issue with this topology. Because of the primary side capacitors and their affect on the source voltage driving the transformer and output inductor, this topology cannot be used readily with current mode control, which is a significant disadvantage from the points of control loop dynamics, audio susceptibility, line regulation, and transistor protection. For this reason this topology has fallen in popularity for midrange sized power supplies, though it may commonly be found in direct coupled lighting applications when used with resonant components. Application Note 9 V1.1, 2002-04
Figure 4 Half Bridge Forward Converter Energy transfer occurs across the isolation transformer, in single ended push-pull. First, when the power transistor Q1 turns on the primary voltage is reflected across the output windings, and rectified by CR1, charging the output inductor. When Q1 turns off, the voltage drive across the transformer primary drops to zero, and energy stored in the leakage inductance and magnetizing inductance causes a turn-off overshoot, which is clamped by the body diode of Q2. In the second stage, Q2 turns on, and the transformer is driven in the opposite direction, resetting the flux balance in the transformer core. The output of the transformer is made with two windings and connected to a half wave rectifier, so the alternating polarity pulse train is rectified into a unidirectional pulse train of twice the frequency. The output inductor and output capacitor store energy and integrate the duty cycle so that the output voltage is proportional to the product of the rectified output voltage and duty cycle. Application Note 10 V1.1, 2002-04
1.5 Two Transistor Forward Converter This SMPS topology (Figure 5) has been widely used because of it's robustness, simplicity, and moderately high performance. It is similar in performance characteristics to the single transistor forward converter, excepting that the two-transistor topology is inherently self-clamping for the magnetizing current reset of the power transformer, making avalanche operation unlikely. Additionally, this topology requires power transistors with only 1/2 the V DS blocking capability of the single transistor version. This reduction in voltage requirements dramatically reduces the R DS[on] for silicon area in the case of conventional MOSFET transistors, with the result that the two smaller transistors usually cost less than the single larger transistor, with lower total losses. The two transistor forward converter is compatible with current mode control, and with the improved operating conditions for the transistor switches due to the lower operating voltage requirements, gives good performance in midrange power applications. It's main drawback compared with the Half Bridge converter is the necessity for a larger power transformer because the flux swing can only operate in one direction, but this also eliminates the necessity for any flux balancing methods in the control circuits. Application Note 11 V1.1, 2002-04
Figure 5 Two Transistor Forward Converter Application Note 12 V1.1, 2002-04
Energy transfer occurs across the isolation transformer, when the power transistors Q1 and Q2 turn on, and the primary voltage is reflected across the output windings, and rectified by CR1, charging the output inductor. When the primary switch turns off, the flyback from the leakage inductance and magnetizing inductance flows through the clamp diodes D1 and D2, clamping the flyback of the primary and returning the energy from the magnetizing inductance of the transformer to the primary power bus (C 1In ). The output inductor and output capacitor store energy and integrate the duty cycle so that the output voltage is proportional to the product of the rectified output voltage and duty cycle. 1.6 Full "H" Bridge Converter The "H" Bridge converter (Figure 6) draws its name from the four switching legs and their common connection to the load or output transformer. It combines some of the best features of the Two Transistor Forward converter and the Half Bridge converter. These include low input voltage requirements for the power switches, smaller transformer size from utilizing both polarities of BH loop excitation, inherent clamping of magnetizing current and leakage inductance transients, and compatibility with current mode control. The latter also provides inherent flux balancing of the power transformer, insuring equal volt seconds across the transformer primary in both directions. Because the H-Bridge topology is capable of operating at effective inductor PWM duty cycles greater than 50%, stable operation in current mode control avoiding sub harmonic oscillation requires slope compensation of the inner control loop, unless a rectifier output topology such as a current doubler is used, which cuts in half the effective duty cycle on the output inductors. The rectified output pulse train is at twice the switching frequency of the primary transistors, which may also allow a reduction in the size of the inductor magnetics for a given output ripple voltage requirement. Application Note 13 V1.1, 2002-04
Figure 6 Full Bridge Converter with Conventional PWM Application Note 14 V1.1, 2002-04
Energy transfer occurs across the isolation transformer in balanced push-pull. First, when the power transistors Q1 and Q4 are turned on, the full bus voltage is applied across the primary winding, and the primary voltage is reflected across the output windings, and rectified by CR1, charging the output inductor. When Q1 and Q4 turn off, the voltage drive across the transformer primary drops to zero, and energy stored in the leakage inductance and magnetizing inductance causes a turn-off overshoot, which is clamped by the body diodes of Q2 and Q3. In the second stage, Q2 and Q3 turn on, and the transformer is driven in the opposite direction, resetting the flux balance in the transformer core. The output of the transformer is made with two windings and connected to a half wave rectifier, so the alternating polarity pulse train is rectified into a unidirectional pulse train of twice the frequency. The output inductor and output capacitor store energy and integrate the duty cycle so that the output voltage is proportional to the product of the rectified output voltage and duty cycle. 1.7 Full Bridge ZVT Converter This topology (Figure 7) is similar in physical and electrical layout to the conventional H Bridge topology, but it's operation and control circuits are in some regards radically different, and significantly more complicated. Like the conventional H Bridge, advantages include small magnetics, low voltage requirements for the power transistors, reduced likelihood of avalanche, and compatibility with current mode control. The complexity comes into play with the control scheme necessary to achieve Zero Voltage Transitions and essentially eliminate switching losses in the power transistors. Instead of using the conventional H Bridge's simple anti-phase PWM modulation scheme where switches Q1 and Q4 are driven from the same control signal (Figure 6) and switches Q2 and Q3 are likewise driven, unique timing signals are generated for each switching transistor, so that each side of the H bridge switches roughly in a square wave fashion at drive points "A" and "B" (Figure 7), and the displacement in phase between the square wave drives produces an effective PWM across the transformer primary. The key advantage to this scheme lies in the fact that the drain to source transitions of each transistor are powered by the energy stored in the leakage inductance, or when necessary, a primary resonant inductor. Each transistor turns on with essentially zero volts drain to source over a wide load range. This requires careful timing of control signals, and adjustable delays between the turn off of one bridge transistor and the turn on the next. The benefit lies in eliminating almost all of the switching losses of the power MOSFET transistors, making possible increases in the operating frequency of the power supply, with attendant reductions in the size and weight of the power transformer and inductors. Application Note 15 V1.1, 2002-04
Figure 7 Full Bridge Converter with Phase Shifted ZVT Application Note 16 V1.1, 2002-04
List of related application notes Energy transfer occurs across the isolation transformer in balanced push-pull, as for the conventional Full Bridge. Key to the operation with no switching losses is the use of a phase shifted modulation scheme, using square waves on both sides of the transformer primary, and controlling the duty cycle across the transformer primary by changing the effective phase between the two square waves. To achieve near lossless switching operation, drain to source transitions occur when one transistor in a leg turns off, and the magnetizing or resonant inductance slews the drain to source voltage to the opposite potential, after which the other transistor is turned on. This requires the delays in switching control shown in Delay 1/2 and Delay 3/4, which shows in an exaggerated manner for clarity the timing delays required. In some implementations this delay is made variable as a function of load current, to optimize the drain to source resonant timing for both light loads and heavy loads. 2 List of related application notes Application Note 1: CoolMOS Selection Guide (AN-CoolMOS-02) Application Note 2: CoolMOS Design In Guidelines (AN-CoolMOS-03) Application Note 17 V1.1, 2002-04
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