TABLE 1 Normalized Differential Phase Shift Between Forward and Backward Wave at 3.5 GHz Normalized Differential Phase Shift Ž k 0. Ž deg. 0 90 10 6.6568e-004 4.6596e-004 20 1.3390e-003 1.3979e-003 30 2.5118e-003 1.5976e-003 45 7.3887e-003 0 60 2.3165e-002 9.5853e-003 75 5.9642e-002 4.6612e-002 90 4.4207e-001 8.7796e-002 served. This might be due to the propagation constants of the forward and backward waves being degenerate at this angular magnetization. The differential phase shift is considerably higher for the transverse magnetization compared to the normal and longitudinal magnetization cases. The characteristic impedance Ž Fig. 7. is observed to reduce with frequency, and the effective dielectric constant of the structure is observed to increase with frequency. At 0 and an operating frequency of 4 GHz, the characteristic impedance is observed to reduce as is varied from 0 to 90, while the characteristic impedance is observed to increase with for 3.5 GHz. The characteristic impedance is observed to increase with for 90 for both 3.5 and 4 GHz. For normal magnetization, the characteristic impedance is observed to be about 30. Hence, arbitrary orientation may assist in matching with the excitation source impedance, but at 4 GHz, the variation of impedance is not seen much. The change in differential phase shift with frequency finds many practical applications in microwave integrated circuits like phase shifters. 4. CONCLUSION A spectral-domain approach is employed in conjunction with matrix methods to estimate the electric and magnetic fields in each layer of the geometry in terms of an exponential matrix characteristic of the material parameters. The resulting integral equation is solved rigorously using the MoM to obtain the dispersion characteristics for both forward and backward waves under arbitrary orientation of dc magnetization. A different orientation of the magnetic field changes the phase constant, current distribution, and characteristic impedance significantly. It is possible to obtain the required phase shift in the microwave band by adjusting the magnetization angle. REFERENCES 1. M. Tsutsumi and T. Asahara, Microstrip lines using yttrium iron garnet film, IEEE Trans Microwave Theory Tech 38 Ž 1990., 1461 1467. 2. M. Tsutsumi and S. Tamura, Microstrip line filters using yttrium iron garnet film, IEEE Trans Microwave Theory Tech 40 Ž 1992., 400 402. 3. J.L. Tsalamengas, N.K. Uzunoglu, and N.G. Alexopoulos, Propagation characteristics of a microstrip line printed on a general anisotropic substrate, IEEE Trans Microwave Theory Tech MTT-33 Ž 1985., 941 945. 4. K. Okubo, A. Sanada, S. Takenawa, and K. Yamane, FDTD analysis of a microstrip line with YIG film magnetized in arbitrary direction, Asia Pacific Microwave Conf 2000, Sydney, Australia, pp. 357 360. 5. I.Y. Hsia, H.Y. Yang, and N.G. Alexopoulos, Basic properties of microstrip circuit elements on nonreciprocal substrate-superstrate structures, J Electromag Waves Appl 5 Ž 1991., 465 476. 6. D. Mirshekar-Syhakal, Spectral domain method for microwave integrated circuits, Wiley, New York, 1990. 7. T. Itoh and R. Mittra, Spectral-domain method for calculating the dispersion characteristics of microstrip lines, IEEE Trans Microwave Theory Tech MTT-21 Ž 1973., 496 499. 8. G. Tyras, The permeability matrix for a ferrite medium magnetized at an arbitrary direction and its eigen values, IRE Trans Microwave Theory Tech Ž 1959., 176 177. 9. E.L. Tan and S.Y. Tan, Concise spectral formalism in the electromagnetics of bianisotropic media, PIER, 2000, pp. 309 331. 10. G. Leon, R.R. Boix, and F. Medina, Efficient full-wave characterization of microstrip lines fabricated on magnetized ferrites with arbitrarily oriented bias field, J Electromag Waves Appl 15 223 252. 11. M. Tsutsumi, T. Ueda, and K. Okubo, Magnetostatic-wave envelope soliton in microstrip line using YIG-film substrate, IEEE Trans Microwave Theory Tech 48 Ž 2000., 239 402. 2001 John Wiley & Sons, Inc. A BROADBAND MICROSTRIP ANTENNA ARRAY FOR LMDS APPLICATIONS Soonsoo Oh, 1 Seongho Seo, 1 Mikyoung Yoon, 1 Changyoul Oh, 2 Eungbae Kim, 2 and Youngsik Kim 1 1 Department of Radio Sciences and Engineering Korea University Seoul 136-701, Korea 2 Radio and Broadcasting Technology Laboratory Electronics and Telecommunications Research Institute Taejon 305-350, Korea Recei ed 25 June 2001 ABSTRACT: In this paper, a broadband 20 20 microstrip antenna array is presented. Once a rectangular microstrip patch antenna with parasitic patches as an element antenna has been optimized at 24.2 26.7 GHz, a parallel-series feed network with minimum feed lengths can be utilized to combine element antennas. From simulation and measurement, the array antenna yields relati ely high gains of 26.0 30 db and narrow 3 db beamwidths of 4.5 5.1 in the H-plane and 4.1 7.1 in the E-plane, respecti ely, in an LMDS band. 2001 John Wiley & Sons, Inc. Microwave Opt Technol Lett 32: 35 37, 2002. Key words: antennas; microstrip antenna; microstrip antenna arrays DOI 10.1002 mop.10084 1. INTRODUCTION Since a microstrip patch antenna was introduced in 1972 1, patch antennas have been investigated widely because of their low profile, low weight, lower cost, and easy manufacturing 2, 3. To broaden the bandwidth of a resonant-type patch antenna, various shapes of microstrip patch antennas have been proposed, such as patch antennas with parasitic elements 4, annular ring antennas 5, and multilayered patches 6. For LMDS applications, a patch antenna with parasitic elements is one of the best candidates among those antennas because of its simple design and implementation for array element antennas. Although a hybrid Ž parallel-series. feed network is well suited for a planar array antenna, each series-fed element antenna phase is not the same. This may be a result of frequency scanning in the LMDS band of 24.2 26.7 GHz 7. To avoid scanning the spacing of the series-fed element, the antennas have been adjusted. This series feed network has been designed with the Dolph Chebyshev array method to reduce sidelobes 8. MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 32, No. 1, January 5 2002 35
Figure 1 Patch antenna element with parasitic patches Figure 3 Top view of 20 20 antenna array Figure 2 Return loss S of antenna element 11 2. ELEMENT ANTENNA Two additional resonators Ž parasitic patches., which are modified in 8, are placed adjacent to the nonradiating edges of a rectangular patch antenna, as shown in Figure 1. The patch antenna in 8 is not adequate for an element antenna because the dimensions of the E- H-plane are larger than 1 0. It was fabricated on a substrate with h 0.508 mm Ž 20 mil., r 2.2, and tan 0.0009. The patch dimensions are L1 L3 3.6 mm, L2 3.7 mm, and S1 S2 0.2 mm. After determining the width of W 2, those of W1 and W3 have been adjusted to obtain a proper bandwidth using IE3D 9 simulation, and the optimized values of W1 and W3 are 1 and 2 mm, respectively. An exponential taper is used to match the input impedance of a single antenna to the 50 transmission line. The simulated and measured return losses are given in Figure 2. Both curves have similar behaviors of two deeps. The return loss is below 10 db in a range of 24.5 29 GHz for both. A bandwidth of 12% for VSWR 2 is obtained from the simulation, and one of 15% from the measurement. Even though its size is smaller than the structure in 9, the bandwidth is as broad as 15.4% of that in 9. Figure 4 Return loss S of 20 1 and 20 20 antenna arrays 11 y-direction. The main feed is placed at the center of the array, and this feed results in a 24 mm spacing at the center vertical. Figure 4 shows the return loss of the 20 1 array antenna and the 20 20 array antenna. The principal E- H-plane radiation patterns of the antenna at 25.5 GHz are shown in Figures 5 and 6, respectively. The simulated patterns are similar to the measured ones. While the desired SLL is 30 3. ARRAY ANTENNA The Dolph Chebyshev method is used to achieve low sidelobes 8. For a 20-element array with a 30 db sidelobe level Ž SLL., the current distribution from the center patch to the end element is given as 1.00 : 0.97 : 0.91 : 0.83 : 0.73 : 0.62 : 0.50 : 0.39 : 0.29 : 0.33. This current distribution is realized with different impedances of the series feed network. To verify the current distribution of the feed network, the 20 1 linear array antenna was simulated and measured. The 20 20 planar array antenna is depicted in Figure 3. The element spacing is 8.5 mm in the x-direction and 8.35 mm in the Figure 5 25.5 GHz E-plane radiation patterns of 20 20 antenna array at 36 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 32, No. 1, January 5 2002
6. S.A. Long and M.D. Walton, A dual-frequency, stacked circular disc antenna, IEEE Antennas Propagat Soc Int Symp, 1978 Dig, pp. 260 263. 7. F. Lalezari and C.D. Massey, mm-wave microstrip antennas, Microwave J 4 Ž 1987., 87 96. 8. W.L. Stuzman and G.A. Thiele, Antenna theory and design, Wiley, New York, 1998. 9. G. Kumar and K.C. Gupta, Nonradiating edges and four edges gap-coupled multiple resonator broad-band microstrip antenna, IEEE Trans Antennas Propagat AP-33 Ž 1985., 173 178. 10. Zeland Software, Inc., IE3D 6.0, New York, NY, 1999. Figure 6 25.5 GHz TABLE 1 H-plane radiation patterns of 20 20 antenna array at Gain and 3 db Bandwidth of 20 20 Antenna Array 3 db Beamwidth deg Frequency GHz Gain dbi H-Plane E-Plane 24.25 26.0 4.8 5.8 24.50 26.4 4.9 6.2 24.75 26.8 5.0 7.1 25.00 27.6 5.1 5.2 25.50 30.0 4.6 4.1 26.00 28.3 4.6 4.6 26.50 27.8 4.6 4.4 26.70 30.7 4.5 4.3 db, the measured and simulated ones are 21 db in the E-plane and 15 db in the H-plane, respectively, due to an imperfect power distribution and element spacing of 24 mm along the vertical center feed line. The gain is 30 db, and the 3 db beamwidth is 4.6 and 4.1 in the E- H-plane, respectively. The gain and beamwidth of the 20 20 array antenna are summarized in Table 1. 4. CONCLUSION A microstrip single patch antenna with parasitic patches was developed, and yielded a bandwidth of 15%. A parallel-series feed network with minimum feed lengths was utilized to combine element antennas. A series feed network was designed from the Dolph Chebyshev method for a 30 db sidelobe level, while the simulated and measured SLL shows 15 db due to the feed network. The array antenna provides a gain of 26.0 30.7 db, and the 3 db beamwidth is 4.5 5.1 in the H-plane and 4.1 7.1 in the E-plane, respectively, at 24.2 26.7 GHz for LMDS applications. REFERENCES 1. J.Q. Howell, Microstrip antennas, IEEE Antennas Propagat Soc Int Symp, 1972 Dig, pp. 177 180. 2. J.R. James and P.S. Hall, Handbook of microstrip antennas, IEE Electromag Wave Ser 28, Peter Peregrinus, London, England, 1989. 3. D.M. Pozar, Microstrip antennas, Proc IEEE 80 Ž 1992., 79 81. 4. H. Pues, J. Vandensande, and A.V. Capelle, Broadband microstrip resonator antennas, IEEE Antennas Propagat Soc Int Symp, 1978 Dig, pp. 68 69. 5. W.F. Richards, J.D. Qu, and S.A. Long, A theoretical and experimental investigation of annular, annular sector and circular sector microstrip antenna, IEEE Trans Antennas Propagat AP-32 Ž 1984., 864 867. 2001 John Wiley & Sons, Inc. IMPROVED TUNING ACCURACY OF FIBER GRATING LASERS USING A LINEAR VARIABLE DIFFERENTIAL TRANSFORMER W. H. Chung, 1 H. Y. Tam, 1 M. S. Demokan, 1 and P. K. A. Wai 2 1 Department of Electrical Engineering Hong Kong Polytechnic University Hung Hom, Kowloon, Hong Kong SAR, P.R. China 2 Department of Electronic and Information Engineering Hong Kong Polytechnic University Hung Hom, Kowloon, Hong Kong SAR, P.R. China Recei ed 2 July 2001 ABSTRACT: A microprocessor-controlled feedback system using a linear ariable differential transformer to measure the compression of a fiber Bragg grating with a iew to impro e the tuning accuracy of a fiber grating laser is reported. This technique o ercomes the large hysteresis of the PZT actuator normally used to compress the grating. A tuning range of about 20 nm with a readout wa elength accuracy of better than 0.05 nm was achie ed. 2001 John Wiley & Sons, Inc. Microwave Opt Technol Lett 32: 37 40, 2002. Key words: fiber lasers; wa elength tuning; fiber Bragg gratings; WDM applications DOI 10.1002 mop.10085 INTRODUCTION Fiber lasers based on fiber Bragg gratings Ž FBGs. are promising candidates for dense wavelength-division multiplexing Ž DWDM. systems because of the ease of producing an FBG with a highly accurate and repeatable Bragg wavelength that matches the ITU wavelength grids. Recently, there has been intense interest in employing tunable sources in DWDM systems to use as spare components in order to reduce cost by stocking fewer spare lasers. Tunable fiber grating lasers offer an important advantage in this respect because an FBG can be tuned over a large wavelength range in comparison with a semiconductor laser which normally can be tuned over just 1 2 nm. FBG tuning is generally achieved by using either thermal or mechanical methods 1 5. The direct thermal tuning technique provides a limited tuning range because of the small wavelength temperature coefficient of an FBG Žthe typical value is 0.012 nm C., and it is also a very slow tuning technique. Mechanically stretching or compressing an FBG enables a wider tuning range of more than 44 nm 5. Contract grant sponsor: Research Grants Council of the Hong Kong Special Administrative Region, China Contract grant number: Project PolyU 5142 97E MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 32, No. 1, January 5 2002 37
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