Balanced antipodal Vivaldi antenna for wide bandwidth phased arrays

Balanced antipodal Vivaldi antenna for wide bandwidth phased arrays J.D.S. Langley P.S. Hall P. Newham ndexing terms: Vivald antenna, Phased arrays, Wide bandlimited arrays, Stripline antenna Abstract: The Vivaldi antenna, a form of tapered slot radiator, has been shown to produce good performance over a wide bandwidth, limited only by the traditionally used slotline to microstrip feed transition. The authors present a new antenna, the balanced antipodal Vivaldi, which incorporates an ultrawide bandwidth transition and overcomes the poor polarisation performance of the antipodal form. Good performance over a 1 to 40 frequency range has been obtained. The use of the antenna in a linear phased array has also been investigated using elements constructed on high permittivity substrate. Wideband wide angle scanning with good cross-polarisation levels is obtained. 1 ntroduction Multi-octave performance phased arrays are important for a number of applications, including dectronic warfare and multiple mode radar systems. Wide bandwidth array action is obtained primarily through the use of wide bandwidth array elements, although arrays incorporating clusters of elements covering sections of the desired bandwidth have been reported [l]. Such elements should have, in addition to wide bandwidth, symmetrical beamwidths to optimise scanning and should be compact to allow sufficiently small element spacing to prevent grating lobe formation at the maximum operating frequency. An additional requirement is that the element should allow integration with transmitheceive modules constructed using a printed circuit transmission medium such as microstrip. There are several ways of creating a wide bandwidth array element. The ridged horn [2] exhibits bandwidths of up to two octaves with a highly symmetric beam and good power handling capability. The spiral antenna [l] has bandwidths in excess of four octaves but requires a wide bandwidth balun. Log periodic antennas [3] have 0 EE, 1996 TEE Proceedings online no. 19960260. Paper first received 24th August 1995 and m revised form 18th December 1995 J.D.S. Langley and P. Newham are with GEC Marconi Defence Systems Ltd., The Grove, Warren Lane, Stanmore, Mid& HA7 4LY, UK P.S. Hall is with the Univensty of Birmingham, Edgbaston, Bkmhgham B15 27T, UK * Formerly wlth the University of Birmingham, UK EE Proc -Mmow Antennas Propag, Val 143, No 2, April 1996 been used in HF or VHF skywave radar to give wide angle scanning. However, none of these elements are of a suitable form for circuit integration. The tapered slot antenna, however, can be fabricated using printed circuit techniques and is thus ideal for circuit integration. The slot antenna can be fabricated in either triplate stripline or microstrip. The stripline version, known as the tapered notch [4], is generally fabricated with an exponential taper. All other types of tapered slot are fabricated on microstrip and incude the Vivaldi, with an exponential taper [5], the linear taper, broken linear taper and constant width slot antennas [6]. All these antennas exhibit low cross-polarisation characteristics in the principle planes, however in the diagonal plane the CO- to cross-polarisation ratio decreases rapidly away from boresight [6]. This group of antennas is now widely used not only in phased arrays but also in radio astronomy, remote sensing, multiple beam satellite communications and spatial power combining techniques. n this study both the tapered stripline notch [4] and the Vivaldi [5] antennas have been tested using identical elliptical tapers, these antennas being fed by stripline and microstrip respectively. Nearly identical performance is noted in our studies, in terms of gain, beamwidths and cross-polarisation, while references [4] and [5] suggest differences in operation. However one difference which is pertinent to phased array operation is that the Vivaldi antenna has an open feed line which can radiate and perturb the radiation pattern. Although both these elements can have equal beamwidths and can in principle be directly connected to an integrated circuit, the slotline to feedline transition limits the bandwidth and requires considerable ingenuity to give broadband performance. This paper describes the development and performance of a new tapered slot antenna element (first described in [7]) that overcomes the transition problem to produce an ultra-wideband element for circuit integration. Performance in a small linear phased array is also presented and performance discussed. 2 Batanced antipodal Vivaldi antenna f the feed transition is made a collinear extension of the slot, then the bandlimiting effect is removed giving very wide bandwidth operation. n the antipodal Vivaldi [S, 91 shown in Fig. 1, a smooth transition between twin line and microstrip is used. The metallisation on either side of the substrate is flared in opposite directions to form the tapered slot. Fig. 2 shows the 97

~ Vivaldi ~ balanced ~ gain input return loss of the antipodal antenna compared to a Vivaldi of similar taper characteristics and substrate material. t is clearly seen that band limitation caused by the Vivaldi transition is removed and wideband action is indeed obtained. The lower frequency limit is now determined by the cut-off mechanism of the flare, namely that at the lowest operating frequency the aperture is half a wavelength wide. However the antipodal nature of the antenna gives rise to very high levels of cross-polarisation particularly at high frequencies as Fig. 3 shows, due to the skew in the slot fields close to the throat of the flare. groundplane twin line dielectric parallel to the metallisation whilst the output transmission medium is triplate stripline. Elliptical radiating tapers were chosen for this antenna because previous work with the Vivaldi and tapered notch antennas showed that this particular taper gave similar E and H beamwidths. The notch length-to-width ratio was set at 2:l to give gain of between 5 and lodbi as found in our previous Vivaldi studies. balanced qroundplanes metalisation flared %-- slot dielectrics i i microstrip Fig. 1 Diagrammatic view of antpodal Vivaldi antenna 0-5 -1 0-1 5 m, -20-25 -3 0-35 E field -40 2 4 6 8 10 12 lir 16 18 20 frequency,ghz Fig.2 nput return loss antipodal Vivaldi (Physical details of antennas; substrate thickness = 1.58mm, E, = 2.32, width = 40mm; flare length = 30mm, width at aperture = 15m, shape elliptical; triplate stripline width = 3mm; microstrip line width 5mm, transition flare length = 30mni, spacing between flares = 20mm, flare shape elliptical, major to minor axis ratio = 3.33). stripline resultant E field Fig. 4 Diagrmmtic view of balanced antipodal Vivaldi antenna -5-1 0-1 5 m, -20-25 -30 V -40-2 436 8 10 512 14 16 L18 20 1 Fig.5 nput return loss Balanced antipodal Vivaldi: ~ ~ theory _ (Details as Fig. 2) frequency,ghz measured 120, A l2 100 + / '10 k/\ 80, ', 8 -- / -35 i 6 7 8 9 b 1; 12 13 lc 15 l'6 i7 lb frequency,ghz Fig. 3 Cross-polarisation antipodal Vivaldi...... antipodal Vivaldi Vivaldi (Details as Fig. 2) To overcome this high cross-polarisation we have added a further layer of metallisation to form a balanced antipodal Vivaldi as shown in Fig. 4. The resultant electric field in the slot region is now oriented 98 o v,,,, T,,, 0 6 7 8 9 10 11 12 13 14 15 16 17 18 frequency, GHz Fig.6 Measured beamwidths and gain of balanced mtpodal Vivaldi antenna _ E-plane ~ H-plane (Details as Fig 2) The input return loss, Fig. 5 is similar to that obtained for the antipodal Vivaldi while the crosspolarisation, Fig. 3, is improved and is typically below -20dB for this particular E~ = 2.32 substrate antenna. Fig. 6 shows that E and H beamwidths are approximately equal and constant over a 6 to l8ghz bandwidth. Fig. 16 also shows that the gain varies from 5 to 1ldBi. Radiation patterns are shown in Figs. 7 and 8. EE Proc -Microw Antennas Propag, Vol 143, No 2, April 1996

~ co-polarisation ~ co-polarisation ~ ------ Although Fig. 2 only extends to 20GHz, for comparison, a larger antenna has been shown to operate from 1 to 40GHz [lo]. -50 1-90 -60-30 0 30 60 90 azimuth angle,deg Fig. 7 Radiation patterns at loghz for balanced antipodal Vivaldi antenna: E-plane (measured) ---- cross-polarisation (measured)..... E-plane co-polarisation (measured) with asymmetric flares (computed) _ cross-polarisation (computed) (Details as Fig. 2; asymmetric flare antenna shown inset, major to minor axis ratio = 5.711-50 1 0 30 60 90 120 150 180 elevation angle, deg Fig.8 Rudmtzon pattern us in Fig. 7. H-plane t can be seen from Fig. 7 that there is a squint of about 15" in the E-plane radiation pattern. This squint appears to be independent of both frequency and permittivity. t is believed that this squint is due primarily to the unequal propagation velocity experienced by the currents on each side of the slot due to their different geometries. Measurements of the antenna aperture fields confirmed the presence of phase asymmetries together with some amplitude asymmetry. Various methods were tried to reduce the squint including cutting away some of the substrate, reducing the distance between the radiating flares and the transition flares, the addition of balancing flares and shorting pins between the two outer flares. However none of these reduced the squint but the use of asymmetric flares allowed the introduction of some asymmetry into the patterns which was found to offset the squint as shown in Fig. 7. The inset shows the shape of the flares used in this example. The tapers in the transition are of a similar elliptical form with length and spacing between tapers chosen to be greater than half a wavelength at the lower operating frequency. We have not performed optimisation on these transitions and it may well be that shorter antennas could be developed. EE Proc.-Microw. Antennas Propag., Vol. 143, No. 2, April 1996 3 Analysis The balanced antipodal Vivaldi antenna has been analysed using the finite difference time domain method. Thiele [l 11 has recently shown the utility of this method for the Vivaldi antenna and we have used a similar technique. The balanced antipodal antenna is modelled in a 79 by 215 by 29 cell volume with h/20 cell size at the highest frequency. The spacing between metallisation layers is modelled by two cells and the excitation is applied to the triplate stripline end by equal electric fields within the 4 by 5 cells representing the geometrical area of the stripline. Both Gaussian pulse and ramped sinusoidal excitation have been used. Far fields are obtained by a frequency domain near to far field transform based on a measurement volume surrounding the antenna spaced away from the structure by 3 cells. Taper curvature is represented by a staircase approximation. Computed results for input return loss and radiation patterns are shown in Figs. 5, 7 and 8 respectively. The input return loss calculation predicts both the low frequency cut-off associated with maximum aperture width and the fast ripple due to slight reflections from the dielectric to air interface. While the calculated radiation patterns for this antenna show good agreement with measured results, both for coand cross-polarisation. Surface currents on the metallisation layers can also be computed using this method. As previously noted [ 121 the current is primarily confined to the metallisation edges within the radiating flares and to the region of the strip in the transitions. Only a small amount of current exists on the flares of the transitions and this suggests that there will be only small perturbations to the radiation patterns from unwanted radiation in these areas. The other notable feature of this result is the large current standing wave on the structure close to the radiating region on the flare. microstrip to stripline transition 1,screened monolithic circuits/j:labsorbing Wall Y------.~. ---- Fig.9 ntegrated antenna - circuit module concept - including the microstrip to stripline transition (O] 4 Stripline to microstrip transition To enable direct integration with a microwave integrated circuit a stripline to microstrip transition was designed so that the complete antenna-circuit module could take the form shown in Fig. 9. Although such a module has not been made within the scope of this study a double-sided transition has been constructed and tested [lo]. The bottom ground plane is continuous whilst the upper one is electrically connected to the lower using via holes. Both the top ground plane and substrate are elliptically tapered while the strip width is tapered in the same fashion to ensure 50Q impedance in each medium. The transition was built on E, = 10.5 substrate for integration with the antennas made on the same substrate described in the next Section. The loss 99

~ 0" ~ 0" for two transitions is less than 2dB with a -12dB return loss across the desired 6-18GHz band (for a single transition) [lo]. 5 Arrays of balanced antipodal Vivaldi antennas Although the balanced antipodal Vivaldi antenna is intended to be used in a dual polarised array, only its performance in E-plane arrays has been demonstrated. To incorporate this wideband element into a scanning array, the elements must be placed at &2 at the highest frequency, where ho is the free space wavelength. Thus at the bottom end of a 3:l band these elements will be spaced by h016. t is well known that tapered slot antennas exhibit a low frequency cut-off which occurs when the maximum flare width at the aperture is hj2, where hs is the wavelength in the slot. Although this may not occur in very large phased arrays due to high mutual coupling at the lowest frequency, it will occur in the small arrays considered here and the following design method is therefore appropriate. Assuming an effective dielectric constant of cs in the slot these two conditions are met when A0 - As - A0 6 2 &2 (1) giving E, = 9. The substrate dielectric constant to achieve this could be approximately derived from uniform slot theory but its value is constrained both by the materials available and the likely materials to be used in the microwave integrated circuit. The value chosen for the initial demonstrator was 10.5, with an additional array being made on E, = 6 material while using separation of alternative elements in the H-plane to prevent grating lobes. The silhouette of the E, = 10.5 array is shown in Figs. 10 and 11 with that for the E~ = 6 array being similar. The seven element array was produced using one manufacturing process, to work over a 3 to 9GHz range. nitial results for a single element using a network analyser in both frequency and time domain mode revealed a substantial reflection from the dielectric edge at the flare aperture. Shaping of a dielectric extension beyond this aperture was found to reduce this reflection, with a semicircular extension as indicated in Figs. 10 and 11, giving optimum performance. This was then used on all array elements, for both the 10.5 and 6 permittivity substrates. The beamwidths for the E, = 10.5 elements are in general very large and therefore the E-plane asymmetry noted in the low permittivity elements is not observed. This is due to surface wave interaction and the fact that the radiation occurs from the front of the dielectric extension. n the E, = 6 case the trapped waves are reduced and some element asymmetry is observed in the E-plane. The principle plane crosspolarisation levels in these higher permittivity elements are similar to those found in the low permittivity versions (5-20dB). However the cross-polarisation in the diagonal planes is generally better in the higher dielectric constant medium. This is due to the fact that the physically smaller elements have shorter longitudinal current paths and therefore an increase in effective cancellation occurs. 0-10 db -20-30 -40-90 -70-50 -30-10 10 30 50 70 90 azimuth angle, deg Fig. 12 Measured scan patterns for the array shown in Figs 10 and 11, 3GHz scan ~ 20" _ scan 40" scan Fig. 10 Silhouette of 7-element E-plane array, outer conductor outline of dielectric showing semi-circular extensions (array details, substrate thickness = 0.64mm, E, = 10.5; radiating flare length = 24mm, width at aperture = 17mm, shape elliptical; stripline width = 0.2mm; transition flare length = 14mm, spacing between flares = 4mm, flare shape elliptical, major to minor axis ratio = 3.33) -90-70 -50-30 -10 10 30 50 70 90 azimuth angie,deg Fig. 13 Measured scan patternsfor the array shown in Figs 10 and 11, 6GHz scan 20' ~ scan ~ 40" scan 6 Array performance... *... Fig. 11 Silhouette of array as in Fig. 10, inner conductor Measured radiation patterns for the 7-element E-plane linear array with E, = 10.5 elements are shown in Figs. 12-14. Well formed beams are obtained in general. However some gain reduction is noted for the at all frequencies due to the element beamwidth of approximately 100". Fig. 15 shows the measured gain 100 EE Proc.-Microw. Antennas Propag., Vol. 143, No. 2, April 1996

~ 5 at the scan angles presented in Figs. 12-14, while Fig. 16 shows the cross-polarisation levels which are below -15dB across the 3 to 9GHz bandwidth. 0-10 db-20-30 :i -40-90 -70-50 -30-10 10 30 50 70 90 azimuth angle,deg Fi.4 Measured scan patterns for the array shown in Figs 10 and 11, 98Hz 0" scan - 20" scan - - 40" scan dbi O 3 4 5 6 7 8 b 9 frequency, GHz Fig.15 Meamredgain for 7-element E-plane array (e, = 10.5) 0" scan - 20" scan - 40" scan - 0,- -10 L- d -1 5 db -20-2 5 3 4 5 6 7 8 9 frequency.ghz Fig. 16 Cross-polarisation for 7-element E-plane array (E, = 10.5) 0" scan -_-- 20" scan - 40" scan Measured inter-element coupling for the E, = 6.0 (triangular lattice) E-plane array is below -20dB for most of the 3 to 9GHz band, with levels of -13dB below 3.5GHz. This is an improvement in coupling of EE Proc-Microw. Antennas Propag., Vol. 143, No. 2, April 1996 1 4dB across the band when compared with the E, = 10.5 array elements. Both the larger element spacing and the reduced amount of surface wave propagation in the E, = 6.0 array help contribute to this improvement. However, because the E-plane element asymmetries are more evident in the E, = 6.0 array, the scan patterns are not as well formed as in the E, = 10.5 array. Similar gain is obtained from both these arrays while the cross-polarisation levels from the E, = 6.0 array (-20dB across the 3 to 9GHz bandwidth) are in general 5dB better than for the E, = 10.5 array. Wideband, wide angle scanning has been achieved using these antennas. However these design techniques are perhaps only applicable to small phased arrays, as used in ECM, ECCM and DF systems. n large phased arrays (number of elements > 100) it is likely that the use of dielectric materials would not be necessary, as the high levels of mutual coupling experienced in such an array would enable operation with elements working well below their individual cut-off frequency [ 131. 7 Conclusions The limitations on the bandwidth of the Vivaldi antenna due to the slotline to microstrip transition have been overcome whilst preserving low cross-polarisation by the development of the balanced antipodal Vivaldi antenna. The new antenna allows simple integration with microwave integrated circuit transmit/ receive modules using an additional stripline to microstrip transition which on E, = 10.5 substrate has been shown to have a loss of less than db. An antenna on E, = 2.32 substrate has been shown to have a bandwidth in excess of 40:1, whilst over a 3:1 bandwidth, cross-polarisation below -20dB is obtained. Radiation patterns are in general well controlled but an E-plane squint is noted which can in principle be compensated for using asymmetrical flares. Performance on E, = 10.5 substrate and to some extent on E, = 6 is affected by the dielectric-air interface at the flare aperture and this mismatch has been reduced with the introduction of a semicircular substrate extension. Two 7-element E-plane arrays of these balanced antipodal Vivaldi elements have been constructed, one on E, = 10.5 and the other on E, = 6.0 with a triangu- 1 lar lattice structure to avoid the formation of grating lobes at the high frequency end of the band. Wideband, wide angle scanning has been achieved with these arrays while maintaining suitable cross-polarisation levels. 8 Acknowledgments This work was supported under an EPSRC CASE studentship by GEC Marconi Defence Systems Ltd, UK and the authors acknowledge the company's permission to publish. References SHVELY, D.G., and STUTZMAN, W.L.: 'Wideband arrays with variable element sizes', EE. Proc. Microwaves, Antennas and Propagation, 1990, 187, (4), pp. 238-240 MONSER, G.J.: 'Design considerations for broadband phasedarray elements beyond two octaves', nt. Con$ Military Microwaves, June 1986, pp. 392-396 101

3 TTTENSOR, P.J., and ORTON, R.S.: Calibration of a multioctave phased array, EE Znt. Con$ Antennas and Propagation, CAP 91, pp. 790-793 4 LEWS, L.R., FASSET, M., and HUNT, J.: A broadband stripline array element, EEE Syrnp. Antennas and Propagation, Atlanta, USA, 1974, pp. 335-337 5 GBSON, P.J.: The Vivaldi Aerial, 9th European Microwave Conference, Brighton, UK, September 1979, pp. 101-105 6 SCHAUBERT, D.H.: Endfire slotline antennas, rnt. conf 8 JNA 90, 13-15 November 1990, pp. 253-264 LANGLEY$ J.D.S., HALL, p.s., and NEWHAM, p.: ultrawide-bandwidth Vivaldi antenna with low crosspolarisation, Electron. Lett., 1993, 29, (23), pp. 2004-2005 1989,37, (12), pp. 1523-1528 GAZT, E.: mproved design of the Vivaldi antenna, EE Proc. Microwaves, Antennas and Propagation, 1988, 135, (2), pp. 89-92 9 FOURKS, N., LOUTAS, N., and SHULEY, N.V.: Parametric study of CO and cross polarisation characteristics of tapered planar and antipodal slotline antennas, ZEE Proc. Micicrowaves, Antennas and Propagation, 1993, 140, (l), pp. 11-22 10 LANGLEY, J.D.S., HALL, P.s.3 and NEWHAM, p.1 Balanced antipodal Vivaldi antenna for multi-octave bandwidth phased arrays, nt. Con$ JNA 94, 8-10 November 1994, pp. 585-588 THELE, E.T., and~taflove, A.: FD-TD Of Vivaldi flared horn antennas and arrays, EEE Trans. Antennas and Propagation, 42, (S), pp. 633-641 12 JANASWAMY, R.: An accurate moment method model for the tapered slot antenna, EEE Trans. Antennas and Propagation, 13 : Phased array workshop discussions, EEE nt. Conf. AP-S, June 1995 102 EE Proc.-Microw. Antennas Propag., Vol. 143, No. 2, April 1996

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