IN THE high power isolated dc/dc applications, full bridge

354 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 21, NO. 2, MARCH 2006 A Novel Zero-Current-Transition Full Bridge DC/DC Converter Junming Zhang, Xiaogao Xie, Xinke Wu, Guoliang Wu, and Zhaoming Qian, Senior Member, IEEE Abstract This paper proposes a novel zero current-transition pulse-width modulation full-bridge dc/dc converter. The proposed converter not only achieves zero current switching for the main switches and auxiliary switch in the entire load ranges, but it also realizes soft commutation for the output rectifier diodes. Furthermore, the auxiliary circuit also helps to turn on the main switches softly. Simulation results and experimental results verify the theoretical analysis. Index Terms Pulse-width modulation (PWM), zero current switching (ZCS), zero current-transition (ZCT), zero voltage switching (ZVS). Fig. 1. Topology of the proposed ZCT FB dc/dc converter. I. INTRODUCTION IN THE high power isolated dc/dc applications, full bridge (FB) type pulse-width modulation (PWM) dc/dc converters are adopted widely. A variety of zero voltage switching (ZVS) full bridge dc/dc converters have been proposed. The phase-shifted full-bridge (PSFB) ZVS converter is the most widely used ZVS topology in practical applications due to its simplicity and good performance. However, in high power level applications (such as higher than 5 kw), insulated gate bipolar transistor (IGBT) devices are usually preferred, due to the fact that the ZVS technique is suitable for metal oxide semiconductor field effect transistor (MOSFET) devices and zero current switching (ZCS) technique is more effective for IGBTs due to its tail current. Generally, there are two types of ZCS techniques, i.e., the resonant type and the PWM type. The resonant FB ZCS converter suffers from large reverse-recovery current of the body diode, which makes it unsuitable for practical applications [1], [7]. The PWM type ZCS converter can be further divided into two categories, i.e., the converter with an auxiliary circuit and without an auxiliary circuit. The most popular ZCS-PWM FB converter without an auxiliary circuit is the PSFB ZCS converter, which is a duality of PSFB ZVS topology [2] [4]. The main advantage of this topology is with simple structure, low device stress, and with ZCS conditions for all switches. However, the primary side switches must be unidirectional switches and the input must be current source, which is not common in the practical applications. The PWM type ZCS converter with auxiliary circuits has two types according to its locations, i.e., at the primary side Manuscript received September 9, 2004; revised August 2, 2005. This work was supported by the China National Science Fund under Contract 50237030. This paper was presented at the APEC 05 Conference. Recommended by Associate Editor F. L. Luo. The authors are with the College of Electrical Engineering, Zhejiang University, Hangzhou 310027, China (e-mail: zhangjm@zju.edu.cn). Digital Object Identifier 10.1109/TPEL.2005.869748 and at the secondary side. If the zero current-transition (ZCT) auxiliary circuit is located at the primary side, then usually two sets of ZCT auxiliary circuits are needed for each switch lag [5] [7], which complicates the whole topology structure. For those topologies, some primary side switches or auxiliary switches should be unidirectional switches and usually a diode should be connected in series with the primary side IGBT devices, which cause high extra conduction loss. Furthermore, the auxiliary switches or the rectifier diodes in these topologies are not softly switched. The interaction problem between the transformer s leakage inductance and the output rectifiers remains unresolved. As an improvement, the auxiliary circuit can also be placed at the secondary side. Due to the unidirectional characteristic of the rectifier diode, no unidirectional switch is needed for both the auxiliary switch and the primary side switches. And only one ZCT auxiliary circuit is needed as the topology structure is simplified, but the switching frequency of the auxiliary switch is doubled. Though the topology proposed in [8] achieves ZCS for all active switches as well as soft commutation for rectifier diodes, the parasitical voltage rings of the rectifier diodes and the duty cycle loss exist due to the resonant inductor in series with the power transformer. The topology proposed in [9] achieves ZCS for all switches and soft commutation for the rectifier diodes, it still suffers from the reverse recovery problem of the anti-paralleled diode of the auxiliary switch, which usually has poor reverse recovery characteristics especially when the MOSFET is adopted as the auxiliary switch. Therefore, two extra fast recovery diodes are usually needed, which also complicates the topology structure. In this paper, a novel ZCT FB dc/dc converter shown in Fig. 1 is proposed. By using a simple auxiliary circuit at the secondary side, the proposed converter not only achieves ZCS for both the main switches and the auxiliary switch in the entire load ranges but also realizes soft commutation for the output rectifier diodes. In addition, the resonant inductor in the auxiliary circuit also helps softly turn on the main switches. The reverse recovery 0885-8993/$20.00 2006 IEEE

ZHANG et al.: NOVEL ZERO-CURRENT-TRANSITION FULL BRIDGE DC/DC CONVERTER 355 Fig. 2. Steady-state operation waveforms. problem of the anti-paralleled diode of the auxiliary switch existed in the topology proposed in [9] is also eliminated. The detailed operation principle will be illustrated in Section II. The design considerations are given in Section III. Section IV will present some simulation and experimental results from a 100-kHz 1.5-KW dc/dc prototype with IGBT devices. II. PRINCIPLE OF OPERATION The proposed ZCT FB dc/dc converter is shown in Fig. 1. A simple auxiliary circuit consisting of a resonant inductor, resonant capacitor, auxiliary clamp diode, and auxiliary switch is attached in the secondary side to achieve ZCS for main switches. refers to the leakage inductance of the transformer, is the magnetizing inductance of the transformer. and are output rectifier diodes.,, and are dc input voltage, output voltage, and output current, respectively. The primary side to secondary side turn ratio of the transformer is. To simplify analysis of the steady-state circuit operation, it is assumed that all the components used in this converter have ideal characteristics, i.e., the leakage inductance is omitted; the output filter inductance is sufficiently large so that the output can be considered as a current source. There are 14 operating modes during a single steady-state switching cycle. The theoretical waveforms and equivalent circuits of each mode are shown in Figs. 2 and 3, respectively. Each operation mode is simply described as follows. Fig. 3. Equivalent circuits for each operation mode. (a) Mode 1. (b) Mode 2. (c) Mode 3. (d) Mode 4. (e) Mode 5. (f) Mode 6. (g) Mode 7. A. Mode1 [ - ] In this mode, S1 and S4 turn on and the power is delivered from input to the output. This operation mode is just the same as a conventional hard switching PWM FB converter. The magnetizing current increases linearly. B. Mode2 [ - ] At, the auxiliary switch turns on, the resonance between and start. At, the current through the resonant inductor reaches, and the transformer primary side current decreases to magnetizing current. The current through the diode decreases to zero. The resonant current in and the resonant voltage across are given as where,, is the turns ratio of the transformer (primary to secondary). C. Mode3 [ - ] At, turns off softly. The resonant capacitor is discharged linearly by the output current. The primary side current decreases to and remains peak magnetizing current. D. Mode4 [ - ] Due to the fact that the magnetizing current is so small that S1 and S4 can be turned off with ZCS at. The resonant capacitor is still discharged linearly by the output current. (1) (2)

356 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 21, NO. 2, MARCH 2006 The voltages across the collector to the emitter of S1 and S4 are charged up linearly by the magnetizing current. The rising slope depends on the magnetizing current and the parasitic capacitance of S1 and S4. E. Mode5 [ - ] At, the voltage across rectifier diode reduces to zero, turns on. Resonance between and starts again but with another path. As soon as the current through rises to, the resonance finished. The primary side current decreases to zero. F. Mode6 [ - ] The output current freewheels through and. If the resonant capacitor has been fully discharged, turns on and clamps to zero. G. Mode7 [ - ] S2 and S3 turn on at. The current through decreases linearly, the current through S2 and S3 rises linearly. Due to the existence of, soft turn on of S2 and S3 is achieved and soft commutation of is also achieved. In very light load condition, the capacitor voltage will not be fully discharged and is always off. And the anti-paralleled diode of is on. However, due to the very small output current and the existence of, the reverse recovery problem is still improved dramatically. From the analysis given above, the proposed topology achieves ZCS for all main switches and soft commutation for all power diodes. The operation modes during the next half switching cycle are the same as those described previously. III. DESIGN CONSIDERATIONS Based on the steady-state operation mode analysis discussed in Section II, the parameter design considerations for the auxiliary circuit and the power stage is discussed as follows. The power loss of the auxiliary circuit is also analyzed in this section. A. Parameters Design In order to achieve ZCS for main switches in all load conditions, the peak resonant current in the auxiliary circuit should be larger than the maximum output current. According to (1), and should satisfy where is the maximum output current. The resonant inductor is also limited by the maximum to overcome the reverse recovery problem of, which is given as (3) (4) The resonant period of and should be limited in a reasonable part in the whole switching period. Usually, the minimum turn on time of the primary switch should be larger than the required turn on time of the auxiliary switch, which is determined by the resonant period. If the resonant period is too long, the required minimum duty cycle will be increased. Otherwise, the ZCS condition will be lost at some particular conditions. Usually, the resonant period should be a small part of the switching period, which can be expressed as where is the resonant period, is the switching period, and represents the maximum acceptable value of, which should be a limited to a reasonable value, such as 20%. Based on (3) (5), the resonant parameters can be calculated, and the smaller one should be adopted in the design. The auxiliary switch should be turned on at a certain moment ahead of the turn off of the main switch as shown in Fig. 2. From Fig. 2, it is clear that the preferred conduction duration of is given as Due to the switching frequency is doubled at the secondary side, the switching frequency of the auxiliary circuit is also doubled. The maximum resonant voltage of is 2 V n. Therefore, the maximum voltage rating of rectifier diode ( ) will reach 3 V n at light load condition if a full-wave type rectifier is adopted, which is 1.5 times larger than the conventional PWM FB converter without consideration of parasitic oscillation. This is a main drawback of the proposed topology. However, the parasitic oscillation is usually very sever in the conventional PWM FB converter, which causes extra power loss and increased voltage rating of the rectifier diode. In the proposed topology, the parasitic oscillation is eliminated due to soft commutation of the rectifier diode. And a higher voltage rating of the rectifier diode will not cause higher power loss. B. Power Loss Analysis The auxiliary circuit in the secondary side causes extra power losses both in the primary side and the secondary side. In this part, these extra power losses compared to the hard-switching FB converter without the auxiliary circuit are analyzed. For simplicity, in the following analysis the magnetizing current of the transformer is omitted, and the output current is considered as a current source. The resonant current caused by the auxiliary circuit causes extra conduction losses both in the main switch and the rectifier diode. The peak resonant current is given in (7). The average current through the primary switch in a switching period is given in (8). For comparison, the average current through the primary switch in a hard-switching topology is given in (9) by simply assuming the duty cycle is the same as both for the (5) (6)

ZHANG et al.: NOVEL ZERO-CURRENT-TRANSITION FULL BRIDGE DC/DC CONVERTER 357 Fig. 4. Increased conduction power loss by resonant current. proposed ZCT topology and hard-switching FB with same input and output voltage (7) where is the duty cycle as shown in Fig. 2, which can be simply given as 2. Therefore, the conduction loss of the primary switch and conduction loss of the secondary side rectifier diode of the proposed topology are given as (8) (9) (10) (11) where is the collector-to-emitter saturation voltage of primary switch, and is the forward voltage drop of the rectifier diode. The conduction loss and are slightly larger than that in hard-switching FB topology. The increased conduction loss with various is shown in Fig. 4. It is clear that if 0.1, the increased conduction losses of both primary switch and the secondary rectifier is about 5%, which is quite low. The additional auxiliary circuit also causes extra power loss itself. In the first half resonant period, the inductor current flows through the auxiliary switch, when changes direction, the body diode of the auxiliary switch turns on. And the auxiliary diode turns on when the resonant capacitor is fully discharged. Therefore, the related power loss of auxiliary switch and the power loss of auxiliary diode are given as (12) (13) where, is the collector-to-emitter saturation voltage of auxiliary switch, and is the forward voltage drop of the auxiliary switch s body diode. is the forward voltage drop of the auxiliary diode. Fig. 5. Simulation waveforms (Trace from top to Bottom: gate drive voltage V of S1 and S2, gate drive voltage V of S, collector current I of S1 and S2, collector to emitter voltage V of S2, diode voltage of D and D, resonant capacitor voltage V ). (a) Full load condition. (b) Light load condition. The calculated power loss of auxiliary diode using (10) can be negative if the output current is only a small part of the peak resonant current. This means that the resonant capacitor is not fully discharged and the auxiliary diode will never turns on in such condition. The reverse recovery problem is much alleviated due to the limited,, and the reverse recovery power loss should be considered with selected devices and circuit parameters individually for each practical application. The power loss of resonant inductor includes core loss and winding loss. This power loss depends on the circuit parameters and inductor design. The positive and negative peak current of the inductor is and, respectively, which determines the core loss with given core and gap. The RMS current of the inductor is given in (14). And the winding loss of can be calculated based on (14) From the power loss analysis given previously, the extra power losses caused by the auxiliary circuit are presented. If the saved switching loss of the primary switch and the reverse recovery power loss of the rectifier diode are larger than these extra power losses, the proposed topology usually has a higher efficiency.

358 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 21, NO. 2, MARCH 2006 Fig. 6. Circuit diagram of the prototype with extra clamp diode. TABLE I KEY PARAMETERS OF THE EXPERIMENTAL PROTOTYPE Fig. 8. Voltage waveform of diode D (D ). Fig. 9. Resonant capacitor voltage V at full load condition. Fig. 7. Collector emitter voltage V and collector current I of S2. IV. SIMULATION EXPERIMENT VERIFICATION To verify theoretical analysis of the proposed topology, the PSIM simulation results at different load conditions are presented in Fig. 5. A 1.5-KW 100-kHz prototype as battery charger in power stations is built to verify the theoretical analysis and simulation results. The input voltage is 300 400 V and the output voltage is 120 V. A clamp diode is added to improve the performance of the auxiliary circuit as shown in Fig. 6. First, is activated as a clamp diode to clamp the voltage of. At the beginning of the turn-on of the primary switch, parasitic ring between,, and parasitic capacitance of occurs. Due to the fact that is usually much larger than the parasitic capacitance of,it can be neglected in the resonance. Therefore, the voltage stress of will be very high due to this parasitic ring. With the added clamp diode, this parasitic ring can be prevented. Second, can also provide a path of current if the auxiliary switch suddenly turns off during the first half of the resonant period. Fig. 10. Collector emitter voltage V and collector current I of S. Fig. 11. Voltage waveform of diode D.

ZHANG et al.: NOVEL ZERO-CURRENT-TRANSITION FULL BRIDGE DC/DC CONVERTER 359 Fig. 14. Efficiency versus input voltage at P = 1250 W, output voltage 120 V. Fig. 12. Current waveform of resonant inductor. it is clear that the efficiency of the proposed topology is about 3% higher at full load condition. The efficiency at full load condition and at different input voltage is shown in Fig. 14. With the increased input voltage, the efficiency slightly decreases. This is mainly caused by the increased resonant current. At 300-V input condition, the efficiency is slightly lower than that at 320-V input. At 300-V input, the peak resonant current is slightly lower than the output current due to parasitic loss of the circuit. Therefore, turn-off loss still has an effect on the efficiency. V. CONCLUSION Fig. 13. Efficiency versus output current at V = 300 V, output voltage 120 V. A SG3525 is adopted as the control integrated circuit (IC). In the prototype, IRG4PC50U without body diode is adopted as an auxiliary switch, and an extra fast recovery diode is needed to parallel with the auxiliary switch. The resonant inductance is 1.75 H, with three turns of the ER28L/3F3 core from Ferroxcube, and the gap is about 1 mm. The magnetizing inductance of the transformer is 2.1 mh with E55/TP3 magnetic core from TDG. The primary winding is 18 turns with two strands 0.20 25 Litz wire in parallel. Each secondary winding is eight turns with two strands of 0.20 25 Litz wire in parallel. The leakage inductance of the transformer is about 3.5 H. The key parameters of the prototype are listed in Table I. The main experimental results are given in Figs. 7 12. From these figures, ZCS of all active switches and soft commutation of power diodes are achieved. From Fig. 10, when turns off, a parasitic resonant between,, and parasitic capacitance of can be founded. Due to the existence of, the voltage is clamped and the resonance is quickly damped. A small current ring due to this resonance exists both in the current of and resonant inductor, which can be seen from Figs. 10 and 12. From the experimental results, it is clear that they match the simulation results and theoretical analysis very well. The measured efficiency at 300-V input is shown in Fig. 13. For comparison, the efficiency from a hard-switching FB converter with the same parameters is also presented. From Fig. 13, In conclusion, this paper proposes a novel ZCT FB dc/dc converter. The proposed converter has very attractive features, such as ZCS for all active switches, no diode reverse recovery problems due to soft commutation of rectifier diodes, simple topology structure, and convenient control strategy. It seems more attractive in high power applications using IGBTs as main switches, and high efficiency can be achieved. Experimental results confirm the theoretical and simulation analysis. REFERENCES [1] R. Farrington, M. M. Jovanovic, and F. C. Lee, Analysis of reactive power in resonant converters, in Proc. PESC 92 Conf., 1992, pp. 197 205. [2] G. Hua, Soft-switching techniques for pulse-width-modulated converters, Ph.D dissertation, Virginia Polytech. Inst. State Univ., Blacksburg, 1994. [3] C. Iannello, S. Luo, and I. Batarseh, Steady-state analysis of full-bridge zero-current switching converter, Electron. Lett., vol. 36, no. 13, pp. 1098 1099, Jun. 2000. [4], Full bridge ZCS PWM converter for high-voltage high-power applications, IEEE Trans. Aerosp. Electron. Syst., vol. 38, no. 2, pp. 515 526, Apr. 2002. [5] D.-Y. Lee, M.-K. Lee, D.-S. Hyun, and I. Choy, New zero-current-transition PWM dc/dc converters without current stress, IEEE Trans. Power Electron., vol. 18, no. 1, pp. 95 104, Jan. 2003. [6] X. H. Wu, D. M. Xu, J. H. Kong, C. Yang, and Z. Qian, High power high frequency zero current transition full bridge dc/dc converter, in Proc. APEC 98 Conf., 1998, pp. 823 827. [7] M. Marx and D. Schroder, A novel zero-current-transition full-bridge DC-DC converter, in Proc. PESC 96 Conf., 1996, pp. 664 669. [8] S. Atoh and H. Yoshike, PWM DC-DC converter with a resonant commutation means, in Proc. INTELEC 91 Conf., 1991, pp. 308 313. [9] D. M. Xu, X. H. Wu, J. M. Zhang, and Z. Qian, High power high frequency half-wave-mode ZCT-PWM full bridge dc/dc converter, in Proc. APEC 00 Conf., 2000, pp. 99 103.

360 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 21, NO. 2, MARCH 2006 Junming Zhang was born in Zhejiang, China, in 1975. He received the M.S. degree and Ph.D. degrees in electrical engineering from Zhejiang University, Hangzhou, China, in 2000 and 2004, respectively. He holds one patent. His research interests include power electronics system integrations, PFC techniques, dc/dc converter, synchronous rectifier and high power inverters. Guoliang Wu was born in Zhejiang, China, in 1981. He received the B.S. degree in electrical engineering from Zhejiang University, Hangzhou, China, in 2004, where he is currently pursuing the M.S. degree. His research interests include soft-switching techniques, dc/dc converters, and power electronics system integrations. Xiaogao Xie was born in Leiyang, China, in 1975. He received the M.S. and Ph.D. degrees in electrical engineering from Zhejiang University, Hangzhou, China, in 2000 and 2005, respectively. His research interests include dc/dc converters and soft-switching techniques. Xinke Wu was born in Jiangsu province, China, in 1978. He received the B.S. and M.S. degrees in electrical engineering from the Harbin Institute of Technology, Harbin, China, in 2000 and 2002, respectively, and is currently pursuing the Ph.D. degree in electrical engineering at Zhejiang University, Hangzhou, China. His research interests include soft switching of power conversion, power factor correction, and power electronics system integration. Zhaoming Qian (SM 92) received the M.S. degree in radio engineering from the Electrical Engineering Department, Zhejiang University, Hangzhou, China, in 1961 and the Ph.D. degree in applied science from the Catholic University of Leuven and the Interuniversity Microelectronics Center (IMEC), Leuven, Belgium, in 1989. Since 1961, he has been teaching and doing research on electronics and power electronics at Zhejiang University. He was promoted to a Professor of the Electrical Engineering Department, Zhejiang University, in 1992. He is currently the Deputy Director of the National Engineering Research Center for Applied Power Electronics, Zhejiang University, and the Deputy Director of the Scientific Committee, National Key Laboratory of Power Electronics, Zhejiang University. He has published one book on EMC design and more than 200 papers. His main professional interests include power electronics and its industrial applications, power electronic system integration, and EMC in power electronic systems etc. Dr. Qian received Excellent Education Awards from the China Education Commission and from Zhejiang University in 1993, 1997, and 1999, respectively, and Science and Technology Development Awards from the China Education Commission, in 1999 and 2003, respectively.