IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 8, NO. 1, DECEMBER 013 5437 An LLC Resonant DC DC Converter for Wide Output Voltage Range Battery Charging Applications Fariborz Musavi, Senior Member, IEEE, Marian Craciun, Member, IEEE, Deepak S. Gautam, Student Member, IEEE, Wilson Eberle, Member, IEEE, and William G. Dunford, Senior Member, IEEE Abstract In this paper, resonant tank design procedure and practical design considerations are presented for a high performance LLC multiresonant dc dc converter in a two-stage smart battery charger for neighborhood electric vehicle applications. The multiresonant converter has been analyzed and its performance characteristics are presented. It eliminates both low- and highfrequency current ripple on the battery, thus maximizing battery life without penalizing the volume of the charger. Simulation and experimental results are presented for a prototype unit converting 390 V from the input dc link to an output voltage range of 48 7 V dc at 650 W. The prototype achieves a peak efficiency of 96%. Index Terms Batteries, dc dc power converters, electric vehicles, resonant converters. I. INTRODUCTION NEIGHBORHOOD electric vehicles (NEVs) are propelled by an electric motor that is supplied with power from a rechargeable battery [1], []. Presently, the performance characteristics required for many electric vehicle (EV) applications far exceed the storage capabilities of conventional battery systems. However, battery technology is improving and as this transition occurs, the charging of these batteries becomes very complicated due to the high voltages and currents involved in the system and the sophisticated charging algorithms [3]. Quick charging of high capacity battery packs causes increased disturbances in the ac utility power system, thereby increasing the need for efficient, low-distortion smart chargers. A smart charger is a battery charger that can respond to the condition of a battery, and modify its charging actions according to the battery algorithm. Conversely, a standard, or simple battery charger supplies a constant dc or pulsed dc power source to a battery being charged. A simple charger does not alter its out- Manuscript received October 11, 01; revised December 3, 01; accepted January 15, 013. Date of current version June 6, 013. This paper was presented at the IEEE Applied Power Electronics Conference and Exposition, Orlando, FL, 01. This work was supported by Delta-Q Technologies Corp. Recommended for publication by Associate Editor S. Crozier. F. Musavi, M. Craciun, and D. S. Gautam are with Delta-Q Technologies Corporation, Burnaby, BC V5G 3H3 Canada (e-mail: fmusavi@delta-q.com; mcraciun@delta-q.com; dgautam@delta-q.com). W. Eberle is with the School of Engineering, University of British Columbia, Okanagan, Kelowna, BC V1V 1V7 Canada (e-mail: Wilson.eberle@ubc.ca). W. G. Dunford is with the Department of Electrical Engineering, University of British Columbia, Vancouver, BC V6T 1Z4 Canada (e-mail: wgd@ece.ubc.ca). Color versions of one or more of the figures in this paper are available online at http://ieeexplore.ieee.org. Digital Object Identifier 10.1109/TPEL.013.4179 Fig. 1. Typical battery charging power architecture. put based on time or the charge on the battery. Therefore, smart chargers are preferred for NEV battery charging applications. The proposed NEV battery charger power architecture includes an ac dc converter with power factor correction (PFC) [4], followed by an isolated dc dc converter, as shown in Fig. 1 [5]. This architecture virtually eliminates the low- and high-frequency current ripple charging the battery without using a bulky filter capacitor. Instead, it uses a high-frequency transformer. The architecture maximizes battery life without penalizing the charger volume. In the work that follows, the front-end ac dc PFC converter is a conventional continuous conduction mode (CCM) boost topology [6], [7]. The second-stage dc dc converter is a half-bridge multiresonant LLC converter [8]. The criteria for choosing these topologies include high reliability, high efficiency, and low component cost. The half-bridge resonant LLC converter is widely used in the telecom industry for its high efficiency at the resonant frequency and its ability to regulate the output voltage during the hold-up time, where the output voltage is constant and the input voltage might drop significantly [9] [1]. Resonant converters have been used in many applications, including induction heating [13], and fuel cells [14]. However, the wide output voltage range requirements for a battery charger are drastically different and challenging compared to telecom applications, which operate in a narrow output voltage range. Fig. illustrates a simplified battery charging profile. It has four distinct operating modes: bulk, absorption, equalization, and maintenance. In the bulk mode, the charger limits the maximum charging current to a preset I MAX value while monitoring the battery voltage. In the absorption mode, the charger elevates the voltage to V ABS while monitoring the current. This voltage is just below the battery gassing voltage. When it has reached this point, the battery is between 70% and 90% state of charge (SOC). When the current decreases to a preset value, I OCT, the charger enters the equalization mode. When it has reached this point, the battery is at 100% SOC. Equalizing is an overcharge performed on lead acid batteries after they have been fully charged. This function equalizes the cell voltages in a battery module. In the maintenance mode, the charging action 0885-8993/$31.00 013 IEEE
5438 Fig.. IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 8, NO. 1, DECEMBER 013 Simplified adaptive 4-step lead acid battery charging profile. Fig. 4. Typical dc gain characteristics of an LLC converter using first harmonic approximation. limits, in addition to the maximum frequency of the LLC converter. Furthermore, in order to revive a fully depleted battery, the charger must operate out of this plane, from near zero voltage which is far below the maximum operating frequency boundary of the converter. Satisfying these requirements requires a new design strategy for an LLC resonant converter. The following sections provide detailed practical considerations for designing an LLC multiresonant converter in a battery charging application. II. DESIGN PROCEDURE Fig. 3. Desired lead acid battery V I plane on a 650-W charger for 48-V system. is finished and the charger only supplies limited current to offset the internal soft discharge. As indicated in Fig., the battery voltage, at the dc dc converter output, can vary from as low as 36 V to as high as 7 V. Therefore, the design requirements for selecting the resonant tank components are significantly different than those in telecom applications featuring a constant output voltage [15]. The authors in [16] [18] addressed wide output range applications; however, the design procedures are given for resonant tank components using the first harmonic approximation proposed in [19] and [0], which is only valid for frequencies close to the resonant frequencies. The resonant tank design guidelines utilize information from the lead acid battery V I plane, provided in Fig. 3. It illustrates the key design points and limitations on a 650-W charger for a 48-V system. This V I plane dictates the design criteria for the half-bridge multiresonant LLC converter, in particular the resonant tank components, Lr, Lm, and Cr. The outer range of the operating plane is constrained by the constant voltage (CV), constant power (CP), constant current (CC), and short circuit The life and capacity of EV batteries depend on several factors, such as cycle count, charge mode, maintenance, temperature, and age. Among these factors, the charge mode has a significant impact on battery life and capacity. EV batteries should be charged with current and voltage levels with low ripple. In addition, the basic requirements for battery chargers are small size and high efficiency, which can be achieved using soft switching techniques. reduce the switching losses that result from high-frequency operation, resonant power conversion can be used. There are several publications and application notes in industry focusing on resonant circuit design. There are two major issues with the existing work: 1) the output voltage is considered constant (e.g., typical for telecom applications), which is not a valid assumption in battery charging; and ) the ratio of the transformer magnetizing inductance and the resonant inductance (including the leakage inductance) is given by some suggested values without considering the effect of the short circuit condition on the resonant network. As a result, the sequence of designing the resonant network is different for battery charging applications. Fig. 4 illustrates a family of typical dc gain characteristics for an LLC converter as a function of normalized switching frequency for seven different load conditions varying from
MUSAVI et al.: ANLLC RESONANT DC DC CONVERTER FOR WIDE OUTPUT VOLTAGE RANGE BATTERY CHARGING APPLICATIONS 5439 no-load to short circuit. Resonance occurs at unity gain, where the resonant capacitors and series resonant inductor are tuned. By design, this is the point where the converter is required to deliver maximum power, and in Fig. 3 it is marked as Design Point f s. A second resonance occurs at the peak of the bellshaped curves, which is the boundary between ZVS (Region ) and ZCS (Region 3). The maximum gain is achieved at the second resonant frequency, where the resonant capacitors are tuned with series resonant inductor and parallel magnetizing inductance. In Fig. 3, this point of maximum gain is labeled Design Point L m. Achieving maximum efficiency requires operation close to frequencies where the resonant tank impedance is very low. Wide range for input and/or output operating voltages requires operation on the steep portion of the curve below resonance in Region. In addition, maintaining ZVS operation [1], while avoiding ZCS operation in Region 3, requires operation in Region. Fast overload and short circuit protection circuitry is required in order to avoid possible ZCS operation in the capacitive Region 3 [], [3]. The following step by step procedure is provided in order to meet these requirements. A. Initial Design Parameters The input voltage range, output voltage range, maximum output power, and resonant frequency are the initial design parameters to be defined. The input voltage to the dc dc stage is determined by the PFC bus output voltage at the dc link capacitors. Unlike telecom power supply applications, there is no hold-up time requirement and the variation seen by the LLC converter is only the low-frequency ripple on the PFC bus capacitors, which is 370 to 410 V with a nominal value of 390 V. The output voltage range of the dc dc stage is from 36 to 7 V with a nominal value of 48 V. The maximum power of 650 W is defined at the nominal output voltage, 48 V. The choice of switching frequency depends on the cost-loss tradeoff between the Litz wire gauge used in the transformer and resonant inductor magnetics, and the cost and power loss in the wire [4]. For the given 390 V input and 650 W application, the optimized Litz wire strand gauge was determined to be 44 AWG at a 00 khz switching frequency using [4]. B. Maximum Switching Frequency, Maximum Dead Time The maximum switching frequency is limited by the voltage controlled oscillator in the control IC and the reflected junction capacitance of the output rectifiers [5], [6]. The ac equivalent circuit of the LLC resonant converter including parasitic components is illustrated in Fig. 5. As it was demonstrated in [5], the normalized dc gain equation of the converter is modified by the inclusion of the rectifier diode junction capacitances and as a result increasing the switching frequency decreases the output voltage until, the circuit resonates with the diode junction capacitances at which point, further increasing the switching frequency tends to increase the output voltage. Limiting the maximum switching frequency is an approach to prohibit the converter from operating at frequencies where the junction capacitances of the rectifier Fig. 5. AC equivalent circuit of LLC resonant converter including parasitic components. diodes and other parasitic elements in the circuit resonate, causing an undesired increase in the output voltage. As a result, the maximum switching frequency should be limited to.5 times the resonant frequency. C. Selecting Transformer Turns Ratio, N n The transformer turns ratio should be selected at the resonant frequency where the gain is unity and can be calculated using (1), where V d represents the diode voltage drop of the output rectifier N n = D. Calculating Resonant Inductor, L r V in(nom) (V o(min) + V d ). (1) The minimum resonant inductor must be selected to limit the maximum output current in the short circuit condition and limit the converter to its maximum switching frequency. The minimum inductance is given by () L r(scc) = N n V in(nom) V o(nom) 8f s max P o. () E. Calculating Resonant Capacitor, C r Once the value of the resonant inductor is determined, the resonant capacitor value can be calculated using (3) C r(res) = 1 ( π f o ) L r(scc). (3) F. Characteristic Impedance and Quality Factor After determining the resonant inductor and capacitor values, the characteristic impedance of the resonant network is given by (4) L r(scc) Z o =. (4) C r(res) The minimum quality factor can then be calculated using (5) and (6) as proposed in [1] Q min = Z o R ac max (5) R ac max = 8N n Vo max π. (6) P o
5440 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 8, NO. 1, DECEMBER 013 G. Calculating Magnetizing Inductance, Lm The choice of magnetizing inductance has to fulfill two conditions. First, to achieve ZVS under no load conditions, a maximum magnetizing inductance, Lm (ZVS), is required as given by (7). In (7), the total capacitance of the half-bridge, CHB, is the sum of stray capacitance of the printed circuit board, the Q1 and Q MOSFET output capacitances, and the transformer and inductor inter-winding capacitances. In order to realize the required maximum gain at the minimum switching frequency as noted in [7], Lm (m ax) is given by (8) tdead.nn.vo(m in) 4 f s 1 m a x t d e a d Lm (ZVS) = (7) CHB Vin(m ax) Lm (m ax) = Lr (scc) fo π f s m i n 1. 4 1 M d c1 m a x (8) The final value for the magnetizing inductance is the minimum of the two values. However, if Lm (m ax) < Lm (ZVS), then a new dead time should be calculated using the value from Lm (m ax), and replaced in (7) to get a new dead time. Finally, the total inductance value must satisfy the energy balance in the total capacitance of the half-bridge, using (9) and (10) 1 (Lm (m in) + Lr (scc) )Im pk Im pk 1 CHB Vin(m ax) Nn Vo(m in) =. 4Lm > Fig. 6. MOSFET current and resonant capacitor voltage. Fig. 7. MOSFET RMS current for different input and output voltage. (9) (10) III. PRACTICAL DESIGN CONSIDERATIONS A detailed design procedure and resonant tank component selection were provided n in the previous section. Additional practical design considerations addressing MOSFET selection, resonant and output capacitor selection, output diode rectifier selection, controller IC selection, and the power and current limit restrictions are provided in the sub-sections that follow. A. MOSFET Selection For MOSFET selection, consideration must be given to practical dv/dt turn-on limits, body diode reverse recovery [8], and losses. Fig. 6 illustrates the voltage across the resonant capacitor and the current through the MOSFET as a function of time at turn-on, where the switching frequency is under the unity gain resonant frequency. With ZVS turn-on and low turn-off loss, MOSFET losses are dominated by conduction loss. In order to calculate the conduction losses, a precise model of the switch current is required. Equations (11) and (1) represent the current through the switch as shown in Fig. 6 π IPK sin t θ, t (11) IM OSFET (t) = N V T Ts n o o, t< 4Lm θ = Arcsin Nn V o T o 4Lm IPK. (1) Using (11), the switch RMS current can be calculated using (13) Ts IM OSFET rm s = (IM OSFET (t)) dt. (13) Ts 0 Fig. 7 shows the MOSFET RMS current as a function of output voltage at minimum, nominal, and maximum input voltages (i.e., the PFC bus voltage). It is noted that this model is only valid for switching frequencies below the unity gain resonant frequency. As shown in Fig. 1 in [9], modern low drain to source on resistance (RDSon ) vertical structure MOSFETs have large Coss capacitance at low VDS voltage. This helps reduce turn-off loss since the drain current at the turn-off instant is divided between
MUSAVI et al.: AN LLC RESONANT DC DC CONVERTER FOR WIDE OUTPUT VOLTAGE RANGE BATTERY CHARGING APPLICATIONS 5441 Fig. 8. Resonant inductor current, MOSFET voltage, and gating signal during switch turn-off. the drain to source (CDS ) and gate to source (CGS ) intrinsic capacitors. When the drain voltage reaches approximately 10 V, Crss starts to decrease suddenly up to 100 times at 40 V (two decades) and the drain current has to continue to flow through CDS and the MOSFET channel. If the MOSFET channel current is zero, this results in lossless turn-off. The CDS value itself has dropped to less than one tenth of its 0 V value so this sudden current redistribution increases the resonant transition dv/dt and triggers a high-frequency (i.e., >100 MHz) oscillation between the CDS and stray plus drain and source package inductances until the opposite MOSFET reduces transition dv/dt when its VDS voltage drops below 40 V. The result is that up to a critical turn-off current the transition is lossless, above this critical value the transition dv/dt and turn-off loss increase rapidly. Unfortunately, there is no easy way to measure this critical current since a drain current measurement cannot differentiate between the MOSFET channel and parasitic capacitance currents, but evidence of this behavior can be observed in the MOSFET VDS and ID waveforms, as shown in Fig. 8. For the ST Microelectronics STB3NM60ND MOSFET, the capacitance inflection point can be approximated at 40 V. Then, the total capacitance of the half-bridge switching node CHB can be modeled as a piecewise effective capacitance of 160 pf for 40 V < VHB < (Vin 40 V), and 5.4 nf for VHB < 40 V and (Vin 40 V) < VHB < Vin. For the half-bridge high-side MOSFET, at the end of the resonant transition, the CDG value increases by a factor of 100, injecting more current into the gate drive circuit in the presence of the undamped transitional high-frequency oscillation, so the gate drive has to be able to sink this current without raising the gate voltage and turning-on the MOSFET prematurely. However, the V I characteristic in Fig. 3 exhibits the charging rate reduction for battery voltages below 1 V/cell in order to limit the turn-off current below Icrit and maintain minimal turn-off loss. B. Resonant and Output Capacitor Selection The resonant capacitor voltage is closely associated with the resonant inductor current, which is partially the same as the Fig. 9. Resonant capacitor voltage for different input and output voltage. MOSFET current. By definition, the voltage across resonant capacitor is given by (14), enabling the capacitor peak-to-peak voltage to be expressed by (15) tal Charge Cr (res) Ts IM OSFET (t) dt = 0. Cr (res) VC p k p k = (14) VC p k p k (15) Assuming the voltage waveform is sinusoidal, the average dc and RMS ac values for the resonant capacitor are given by (16) and (17), respectively Vin VC p k p k. = VC d c = VC a c rm s (16) (17) Curves of the resonant capacitor voltage are provided in Fig. 9 as a function of output voltage at minimum, nominal, and maximum input voltages (i.e., the PFC bus voltage). This model is only valid for switching frequencies below the unity gain resonant frequency. There are two technologies available to satisfy the resonant capacitor value and voltage stress requirements: ML ceramic capacitors and film capacitors. The problems with ML ceramic capacitors include maximum size limitations due to failure under board flexing and a failure mode that results in a short circuit. Thus, they are undesired for resonant and output filter capacitors. Alternately, polypropylene film capacitors have several characteristics that make them a good candidate for the resonant capacitor. They are more stable with temperature, have higher temperature ratings, exhibit self-healing characteristics, and are relatively inexpensive. The capacitor selected for the resonant network was an EPCOS MKP series B360L. The permissible voltage across the capacitor given in Fig. 9 must match the voltage given by the data sheet [30]; page 15 at 700 V ac. The output capacitors have to handle very high ripple current as well as all the requirements given for resonant capacitors.
544 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 8, NO. 1, DECEMBER 013 Fig. 11. Photo of the prototype LLC dc dc converter. D. Power Limit and Current Limit Restrictions Fig. 10. voltage. Output capacitor ac ripple current for different input and output In order to express the output capacitor ac ripple current, the secondary side instantaneous current is required and is given by (18), enabling the output capacitor ac ripple current to be calculated using (19) (1) sec (t) = Nn IM OSFET (t) + sec ac = sec rm s = sec ave = 1 sec rm s 1 sec Nn Vo 4Lm 1 (18) 4 sec ave (19) (t) dt (0) 0 sec (t) dt. (1) 0 Fig. 10 illustrates the output capacitor ac ripple current as a function of output voltage for different input voltages. As can be noted, the ac ripple current is at its peak, when both the input voltage and output voltage are at their minimum values. For this application, metalized polyester film capacitors are best suited. Accordingly, an EPCOS MKT series B356 output capacitor was selected. C. Output Diode Rectifier Selection The output diode rectifiers exhibit ZCS for both turn-on (zero di/dt) and turn-off. Hence, the parameters to consider in the selection of the diodes are forward drop, VFW D and junction capacitance, Cj. The available technologies include Schottky and ultrafast diodes. Schottky diodes demonstrate lower forward drop, but have relatively higher junction capacitance compared to ultrafast diodes. Higher junction capacitance limits the maximum switching frequency of the converter and no load regulation of output voltage. This is usually dealt with using a dummy load. If there is no dummy load at the output, then ultrafast diodes should be chosen. The V I plane provided in Fig. 3 illustrates the limitations on the output voltage, output power, and output current. The output current and output voltage are controlled by the feedback control circuit as per the voltage and current references provided by the battery charging algorithm requirements. However, the output power limitation has two distinct curves. The actual power limit is implemented by software, as a hyperbolic function given by () = Po (Const.). Vo () In addition, there is a faster power limit implemented by hardware, which is a straight line connecting the constant current limits to the constant voltage limits. The deviation of the hardware limit from the actual power limit is 5 W at.5 V/Cell. E. Control IC Selection There are several commercial off-the-shelf control ICs available for LLC resonant converters, each having distinct characteristics and limitations. The key requirements for selecting a controller are switching frequency range, operating temperature range, ability to control the secondary side, programmable deadtime, programmable soft start, protection and brownout detection. Given the requirements for the battery charging application investigated, the NCP1395B control IC from On Semiconductor was selected. IV. EXPERIMENTAL RESULTS A prototype of the half-bridge LLC multiresonant converter was built to provide a proof-of-concept and verify the analytical work presented in this paper. Fig. 11 shows a picture of the LLC dc dc multiresonant converter prototype. Table I provides the design criteria for the prototype LLC converter. In Table II, the key components used in the prototype converter are given. Curves of the efficiency of the converter as a function of load are given in Fig. 1 for output voltages of 48, 60, and 7 V. These measurements were taken with the output relay, common mode EMI inductor, and output fuse included. Experimental waveforms of the resonant tank current, resonant capacitor voltage, and voltage across bottom MOSFET, Q are provided in Figs. 13 and 14 at Vin = 390 V, and Po = 650 W.
MUSAVI et al.: AN LLC RESONANT DC DC CONVERTER FOR WIDE OUTPUT VOLTAGE RANGE BATTERY CHARGING APPLICATIONS 5443 TABLE I DESIGN SPECIFICATIONS Fig. 13. Resonant tank current (IL r ), resonant capacitor voltage (V C r ), and voltage across Q (V Q ) at V o = 48 V, P o = 650 W, and fs = 11 khz (Y -axis: Ch1 = V Q 100 V/div. Ch = V C r 100 V/div. Ch4 = IL r A/div. X -axis: 1 μs/div). TABLE II COMPONENTS USED IN THE PROTOTYPE CONVERTER Fig. 14. Resonant tank current (IL r ), resonant capacitor voltage (V C r ), and voltage across Q (V Q ) at V o = 7 V, P o = 650 W, and fs = 11 khz (Y -axis: Ch1 = V Q 100 V/div. Ch = V C r 100 V/div. Ch4 = IL r A/div. X -axis: 1 μs/div). Fig. 1. Measured efficiency versus output power at V o = 48 V and fsw = 11 khz, V o = 60 V and fsw = 170 khz, V o = 7 V and fsw = 15 khz. The waveforms in Fig. 13 are given close to the unity gain resonant frequency, fs = 11 khz, and output voltage, Vo = 48 V. The waveforms in Fig. 14 are given at fs = 15 khz, and an output voltage of Vo = 7 V. In both figures, the potential for ZVS turn-on is noted, since the resonant current lags the voltage across MOSFET Q. Fig. 15. Output current ripple ( R ip p le ) and output voltage ripple (V o R ip p le ) at V o = 48 V, = 13.5 A and battery simulator load (Y -axis: Ch1 = V o R ip p le 500 mv/div. Ch4 = R ip p le 100 ma/div. X -axis: 5 ms/div). Fig. 15 provides waveforms of the output voltage and current low-frequency ripple at full load. The low-frequency current ripple is limited to 537 ma at a dc load current of 13.5 A. The low-frequency voltage ripple is 140 mv at an output voltage of 48 V.
5444 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 8, NO. 1, DECEMBER 013 V. CONCLUSION A resonant tank design procedure and practical design considerations were presented for a high performance LLC multiresonant dc dc converter in a two stage wide output voltage range smart battery charger for NEV applications. The multiresonant converter has been analyzed and its performance characteristics presented. It eliminates both low- and high-frequency current ripple on the battery, thus maximizing battery life without penalizing the volume of the charger. Experimental results were presented for a prototype unit converting 390 V from the input dc link to an output voltage range of 48 7 V dc at 650 W. The prototype achieves a peak efficiency of 96%. REFERENCES [1] D.W. Gao, C. Mi, and A. Emadi, Modeling and simulation of electric and hybrid vehicles, Proc. IEEE, vol. 95, no. 4, pp. 79 745, Apr. 007. [] A. Emadi, S. Williamson, and A. 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Electron., vol. 56, no. 5, pp. 140 147, May 009. [7] Y. Gu, Z. Lu, L. Hang, Z. Qian, and G. Huang, Three-level LLC series resonant DC/DC converter, IEEE Trans. Power Electron., vol. 0, no. 4, pp. 781 789, 005. [8] Won-suk Choi, Sung-mo Young, and Dong-wook Kim, Analysis of MOS- FET failure modes in LLC resonant converter, in Proc. Int. Telecommun. Energy Conf., INTELEC, 009, pp. 1 6. [9] STB3NM60ND, N-channel 600 V, 0.150 Ω typ., 19.5 A, FDmesh TM II Power MOSFET, STMicroelectronics, 01. [30] Film Capacitors: Metallized Polypropylene Film Capacitors (MKP), Series/Type: B3601 L B360 L, EPCOS AG, 009. Fariborz Musavi (S 10 M 11 SM 1) received the B.Sc. degree from Iran University of Science and Technology, Tehran, Iran, in 1994, the M.Sc. degree from Concordia University, Montreal, QC, Canada, in 001, and the Ph.D. degree in electrical engineering with emphasis in power electronics from the University of British Columbia, Vancouver, BC, Canada. Since 001, he has been with several high tech companies including EMS Technologies Inc., Montreal; DRS Pivotal Power, Bedford, NS, Canada; and Alpha Technologies, Bellingham, WA. He is currently with Delta-Q Technologies Corporation, Burnaby, BC, Canada, where he is the Manager of Research, Engineering and is engaged in research on the simulation, analysis, and design of battery chargers for industrial and automotive applications. His current research interests include high-power high-efficiency converter topologies, high-power-factor rectifiers, electric vehicles, and sustainable and renewable energy sources. Dr. Musavi is a Registered Professional Engineer in the Province of British Columbia, Canada. He was the recipient of the First Prize Paper Award from the IEEE Industry Applications Society Industrial Power Converter Committee in 011. He was also a recipient of an award from the Power Sources Manufacturers Association to present papers at conferences.
MUSAVI et al.: ANLLC RESONANT DC DC CONVERTER FOR WIDE OUTPUT VOLTAGE RANGE BATTERY CHARGING APPLICATIONS 5445 Marian Craciun (M 00) received the B.Sc. degree in electronics engineering at The Polytechnic Institute of Bucharest, Bucharest, Romania. He has over 0 years of experience in developing telecom and industrial power electronics products and sustaining engineering. His industrial experience includes positions at Energorepairs RENEL and Asea Brown Boveri Ltd, Bucharest, Romania; Argus Technologies Ltd and Alpha Technologies Ltd, Burnaby, BC, Canada. He is currently a Power Electronics R&D Engineer at Delta-Q Technologies Corp in Vancouver. His current research interests include high-power high-efficiency converter topologies, high-power-factor rectifiers, resonant converters, electric vehicles, and sustainable and renewable energy sources. Deepak S. Gautam (M 09 S 11) received the B.E. degree in electronics engineering from Mumbai University, Maharashtra, India, in 000, and the M.A.Sc. degree in electrical engineering from the University of Victoria, BC, Canada, in 006. He is currently pursuing the Ph.D. degree in electrical engineering in the field of power electronics from the University of British Columbia, Vancouver, BC Canada. From 000 to 003, he worked as a Research and Development Engineer for the Power Conversion and Control division of Aplab Limited, Mumbai, India where he was involved in development of linear, switch-mode and programmable power supplies for industrial and telecom industries. Since 007, he has been working for Delta-Q Technologies Corp., Burnaby, BC, Canada as a Power Electronics Engineer where his main responsibility is to develop highfrequency switch-mode battery chargers for automotive and industrial application. His research interests are dc dc converters, ac dc power factor correction converters, resonant converters, and feedback control circuits. Prof. Gautam is the recipient of the University of Victoria fellowship, Andy Farquharson award for excellence in graduate student teaching, and best poster presentation award at the APEC 01 conference in Orlando, FL, USA. He also has won travel grants from the Power Source Manufacturer s Association and IEEE Industry Application Society and Power Electronics Society to present papers at conferences. Wilson Eberle (S 98 M 07) received the B.Sc., M.Sc., and Ph.D. degrees from the Department of Electrical and Computer Engineering, Queen s University, Kingston, ON, Canada, in 000, 003, and 008, respectively. He is currently an Assistant Professor with the School of Engineering, The University of British Columbia, Kelowna, BC, Canada. His industrial experience includes positions with Ford Motor Company, Windsor, ON, Canada, and with Astec Advanced Power Systems, Nepean, ON, Canada. He is the holder of one U.S. patent. His current research interests include highefficiency high-power-density dc dc converters and ac dc power factor correction circuits. He is the author or coauthor of more than 40 technical papers published in various IEEE international conference proceedings and IEEE journals. Dr. Eberle was the recipient of the Ontario Graduate Scholarship and has been the recipient of awards from the Power Sources Manufacturers Association and the Ontario Centres of Excellence. He currently holds research grants from the Natural Sciences and Engineering Research Council of Canada, the Canada Foundation for Innovation, the University of British Columbia, and the Kaiser Foundation for Higher Education. William G. Dunford (S 78 M 81 SM 9) received the B.S. degree from Imperial College London, London, U.K., and the Ph.D. degree from the University of ronto, ronto, ON, Canada. He has been a Faculty Member with Imperial College London and the University of ronto and is currently a Faculty and Senate Member with the University of British Columbia, Vancouver, Canada. Industrial experience includes positions with the Royal Aircraft Establishment (now Qinetiq), Schlumberger, and Alcatel. He has had a long-term interest in photovoltaic powered systems and is also involved in projects in the automotive and distributed systems areas. Dr. Dunford has served in various positions on the Advisory Committee of the IEEE Power Electronics Society and chaired IEEE PESC in 1986 and 001.
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