NEW ACTIVE POWER FILTER WITH SIMPLE LOW COST STRUCTURE WITHOUT TLJNED FILTERS

NEW ACTIVE POWER FILTER WITH SIMPLE LOW COST STRUCTURE WITHOUT TLJNED FILTERS Gu H. Jung* and Gyu H. Cho *Dept. of Electrical Engineering, Korea Advanced Institute of Science and Technology(:KAIST), 373-1, Kusong Dong, Yusong Gu, Taejon, 305-701, Korea TEL : +82-42-869-3424, FAX : +82-42-869-3410, E-mail : s-ghjung@cais.kaist.ac.kr Abstract - This paper presents a new active power filter connected in series with a bank of capacitors having no tuned filters and no resonance problem related with harmonic loads. Capacitor of appropriate small size is determined considering much small power capacity of converter compared with load power. Through frequency analysis of system, dual loop controller is designed for eliminating harmonics of load current and reducing fundamental voltage component of converter. Finally, the performance of the presented scheme is confirmed using simulation with lokva 6-pulse load and experimental results with 1 KVA 6-pulse load. I. INTRODUCTION Recently, as harmonic-producing loads such as diode or thyristor converters and cycloconverters increase, the problem of EMI(e1ectro-magnetic interference) becomes more serious in power systems. In this trend, the IEEE-5 19-1992 specifies limits for maximum distortion in the current drawn by individual loads. In the meantime, active filter was presented as one of various approaches for reducing utility current distortion in 1976 and thus has been successfully applied to nonlinear harmonic producing loads for minimizing power quality problems [ 11. From old times, shunt passive filters consisting of a bank of tuned LC filters have been broadly used to suppress harmonics because of its low initial cost and high efficiency. However, shunt passive filters have some problems between the source impedance and shunt passive filter, i.e., parallel resonance caused by harmonic load current and series resonance by harmonic source voltage. Therefore, a seriesactive hybrid filter consisting of series active filter and a set of tuned passive filters connected across the load was presented in 1990's, where the active filter solved the above problems of the conventional passive filter with small VA capacity [2]. After that, numerous topologies and control methods of hybrid filters have been presented for smaller VA rating of active filter in high power applications [3]-[5]. However, these hybrid filters have active filters connected in shunt or series with bulky tuned filters, from which series and parallel resonance caused by source or load can occur in other frequencies except the characteristic ones(5", 7" etc.) because of neighboring multiple loads, low quality factor Q, variation of Q values due to aging of elements. In addition, the complicated control methods in them are used to solve these problems. Therefore, this paper presents a new active power filter connected in series with a bank of capacitors of simple low cost structure, which has no resonance problem related with harmonic loads and simple dual loop control structure because there are no tuned filters. Specially, this topology is suitable for high power applications and also useful for single phase utilities, covering multiple harmonic-producing loads. Also, the usefulness of the presented topology is verified by simulation using 10KVA. 6-pulse diode load and experiment using lkva load. Gate Signal Generator Controller Fig.1. An overall view of the presented active filter without tuned filter. 0-7803-4489-8/98/$10.00 0 1998 IEEE 217

'Mh 11. PRINCIPLE OF OPERATION Fig.1 shows the presented active power filter using twolevel voltage source inverter(vs1) coupled with a transformer connected in series with a bank of capacitors CF, which is connected in parallel with harmoni>,c load. Here, inductance L, is used as a filter for reducing harmonic current in inverter and capacitor C, for eliminating switching frequency harmonics of znd side transformer voltage VM,&. Meanwhile, resistor R, connected in series with C, is inserted for damping parallel resonance between the source impedancels and c,, where both of resistor R, and capacitor C, can be canceled in case of low switching frequency components existing in vm,abc, As it is shown, the distinctive feature of this topology is that only three capacitors instead of multiple number of tuned filters are used in series with active filter compared to conventional hybrid filter. The role of the capacitors is to reduce the rating of the active filter to a small value(about 5% for 6-pulse and 1.25% for 12-pulse load rating). Bulky inductors and tuning problems do not exist in this topology unlike the conventional ones. The compensation of harmonic components generated by load is achieved by dual loop controller using SPWM technique, which makes the source current sine wave with no harmonics and removes fundamental component of v M,abc. The converter connected in series with capacitor banks through transformer operates as a dependent voltage source in the fundamental and harmonic frequencies, which supplies compensation current ic,abc to draw harmonic currents from source line. As this scheme uses no shunt tuned filters for absorbing harmonic current of load, series and parallel resonance do not occur between the source impedance and the shunt passive filters[2]. As shown in Fig. I, three phase reference harmonic current i&abc is desired as reference signal for dual loop controller, which is obtained from the three-phase terminal voltage vt.abc and load current il,abc by the following instantaneous dq-transformation theory[3]. Firstly, these variables on abc-axis are transformed into dq-axis ones using transformation matrix K, from which instantaneous real and imaginary power pl and ql of load are obtained as follows. the desired three phase reference harmonic current i:h,abc are extracted by inverse transformation matrix K-' as follows: (3) Using these current references, dual loop controller produces three phase reference voltage v,,~,~,,~ for converter which is applied to SPWM gate signal generator with lokhz carrier frequency. Fig. 2. Simplified per-phase equivalent circuit of the presented filter: (a) for the fundamental frequency, (b) for the harmonic frequencies. Fig. 2 shows simplified per-phase fundamental and harmonic equivalent circuits of the proposed filter, where the converter has a role of dependent voltage sources. At the fundamental frequency of Fig. 2(a), the transformer znd voltage of converter is operated to be short ( VM, = 0) and source current is, to be sum of capacitor current i, and load current i,,. At the harmonic frequencies of Fig. 2(b), the converter supplies compensation harmonic current i, (= i, ) to load and then the harmonic voltage of vmh is made equal to the harmonic voltage drop in the compensation capacitor C,. Also, the source harmonic current is, becomes zero with equivalently shorted terminal harmonic voltage( 'Th = 0) and no source impedance appears in harmonic frequencies. Hence, no series and parallel resonance occur between source impedance Z, and filter impedance Z,, which is effective in compensating harmonics of non-characteristic frequencies generated by neighboring multiple loads and variation of quality factor Q. Additionally, the equations related to is, i,, i, and vm are expressed as follows: (4) is = is, + is, = i, - i,,, i, = i,, + i,, (5) ic = ic, + t,,, VM = vml + ZFiCh ' (6) Considering filtering characteristics for the load harmonic current i, using impedance concept given in Fig. 2(b), source harmonic current is, generated by i, can be expressed as the following equation: Secondly, instantaneous harmonic powers PLh and qlh of load are obtained through two high pass filters using 4-th order low pass filters with cutoff frequency of 30Hz, and thus 218

Assuming that sum of converter impedance Z, and filter impedance Z, is controlled to become zero in (7) by complete closed loop controller, that is, no source harmonic current exists( is, =O). Voltage and Current of Converter 6 15 0 100 200 300 400 500 Capacitor C, [uf] (4 an acceptable size, C, of 240uF is selected. Meanwhile, with curves similar to Fig. 3, power rating P, of converter in 12- pulse diode load becomes about 1.2% of load power. IV. DUAL ILOOP CONTROL To understand open-loop characteristics of the presented system with frequency variation, a per-phase equivalent circuit of Fig. 1 is shown in Fig. 4. Here, assuming that turn ratio of transformer is 1: n and output voltage of inverter is vi, equivalent output voltage and filter inductance seen on 2nd side of transformer become n vi and L,, (= n' L,,,). In the meantime, harmonic load impedance as current source i, is thought to be open, and there are no source harmonic current (is, =0) if compensation current i, equals to i, in harmonic frequencies. Based on these facts, transfer function Gp(s) of i, with respect to vi is obtained as follows: 5 Power Rating of Capacitor & Converter I ' 11 where WO = 2 j 0 = - 1 fin11 + LS)cF ' a" 0 100 200 300 400 500 Capacitor C, [uf] (b) Fig. 3. The converter characteristics with variation of C, in 10kVA[1 p.u.1 6-pulse diode rectifier load, (a) compensation rms current 1,, compensation harmonic rms current I,,, converter rms voltage V,, (b) rated power PM [%] of converter, Pc [P.u.] of capacitor. 111. SELECTION OF CAPACITOR C, Voltage, current and power rating of converter are plotted in Fig. 3 by varying size of filter capacitor C, for lokva 6- pulse diode rectifier load. As it is shown, the power rating of the converter depends on the value of the capacitor C, used in this topology. That is, as the size of C, goes larger, harmonic compensation current I,, is fixed with nearly constant value but the fundamental compensation current I,, increases in proportion with C,. As a result, total compensation rms current I, (= Icl + I,, ) in converter increases together. Meanwhile, as converter voltage V, becomes equal to ripple voltage of capacitor C, caused by I,,, V, becomes smaller with larger C,. Fig. 3(b) shows that the power rating PM of converter approximately becomes about 5%(=0.5kVA) of load power for above loouf and P, of capacitor C, increases linearly. Assuming that Pc is O.S[p.u.](=SkVA) which is thought to be Fig. 4. Per-phase equivalent circuit of the presented filter for frequency ana.lysis. I meaning Table I. System Parameters I I ] sy;bol ~ ; : ; v, fundamental frequency I I I I rated load power [I p.u.1 I VA 10 [kva] I source impedaince I Ls I 100 [uh] converter filter inductor converter filter capacitor damping res i s I: o r transformer turn ratio I gain of ~-control~er I K, I 0.8 219

Bode diagram of G,(s) can be plotted as in Fig. 5(a) using circuit parameter of Table I. At low frequency range below f, (=296Hz), filter inductor L,, is negligible and only the filter capacitor C, exists, which means that most of terminal voltage appears across C, and the fundamental transient response depends on its size. In contrary, at harmonic frequencies above f,, only the filter inductor L,, remains with filter capacitor C, shorted. Thus, L,, of appropriate small size must be selected for the presented filter to have filtering bandwidth enough to eliminate load harmonics. controller, closed loop transfer function G,,(s) of i, with respect to i, can be written as follows: where 0, 10. ioo IO io2 io3 io4 Frequency (radisec) Therefore, to obtain good filtering characteristics of wide bandwidth with low Q, we can see that the size of L,, and L, must be small, while that of turn ratio n, gain K, and dc voltage V,, as large as possible. Ref=O VM IO- ioo IO io2 io3 io4 Frequency (radisec) (a) open-loop transfer function G, (s) VT ICh* Fig. 6. Per-phase block diagram of overall control loop with dual loop structure. -40 I I I oo 1 o2 1 o4 1 o8 Frequency (radisec) Frequency (radlsec) (b) closed-loop transfer function GCL (s) Fig. 5. Bode plot of the presented filter system for frequency analysis. Meanwhile, overall control loop has dual loop structure as illustrated in Fig. 6, in which lower loop with P-controller is for compensate harmonic load current and upper ones with two PI-controller are for reducing the fundamental component of voltage vm. In the lower loop, the fundamental reference current je1 for compensation capacitor C, is derived through differentiator and thus final reference current i for control becomes the sum of i:, and harmonic reference current ich obtained by the instantaneous dq-theory of chapter 11. If the G,(s) given in (8) is applied to lower loop with P- Fig. 5(b) is bode plot of G,,(s) obtained using system parameters of Table I, where region of constant magnitude between 60Hz and 1.5KHz becomes bandwidth of the presented filter. However, error of magnitude and phase existing in ic at the fundamental frequency 60Hz generate some fundamental component in vm, which must not exist in theory and otherwise causes power rating of converter much larger than 5% estimated in Fig. 3(b). Therefore, this paper presents another upper voltage loop for reducing the fundamental component of vm as given in Fig. 6. Here, only peak value of fundamental sine(cosine) component existing in vm is derived through LPF after multiplied by fundamental sine(cosine) source. Using a PI controller and sine(cosine) source, desired reference signal for reducing sine(cosine) component of v,,,, is generated and added to SPWM reference signal v,,~ as shown in Fig. 1. Thus, simulational VA rating of converter approaches the above theoretical estimare(5%) by this additional loop. V. SIMULATION The possibility of the presented scheme is confirmed by simulation results for lokva 6-pulse diode load where parameters of the system is given in Table I. As shown in the waveform of Fig. 7, a-phase source current is, has sine wave with harmonics nearly eliminated from a-phase load current i, and the fundamental component of is, equals the sum of 220

I compensation OKHz the fundamental current of capacitor and load. Also, Fig. 7(a) converter voltage vma : 80 [V/div], source current is, : shows that a-phase converter voltage vma is nearly zero in loo[a/div], load current i, : loo[a/div], filter average dc value and composed of most harmonic compensation current i, : loo[a/div]. components where some fundamental component still exists in vma due to incomplete characteristics of LPF used in upper Ttlk Acqrrps'tions -T 1 - - 4 voltage loop of Fig. 6. In this case, power rating of three phase converter becomes about 8%(0.8KVA) of load power, which is a little larger than the former estimated value(5%) of ideal case because of some fundamental component existing in vma. Also, as illustrated in frequency spectrum of Fig. 7(b), most of 5, 7, 11, 13" harmonics of load current ila is almost eliminated in source current is, with filtering bandwidth of about 1.5KHz mentioned in Fig. 5(b). However, spectrum of vma has some fundamental value which can be reduced to nearly zero by optimal design of upper loop. (a) TeK Slouurd *. 603 Acctiiisitions I.. T. 1 SEL>> OA j +--- (-11 SO OmV CF2700mV 4 M5 OOms Chi J '58mV' (113 'io OmV [4EKl 20 OinVR - - - j -- - - -, (b) 50AT---------~----.-------------- 1 Fig. 8. Experimental results of a-phase with lkva 6- pulse diode load, terminal voltage vta vta : 100[V/div], converter voltage vma : loo[v/div], source current - is, : lo[a/div], load current i, : lo[a/div], filter i current i, : 6.25[A/div]. VI. EXPERIMENTAL RESULTS fl ;EL=-=- Similarity between simulational and experimental results is ov --_- - -- 1 - - - - - verified through experiment of reduced 1 KVA rated 6-pulse OHz 1. OKHz 2 ~ diode load. As shown in Fig. 8(a), source current is, has sine Frequency waveform with harmonics eliminated from load current i, whose harmonics was not completely removed due to AD (b) frequency spectrum converter with s!ow conversion time of 140 usec for sampling Fig. 7. Simulational results of a-phase with lokva 6- signals and slow switching frequency of 5kHz. Fig. 8(b) pulse diode load, terminal voltage vta : 2OO[V/div], shows that converter voltage vma has harmonic and some 221

fundamental component, where upper voltage compensation loop used in the simulation was not applied in the experiment for simple control. Additionally, applying faster AD converter, higher switching frequency and upper voltage loop to experiment, it is thought that the experimental results similar to simulation ones can be obtained. VII. CONCLUSION This paper presents a new active power filter with simple low cost structure which has no multiple tuned filters and no resonance problems related with harmonic loads. Filter capacitor of appropriate size was selected considering power rating of converter related to various filter size. Thus, dual loop controller is designed based on frequency analysis using open and closed loop transfer function. Finally, the usability of the presented topology is confirmed by simulation with lokva 6-pulse diode load and experimental results with lkva load. REFERENCES [1] L. Gyugyi and E. C. Strycula, Active AC Power Filters, IEEELAS Annual Meeting, pp. 529-535, 1976. [2] F. Z. Peng, H. Akagi, and A. Nabae, A New Approach to Harmonic Compensation in Power Systems - A Combined System of Shunt Passive and Series Active Filters, IEEE Trans. Ind. Appl., vol. 26, no. 6, pp. 983-990, Nov./Dec., 1990.. [3] H. Fujita and H. Akagi, A Practical Approach to Harmonic Compensation in Power Systems - Series Connec tion of Passive and Active Filters, IEEE Trans. Ind. Appl., vol. 27, no. 6, pp. 1020-1025, Nov./Dec. [4] 1991. M. Rastogi, N. Mohan, A. Edris, Filtering of Harmonic Currents and Damping of Resonances in Power Systems with a Hybrid-Active Filter, IEEEAPEC Annual Meeting, pp. 607-612, 1995. [5] A. van Zyl, J. H. R. Enslin, W. H. Steyn, A New Unified Approach to Power Quality Management, IEEE/PESC Annual Meeting, pp. 183-188, 1995. 222