A Current-Source Active Power Filter with a New DC Filter Structure Mika Salo Department of Electrical Engineering, Institute of Power Electronics Tampere University of Technology P.O.Box 692, FIN-3311 Tampere, Finland Abstract- The main drawback of the current-source active power filter is the heavy and bulky dc side filter. The large dc filter is needed to store the energy of the compensated harmonic components. In this paper a new smaller dc filter structure is proposed for the current-source active power filter. In the presented dc filter structure the energy of the most important harmonics are stored in resonant circuit which makes it possible to decrease the overall size of the filter. The function of the proposed dc filter structure is examined with both simulations and experimental tests. Power supply u s i lda i ta Load L f isa L s Supply filter i ra T 1 Rectifier bridge T 3 T dc-filter i dc L dc I. INTRODUCTION In recent years, active power filters have been widely investigated for the compensation of harmonic currents in electrical power systems. These active power filters are divided into two types: voltage-source active filter (VSAF) and current-source active filter (CSAF). CSAF has advantages of excellent current control capability, easy protection and high reliability over VSAF [1]. The main drawbacks of the CSAF has been so far the lag of proper switching devices and large dc side filter. The new IGBTs with reverse blocking capability are being launched on the markets which are suitable for CSAF [2]. However, the bulky and heavy dc side filter is still a problem. Fig. 1 shows the most common main circuit structure of the current-source active power filter (e.g. [3]-[]). The line current characteristics are improved by injecting the current components opposite to the harmonics of the load current. The energy of the injected harmonic components is stored in and restored from the dc circuit which makes ripple in the dc current. In order to keep this ripple in an acceptable level relatively large dc filter inductor is needed. In this paper a new smaller dc filter structure is presented for the current-source active power filter. The proposed dc filter structure is shown in Fig. 1. In the presented dc filter the energy of the most important harmonics are stored in resonant circuit which makes it possible to decrease the overall size of the filter. II. PROPOSED DC FILTER STRUCTURE The most important harmonics in typical nonlinear loads are th and 7th harmonic components which produce 6th harmonic component in the dc circuit of the active power filter. Next important harmonic component in the dc circuit of the CSAF is the 12th harmonic which is caused by the compensation of 11th and Power i lda supply i ta u s 13th load current harmonics. Other components in the dc circuit have order of 18, 24, 3 etc. However, the 6th harmonic components makes the biggest ripple in the dc current because of it s large magnitude and low frequency. The proposed dc filter structure shown in Fig. 1 is designed to damp the 6th harmonic component effectively. The proposed dc filter structure uses a parallel resonant circuit which is tuned for 6th harmonic component. Other harmonics of the dc circuit are filtered with the inductor L dc connected in series with the resonant circuit. With the proposed modified dc filter structure the amount of the total filter inductance can be significantly reduced. Impedance of the conventional dc filter structure is R dc Load L f isa L s Supply filter Fig. 1. The current source active power filter with conventional dc filter structure and with modified dc filter structure. Zs ( ) = R dc sl dc where is the resistance of the dc filter. For modified dc filter structure can be written as L r s R Zs ( ) = r R dc sl dc --------------------------------------- L r C r s 2 R r C r s 1 C s C s i ra T 1 T 2 T 2 T 3 T 4 T 4 Rectifier bridge T T 6 T 6 i dc L r dc filter L dc i cr C r (1) (2)
where L r, C r and R r are the inductance, capacitance and resistance of the parallel resonant circuit respectively. Impedances of the conventional (--) and modified (-) dc filter structures are plotted in Fig. 2 as a function of the angular frequency. With conventional dc filter L dc =17 mh and R dc =8Ω. The modified dc filter have parameter values: L dc =3 mh, R dc =3Ω, L r =1 mh, R r = 2Ω and C r =18.7 µf. The resistances are approximate values at 3 Hz. Fig. 2 shows also the impedance curve with 4 mh dc filter inductor (.-) which is the total inductance of the modified dc filter structure. These parameter values are used with the active power filter of which nominal power is kw. Fig. 2 shows that impedance of the modified dc filter is increased rapidly at 19 rad/s (3 Hz). This is caused by the parallel resonant circuit which is tuned for this frequency (6 th harmonic component). At this frequency the impedance of the modified dc filter is higher than the impedance of the conventional filter which indicates that the proposed filter structure can store effectively the energy of the filtered th and 7th load current harmonics without large ripple in dc current. Fig. 2 shows also that the impedance of the modified filter is very low at 27 rad/s (42 Hz). This is caused by the series resonance of the proposed dc filter structure. The frequency of the series resonance is determined by C r and the parallel connection of L dc and L r. When selecting the parameters for the modified dc filter structure it should be made sure that the frequency of the series resonance is not near to 12th harmonic component which is the next important harmonic component in the dc circuit of the active filter. Otherwise, the series resonance should neither be near 6th harmonic component. This can be avoided when the ratio of L r and is between 1/2-2. L dc III. CONTROL OF CSAF The proposed dc filter topology can be used with any control system of CSAF. Fig. 3 shows a control system [6] which is practical for current-source topology. It is realized in the synchronously rotating reference frame where the active power of the active filter can be simply controlled with i sx i sy real axis component and the reactive power with imaginary axis component of the filter current. The superscript s in space vector variables and x/y in space vector components refers to a synchronously rotating coordinate system. The harmonic compensation is based on the feedforward control of i sxy the load currents. The active filter currents are controlled in an open-loop manner. The reference values for active filter current vector are calculated as follows: i sx = i ffx i dcx (3) Impedance[dB] and 8 6 4 2 1 2 1 3 1 4 Angular frequency [rad/s] Fig. 2. Impedances of the convential dc filter when L dc =17 mh (--) and L dc =4 mh (.-) and impedance of the modified dc filter (-) as function of the angular frequency. i sy = i ffy i qy (4) where i ffxy (both components combined in one expression), i and i dcx qy are outputs of the feedforward, dc current and the reactive power controls respectively. These two components form the rectifier current reference vector i s r which is transformed to the stationary reference frame and fed to the modulator. Due to the open-loop control of the active filter currents the currents references are not realized accurately because the supply filter takes capacitive currents. Also, oscillations may occur in supply currents due to the LC-filter resonance if the active filter current references are rapidly changed. Detailed describtion of the control methods to solve both of these problems can be found in [6]. A. Dc current control The task of the active filter is to compensate the harmonics of the non-linear load. The magnitude of the dc current is changed as the energy of the harmonic components is stored in and restored from the dc circuit. This ripple in the dc current is the basic feature in the active power filter and for that reason the dc current control should not try to remove it. However, the dc current control should work effectively when the reference value of the dc current is changed. For that reason, a non-linear PID controller, where the input of the controller is the square of the error signal, is proposed for the control system shown in Fig. 3. With small error values the controller acts slowly and when the error value is increased faster control dynamics is achieved. Fig. 4 shows the block diagram of the non-linear PI controller where the modified proportional gain P dc k 1 depends on the error value. In practice, it is reasonable to limit P dc k 1 between and max P dc, which is done with Saturation block. To understand how i dcx is constructed in the dc-current control we can first consider that 3 --u 2 sx i dcx = u dcbr i dc ()
Load L f i lda Power CSAF supply i i i dc ta sa i ra L dc u dcbr Lr Cr 3->2 i ld e jθ s i ld s i ldx i ldy HPF HPF ^ i ^ ldx i ldy -1-1 CDC θ CDC s u s Reactive power control i qy L s i ffy C s θ s i sy Modulator e jθ s i r i sx i ffx FG i r s = is s i dc i dc PID(e 2 ) u dcbr c 1 i dcx Feedforward control Fig. 3. Control system of CSAF with modified dc filter topology. Dc current control i.e. that the ac and dc active powers of the converter are equal in steady state if the converter losses are ignored. u dcbr is the dc-voltage of the rectifier bridge. By solving () for i dcx and by using the reference values of i dcx and u dcbr we have = ----------u dcbr i dc = 2 i dcx 3u sx cu dcbr i dc which is used in Fig. 3 to transform the dc-voltage reference of the rectifier bridge to vector variable. IV. SIMULATION RESULTS The proposed control methods are tested with the simulation model. The simulation model is built in discrete form to have close analogy with the microcontroller implementation. Simulation is based on the control system shown in Fig. 3. Sampling time of the feedforward and dc current controllers is µs and the modulation frequency 1 khz. The supply filter is realized with parameters: L s =.6 mh and C s =8 µf. Figs. and 6 show the simulation results of CSAF with modified and conventional dc filter structure respectively. As a nonlinear load a three-phase diode rectifier with RL-load is used. The parameters of the modified dc filter are L dc =1 mh, L r =1 mh and C r =28.1 µf and conventional L dc =1 mh. (6), k1 k1 i dc i dc abs P dc P k1 dc Saturation T s T dc Delay Delay Saturation T s In both simulations the the ripple of the dc current is about 1 A. The total harmonic distortion (THD) of the load current in both simulations is 26.8%. THDs of the supply current with modified dc filter is 4.2% and with conventional filter 4.%. Fig. 6 shows the current i cr of the resonant circuit capacitor. It s amplitude is about 3A and frequency 3 Hz. This ac current is caused by the 3 Hz ac voltage component across the resonant circuit. This 3 Hz ac voltage component is caused by the compensation of the th and 7th load current harmonics as was explained earlier. Figs. 7 and 8 show the simulation results of CSAF with D dc Fig. 4. Non-linear PID controller for dc current control. Saturation k u, 1 dcbr
1 1 1 1 - - -1-1 -1.2.4.6-1.2.4.6 1 1 1 1 -.2.4.6-1.2.4.6 Fig.. Simulation results of CSAF with modified dc filter structure when the current harmonics are produced using a three-phase diode rectifier with RL-load. Load current, supply current, dc current and current of the resonant circuit. i lda i ta i dc i cr -1 1 1 - -1-1.2.4.6 1 1 1 - -1-1.2.4.6 1.2.4.6 Fig. 6. Simulation results of CSAF with conventional dc filter structure when the current harmonics are produced using a three-phase diode rectifier with RL-load. Load current, supply current and dc current. i lda i ta i dc modified and conventional dc filter structure respectively when as a non-linear load a three-phase diode rectifier with RC-load is used. In this case the parameters of the modified dc filter are L dc =3 mh, L r =1 mh and C r =18.7 µf and conventional L dc =17 mh. The size of L dc is increased in both modified and conventional dc filter solutions because RC-type diode load contains more harmonics than RL-type load. Also, the size of L r is increased in order to keep the series resonance of the modified filter far enough from the parallel resonance. Furthermore, smaller C r decreases the current of the resonant circuit capacitor. Anyway, Figs. and 7 shows that i cr is much larger with RC-type load than RL-type load due to the larger amount of harmonics included in RC-load which increases also the harmonics of the dc circuit. THD of is i lda
1 1 1 1 - - -1-1 -1.2.4.6-1.2.4.6 1 1 1 1 - -1.2.4.6-1.2.4.6 Fig. 7. Simulation results of CSAF with modified dc filter structure when the current harmonics are produced using a three-phase diode rectifier with RC-load. Load current, supply current, dc current and current of the resonant circuit. i lda i ta i dc i cr 1 1 - -1-1.2.4.6 1 1 1 - -1-1.2.4.6 1.2.4.6 Fig. 8. Simulation results of CSAF with conventionald dc filter structure when the current harmonics are produced using a three-phase diode rectifier with RC-load. Load current i lda, supply current i ta and dc current i dc. 87.9% and THDs of i ta with modified and conventional dc filter 7.3% and 7.6% respectively. V. EXPERIMENTAL INVESTIGATION The prototype of CSAF is built using 12 V, A IGBTs. The control system realization is based on the Motorola MPC 32-bit single-chip microcontroller. The supply filter parameters and the sampling times of the control system are the same as used in simulation model. Figs. 9 and 1 show the experimental results of CSAF with modified and conventional dc filter structure respectively. As a non-linear load a three-phase diode rectifier with RL-load is used. The dc filter parameters are same as used in simulations.
1 1 1 1 - - -1-1 -1.2.4.6-1.2.4.6 1 1 1 1 - -1-1.2.4.6.2.4.6 Fig. 9. Experimental results of CSAF with modified dc filter structure when the current harmonics are produced using a three-phase diode rectifier with RL-load. Load current i lda, supply current i ta, dc current i dc and current of the resonant circuit i cr. 1 1 - -1-1.2.4.6 1 1 1 - -1-1.2.4.6 1.2.4.6 Fig. 1. Experimental results of CSAF with conventional dc filter structure when the current harmonics are produced using a three-phase diode rectifier with RL-load. Load current i lda, supply current i ta and dc current i dc. By comparing Figs., 6, 9 and 1 it can be seen that the simulation and experimental results are in good agreement. THD of i lda in Figs. 9 and 1 is 27.1% and THDs of i ta with modified and conventional dc filter 3.3% and 3.2% respectively. Figs. 11 and 12 show the experimental results of CSAF with modified dc filter in two cases when RC-type diode rec- tifier load is used. In the first case shown in Fig. 11 the load current contains only 1.3% of 3rd harmonic current. In the case of Fig. 12 the amount of 3rd harmonic is 7.1%. The 3rd harmonic component in load currents causes 2nd harmonic voltage component in the dc circuit. However, the impedance of the modified dc filter for 2nd harmonic component is very low as can be seen in Fig. 2. As a result, the ripple in dc cur-
1 1 - -1-1.2.4.6 1 1 1 1 - -1-1.2.4.6 1 1 - -1-1.2.4.6.2.4.6 Fig. 11. Experimental results of CSAF with modified dc filter structure when the current harmonics are produced using a three-phase diode rectifier with RC-load. Load current contains only small amount of 3rd harmonic component. Load current i lda, supply current i ta, dc current i dc and current of the resonant circuit i cr. 1 1 1 1 - - -1-1.2.4.6 1 1-1 -1.2.4.6 1 1 - -1.2.4.6.2.4.6 Fig. 12. Experimental results of CSAF with modified dc filter structure when the current harmonics are produced using a three-phase diode rectifier with RC-load. Load current contains large amount of 3rd harmonic component. Load current i lda, supply current i ta, dc current i dc and current of the resonant circuit i cr. -1 rent is much larger in Fig. 11 than in 12 due to the significant 1 Hz component (2nd harmonic). In principle, symmetric three-phase load should not contain 3rd harmonic component which was also confirmed with simulations. The 3rd harmonic components seen in measured load current is caused propably by the distorted supply voltages. The experimental results of the conventional dc filter are shown in Fig. 13. In Figs. 11-13 the THD of i lda is around 82%. The THDs of i ta shown in Figs. 11, 12 and 13 are.7%, 7.% and 6.%. According to simulations and experimental investigation it can be concluded that the proposed dc filter structure works well if the load currents are symmetrical and do not contain
1 1 1 1 - - -1-1 -1.2.4.6-1.2.4.6 1 1.2.4.6 Fig. 13. Experimental results of CSAF with conventional dc filter structure when the current harmonics are produced using a three-phase diode rectifier with RC-load. Load current i lda, supply current i ta and dc current i dc. 3rd harmonic component. It seems that in practice threephase diode rectifier with RC-type load generates 3rd harmonic in load currents and is not practical for proposed filter structure. However, the modified filter can be used if the 3rd harmonic component is not compensated. In the case of RLtype diode rectifier load the amount of 3rd harmonic is minimal and the proposed filter structure can be used. In this case the amount of total inductance needed in the modified dc filter is decreased to 1/4 compared to the conventional dc filter. VI. CONCLUSIONS In this paper a new smaller dc filter structure is proposed for the current-source active power filter. In the presented dc filter structure the energy of the most important harmonics are stored in resonant circuit which makes it possible to decrease the overall size of the filter. After simulations and experimental tests it was found that the proposed filter structure works well with symmetrical loads if the load current doesn t contain 3rd harmonic component. REFERENCES [1] Y. Hayashi, N. Sato and K. Takahashi, A novel control of a currentsource active filter for ac power system harmonic compensation, IEEE Trans. Ind. App., Vol. 27, No. 2, pp. 38-38, March/April 1997. [2] A. Lindemann, Characteristics and applications of a reverse blocking IGBT, PCIM Europe, pp.12-16, January-February 21. [3] S. Fukuda and T. Endoh, Control method and characteristics of active power filters, th European Conference on Power Electronics and Applications, Vol 8, pp. 139-144, 1993. [4] S. Fukuda and T. Endoh, Control method for a combined active filter system employing a current source converter and a high pass filter, IEEE Trans. Ind. App., Vol. 31, No. 3, pp. 9-97, 199. [] M.-X W ang and H. Pouliquen, Performance of an active filter using PWM current source inverter, th European Conference on Power Electronics and Applications, Vol. 8, pp. 218-223, 1993. [6] M. Salo and H. Tuusa, H., A novel open-loop control method for a current-source active power filter, IEEE Trans. Ind. Electr., Vol., No. 2, pp. 313-321, 23.