IN recent years, the development of high power isolated bidirectional

IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 23, NO. 2, MARCH 2008 813 A ZVS Bidirectional DC DC Converter With Phase-Shift Plus PWM Control Scheme Huafeng Xiao and Shaojun Xie, Member, IEEE Abstract The current-voltage-fed bidirectional dc dc converter, which refers to a current-fed inverter at low voltage side and a voltage-fed inverter at high voltage side, can realize zero voltage switching (ZVS) for the switches with the use of phase-shift (PS) technology. However, the current-fed switches suffer from high voltage spike and high circulating conduction loss. In order to solve these problems, a novel phase-shift plus pulsewidth modulation (PSP) control ZVS bidirectional dc dc converter is proposed in this paper. By adopting active clamping branch and PSP technology, the converter can realize ZVS for all switches in a wide range of load variation while input or output voltage varies. In addition, a novel control strategy with one port voltage regulation and another port current regulation is proposed to make energy bidirectional conversion freely. The operation principle is analyzed and verified by a 28V/270V conversion prototype rated at 1.5kW. Index Terms Active clamping, bidirectional dc dc converter, phase-shift plus pulsewidth modulation (PSP), pulsewidth modulation (PWM), zero voltage switching (ZVS). s NOMENCLATURE Voltage of the converter port. Voltage of the converter port. Current of the converter port. Inductance in the converter side. Instantaneous current through inductance, and, respectively. Total effective inductance in series with the winding of transformer. Instantaneous current through inductance. Clamping capacitor. Instantaneous voltage across clamping capacitor c. Number of turns of primary winding, and secondary winding, respectively. Capacitor in the converter side. Phase-shift angle between and. Duty cycle of the switches and. Angular frequency. Switching frequency. Conversion efficiency. Manuscript received March 1, 2007; revised July 26, 2007. Recommended for publication by Associate Editor C. Canesin. The authors are with the College of Automation Engineering, Nanjing University of Aeronautics and Astronautics, Nanjing 210016, China (e-mails: saloulin@ynet.com, or xiaohf@nuaa.edu.cn; eeac@nuaa.edu.cn). Digital Object Identifier 10.1109/TPEL.2007.915188 Output power. Effective value of the current. Instantaneous voltage across the primary winding of transformer. Instantaneous voltage across the secondary winding of transformer. Output of the phase-shift angle controller. Output of the duty cycle controller. I. INTRODUCTION IN recent years, the development of high power isolated bidirectional dc dc converters (BDC) has become an important topic because of the requirements of electric vehicle, uninterruptible power supply (UPS), distributed generation, energy storage, and aviation power system [1] [12]. In a typical UPS system, the battery is charged when the main power source is normal and discharges to supply power in case of the failure of lose of the main power source. In the aircraft high voltage direct current (HVDC) power supply system [6], when the 270 V HVDC generator is in gear, it charges the 28 V battery and supplies the 28 V key load by the BDC, and when the generator is in failure, the 28 V battery discharges to supply 270 V key load by the BDC. The high-low voltage conversion and electrical isolation are necessary in the above-mentioned conditions. The current-voltage-fed BDC is suitable for such system due to its high voltage conversion ratio and low current ripple in the current-fed port. A dual active full bridge dc dc converter was proposed for the high power BDC in [9] and [10], which employs two voltage-fed inverters to drive each side of a transformer. Its symmetric structure enables the bidirectional power flow and ZVS for all switches. A dual active half bridge current-voltage-fed soft-switching bidirectional dc dc converter was proposed with reduced power components [11]. However, the current stresses in switches and are asymmetric. When the voltage amplitude of two sides of the transformer is not matched, the current stress and circulating conduction loss become higher in [9], [10], and [11]. In addition, these converters can not achieve ZVS in a wide range of load variation while input or output voltage varies. These disadvantages make them not suitable for large variation of input or output voltage condition. An asymmetric bidirectional dc dc converter with PWM plus Phase shift (PPS) control was proposed in [12]. The circulating conduction loss is reduced, however, it results the asymmetric stresses in the main switches and a bias of the magnetizing current which decreases the utilization of the transformer. So, it is not suitable for high power bidirectional conversion. 0885-8993/$25.00 2008 IEEE

814 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 23, NO. 2, MARCH 2008 Fig. 1. Novel PSP ZVS BDC (a) Main circuit. (b) Key waveforms in Boost mode. (c) Key waveforms in Buck mode. It is proposed that a current-voltage-fed PSP ZVS BDC based on a current-fed half bridge and a voltage-fed half bridge guaranteeing volt-second balance of the transformer by its capacitors and in this paper, as shown in Fig. 1(a). The converter utilizes an active clamping branch and to avoid the voltage spike, achieve ZVS of and, and also restrain the start-inrush current [13]. By PWM control of and, the amplitude of and is well matched while input or output voltage varies, which can reduce circulating conduction loss, and realizes ZVS in a wide range of load variation. The control strategy of Phase-shift (PS) plus PWM is realized by two individual controllers. The operation principle of PSP ZVS BDC is analyzed in detail. A 22 32 V/270 V 1.5 kw prototype is built to verify the operation principle of the proposed converter. II. OPERATION PRINCIPLE The BDC has two operation modes. It is defined as Boost mode when energy flowing from side to side, and the counterpart is defined as Buck mode. Before analysis, the following assumptions are given: 1) All the active power devices are ideal switches with parallel body diodes ( and ) and parasitic capacitors (, and ); 2) The inductance and are large enough to be treated as two current sources with value of ; 3) The transformer is an ideal one with series leakage inductor. Fig. 1(b) shows the key waveforms in Boost mode. One complete switching cycle can be divided into twelve stages. Because of the similarity, only a half switching

XIAO AND XIE: ZVS BDC WITH PHASE-SHIFT PLUS PWM CONTROL SCHEME 815 Fig. 2. Equivalent circuits in Boost mode for a half switching period (a) Stage 0 [before ] (b) Stage 1[ ] (c) Stage 2[ ] (d) Stage 3[ ] (e) Stage 4[ ] (f) Stage 5[ ] (g) Stage 6[ ]. cycle is described in detail. The equivalent circuits are shown in Fig. 2. As the two sides of the topology are symmetrical, the operation principles in Buck mode are similar to those in Boost mode. Fig. 1(c) shows the key waveforms in Buck mode. 1) Stage 0 [Before ]: Refer to Fig. 2(a). and are conducting. At this stage,. The power flows from side to side. 2) Stage 1 : Refer to Fig. 2(b). At is turned off. and begin to resonate, is discharged and is charged. 3) Stage 2 : Refer to Fig. 2(c). At, the voltage across attempts to overshoot the negative rail. is therefore forward biased. During this period, can be turned on under zero voltage. The voltage across is clamped at. At this stage,. 4) Stage 3 : Refer to Fig. 2(d). At is turned off. and begin to resonate, is charged, is discharged. At this stage,. 5) Stage 4 : Refer to Fig. 2(e). At, the voltage across attempts to overshoot the negative rail. is therefore forward biased. During this period, can be turned on under zero voltage. The voltage across is clamped at. The current of rises to a positive value.

816 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 23, NO. 2, MARCH 2008 6) Stage 5 : Refer to Fig. 2(f). At is turned off. and begin to resonate, is discharged and is charged. 7) Stage 6 : Refer to Fig. 2(g). At, the voltage across attempts to overshoot the negative rail. is therefore forward biased. During this period, can be turned on under zero voltage. The voltage across is clamped at. At this stage,. The power flows from side to side. At, the second half cycle starts, which is similar to the first half cycle. III. CHARACTERISTICS OF THE NOVEL BDC A. Output Power The phase shift angle between and [referring to Fig. 1(b) and (c)], which is defined to be positive when is leading to in phase, is used to control the magnitude and direction of the transmitted power. The duty cycle of and is used to match the amplitude of and, that means the current keeps constant in stage 0 and stage 6. Referring to the Appendix A, the duty cycle of and is given by (1) Fig. 3. Curves of the normalized output power versus the phase-shift angle (V =22 32 V;V =270V;N =N =2:1). B. Circulating Current When the transmitted power is, the current RMS of under PS control in Boost mode is (Referring to the Appendix C) Under PS control, the output power is [10] Under PSP control, the output power is (Referring to the Appendix B) P = (2) 2(N V ) (10d)(jj+d01:5) ; [0;02(1 0 d)] (2N )!L (N V ) [ +2(10d)0(10d)(2d01) ] ; [02(1 0 d); 0] (2N )!L 2(N V ) (10d)[0(d00:5)] ; [0; (2d 0 1)] (2N )!L (N V ) [0 +2d0d(2d01) ] ; [(2d 0 1); ]: (2N )!L (3) Fig. 3 shows the relations between the output power (normalized by ) and phase-shift angle under PS and PSP control. The bold curves are output power versus under PSP control. The intersection curves are output power versus under PS control. When the amplitude of and is matching ( V, V), the both curves are superposed under PS control and PSP control. However, the maximum of output power under PSP control is higher than that of PS control in low battery voltage. Evidently, PSP control improves the capability of power transmission. where is equal to 2, and is equal to. Under PSP control, the current RMS of in Boost mode is (Referring to the Appendix D) as (5), shown at the bottom of the page. Fig. 4 shows the comparison of the current RMS of under PS control and PSP control in Boost mode. It is evident that the circulating current is low under PSP control, which can improve the conversion efficiency in low battery voltage. C. Range for Achieving Soft Switching From Section II, it can be known that in order to achieve ZVS for all switches, (6) should be satisfied in Boost mode. (4) (6) (5)

XIAO AND XIE: ZVS BDC WITH PHASE-SHIFT PLUS PWM CONTROL SCHEME 817 Fig. 4. RMS value of i.(v = 22 32 V;V = 270 V;N : N = 2:1;P =1:5 kw, f = 100 khz, L =1:2 H). Also, (7) should be satisfied in Buck mode. The conventional dual active bridge converter with PS control scheme can achieve full control range under soft switching while the amplitude matching of and is naturally matching. However, when the amplitude of and is not matching, the soft switching range is rapidly reduced [10]. By adopting PWM control of and in this paper, the amplitude matching of and is completely guaranteed in different battery voltage. Therefore, this converter can satisfy (6) or (7) well from no load to full load under PSP control [12]. In other words, compared with PS control, PSP control can expand the ZVS range to maximum in entire battery voltage range. (7) IV. CONTROL STRATEGY The control strategy of PSP is realized with two individual controllers, as shown in Fig. 5. The BDC is difficult to control stably because of the different small signal characteristics in different operation mode. In this paper, a novel control strategy with one port voltage regulation and another port current regulation is proposed. By sampling one port voltage (the port, ) and another port current (the battery port, ), the controller can realize the voltage regulation and current regulation in different energy transmission direction, respectively. The control strategy unifies the control system, simplifies the control circuit, and makes energy bidirectional conversion free [5]. The block diagram of phase-shift angle controller is shown in Fig. 5, which is used to control the magnitude and direction of the transmitted power. When the voltage value on side is higher than the reference, the converter operates in buck mode and is controlled by single current closed-loop. The constant-current setting for the low voltage side is decided by the current limiter, which can be regulated according to the charge condition of the battery. When the voltage value on side is lower than the reference, the converter will operate in boost mode, and be controlled by current and voltage dual closedloops. The maximum discharge current of the battery is limited by the current limiter. By selecting appropriate control parameters, this variable structure controller can improve the steady and dynamic performance of the system. The further studies about theoretical model analysis and choice criteria of the controller parameters will be presented in a coming paper. The duty cycle controller realizes the amplitude matching of and when varies. By sampling the voltage of clamping capacitor and the voltage /2 of the port, the controller can yield a signal which is transferred to the PWM chip (SG3525).Here the voltage equals the amplitude of, and the voltage is equal to the amplitude of the secondary voltage reflected to the primary. When the voltage value is higher than the clamping capacitor voltage, the duty cycle controller makes the signal rising. Sequentially, the duty cycle of and is increased to raise the voltage in accordance with the signal. Finally, the amplitude matching of and can be achieved. Contrarily, the duty cycle of and is reduced to match the amplitude of and. As can be seen in Fig. 5, the duty cycle can be independently modulated when varies. V. EXPERIMENTAL RESULTS AND DISCUSSIONS In order to verify the operation principle of the proposed converter, a 1.5 kw prototype was built in laboratory. The specifications of the converter are given as follows: 1) The battery voltage of side: VDC. 2) The rated voltage of side: VDC. 3) Rated power: kw. 4) The turns ratio of the transformer: :. 5) The leakage inductance of the transformer: H. 6) The inductance: H. 7) The clamping capacitor: F. 8) The capacitors: F. 9) Switches and : APT20M11JFLL. 10) Switches and : APT77N60JC3. 11) Switches and : APT20M16LFLL. 12) Switching frequency: khz. Fig. 6(a) and (b) show the experimental waveforms of the leakage inductor current, the primary voltage, and the secondary voltage at V in Boost mode under PSP and PS control, respectively. Since the amplitudes of and are matched in this case, the maximum current of under PSP control and PS control is the same. Fig. 6(c) and (d) show the experimental waveforms of the leakage inductor current, the primary voltage, and the secondary voltage at V in Boost mode under PSP control and PS control. In this case, the amplitudes of and are not matched under PS control. Therefore, the current stress of with PS control rises rapidly. As can be seen from Fig. 6(a) and (c), the amplitude matching of and is guaranteed in different battery voltage. Therefore, the duty cycle controller is valid. Fig. 7(a), (b) and (c) show the gate drive signal, voltage across the drain and source, and the drain current of and respectively, at V in Boost mode with 1.5 kw output power under PSP control. Fig. 8(a), (b) and (c) show the gate drive signal, voltage across the drain and source, and the drain current of and, respectively, at V and

818 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 23, NO. 2, MARCH 2008 Fig. 5. Control scheme (UC3875 is a phase shift resonant controller, CD4098 is a CMOS dual monostable multivibrator, SG3525 is a regulating pulse width modulator, and IR2110 is a high and low side driver). Fig. 6. Experimental waveforms at V =32V and V =22V. (a) PSP control at V =32V, V =270V, f =0:35;d=0:5; and P = 1500 W. (b) PS control at V =32V, V = 270 V, =0:35;d =0:5, and P = 1490 W. (c) PSP control at V =22V, V = 270 V, =0:88;d=0:65, and P = 1410 W. (d) PS control at V =22V, V = 270 V, =0:08;d =0:5, and P = 250 W. A in Buck mode with 1.5 kw output power under PSP control. Fig. 7 and Fig. 8 illustrate that all the switches realize ZVS. The experimental results are in agreement with the theoretical analysis well.

XIAO AND XIE: ZVS BDC WITH PHASE-SHIFT PLUS PWM CONTROL SCHEME 819 Fig. 7. Gate drive signal, the voltage across the drain and source, and the drain current of the switches at full load and V = 30V in Boost mode. (a) S. (b) S. (c) S. Fig. 9 shows the dynamic experimental waveforms of energy bidirectional conversion process, from up to bottom are voltage and current. When the voltage on port is higher than the reference value, the bidirectional dc dc converter charges the battery with constant current. When the voltage on port drops, the battery turns to discharge and maintains the voltage at 270 VDC. The experimental results convinced that the novel control strategy with one port voltage regulation and another port current regulation can control energy bidirectional conversion freely. The response time of voltage rebuilding is 10ms. Therefore, this converter has high steady and dynamic performance. Fig. 10(a) shows the overall efficiency curves at different load, different transmission direction, and different voltage with the PSP control. In this figure, the power transmitted from to is defined as positive, and the power transmitted from to is negative. We can see that the efficiency is higher in high battery voltage (such as V, the highest in Boost mode). Unfortunately, the efficiency is lower in low battery voltage (such as V, the highest in Boost mode). This degradation is due to the increase of conduction loss with the battery voltage decreases. Fig. 10(b) shows the efficiency curves of the converter under PSP control and PS control in Boost mode. From Fig. 10(b), it can be easily found that PSP control can achieve higher efficiency than PS control, especially in low battery voltage. The experimental results are in agreement with Fig. 4. VI. CONCLUSION A novel ZVS bidirectional dc dc converter with PS plus PWM control is proposed in this paper, which has the following advantages. 1) All switches realize ZVS in a wide range of load variation while input or output voltage varies. 2) The PS plus PWM control reduces the circulating current. 3) The converter avoids the voltage spike of and with the use of an active clamping branch and. 4) The control strategy realizes energy conversion freely, which has high steady and dynamic performance. These merits are verified by a 22 32 V/270 V 1.5 kw prototype. It can be concluded, this kind of converter is extremely suitable for aircraft HVDC power supply system and UPS system.

820 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 23, NO. 2, MARCH 2008 Fig. 8. Gate drive signal, the voltage across the drain and source, and the drain current of the switches at full load and V = 300 V in Buck mode. (a) S. (b) S. (c) S. The following equation is satisfied: (A1) The average voltage of in one switching period is zero (A2) Further (A3) Substituting (A2) and (A3) into (A1), the following is found: Fig. 9. Waveform of energy bidirectional Transmitted. (A4) APPENDIX A From Fig. 1(b), we can see that the controlling of and is to match the amplitude of and in the stage. APPENDIX B This Appendix is provided to derive the relation of output power versus phase-shift angle and duty cycle, the process can be divided into four intervals.

XIAO AND XIE: ZVS BDC WITH PHASE-SHIFT PLUS PWM CONTROL SCHEME 821 b) (B4) (B5) Substituting (B4) into (B5), the following is found: (B6) c), referring to Fig. 1(c) (B7) (B8) Fig. 10. Conversion efficiency (The power transferred from V to V is defined as positive, and the power transferred from V to V is defined as negative.) (a) Efficiency with PSP control under different output power, V, and V voltage. (b) Efficiency comparison in Boost mode under the PSP control and PS control. Substituting (B7) into (B8), the following is found: a), referring to Fig. 1(b) (B9) d) (B1) (B10) (B11) (B2) Substituting (B10) into (B11), the following is found: Substituting (B1) into (B2), the following is found: (B12) (B3) Combining (B3), (B6), (B9), and (B12), the expression (3) can be obtained.

822 IEEE TRANSACTIONS ON POWER ELECTRONICS, VOL. 23, NO. 2, MARCH 2008 APPENDIX C A Bidirectional dc dc converter with Phase-shift control strategy was introduced in [10], the following is listed: ( V, ) can be yielded from (D1), and (D2). a) When, referring to Fig. 1(b)) a) b) (C1) (C2) (D3) (D4) The RMS value of can be expressed as follows: (C3) (C4) Substituting (C1), (C2), and (C3) into (C4), you can find (C5), shown at the bottom of the page. (D5) Substituting (B1), (D3), (D4), and (D5) into (C4), can be expressed as (D6), shown at the bottom of the page. b) When (D7) APPENDIX D In order to derive the RMS value of in full load in Boost mode under PSP control, first, we should decide which interval the phase-shift is in under different. Referring to Appendix B, the following equations can be listed: (D1) Substituting (B4), (D7), (D8), and (D9) into (C4), can be expressed as (D8) (D9) (D2) (D10) In the proposed conditions ( V, : kw, khz, H), the results Combining (D6), and (D10), the expression (5) can be obtained. (C5) (D6)

XIAO AND XIE: ZVS BDC WITH PHASE-SHIFT PLUS PWM CONTROL SCHEME 823 ACKNOWLEDGMENT The authors would like to thank M. Shi, NUAA, Y. Tang, NUAA, L. Guo, NUAA, and F. Lin, MF, Inc., for their help during the experiments and revisions. REFERENCES [1] S. Inoue and H. Akagi, A bidirectioanl isolated dc dc converter as a core circuit of the next-generation medium-voltage power conversion system, IEEE Trans. Power Electroni., vol. 22, no. 2, pp. 535 542, Mar. 2007. [2] F. Zhang, L. Xiao, and Y. Yan, Bi-directional forward-flyback dc dc converters, in Proc. IEEE PESC, 2004, pp. 4058 4061. [3] L. Zhu, A novel soft-commutating isolated boost full-bridge ZVS-PWM dc dc converter for bidirectional high power applications, IEEE Trans. Power Electron., vol. 21, no. 2, pp. 422 429, Mar. 2006. [4] H.-J. Chiu and L.-W. Lin, A bidirectional dc dc converter for fuel cell electric vehicle driving system, IEEE Trans. Power Electron., vol. 21, no. 4, pp. 950 958, Jul. 2006. [5] H. Xiao, D. Chen, and S. Xie, A ZVS Bi-directional dc dc converter for high-low voltage conversion, in Proc. IEEE IECON, 2005, pp. 1154 1158. [6] A. Emadi and M. Ehsani, Aircraft power systems: Technology, state of the art, and future trends, IEEE AES Syst. Mag., vol. 15, pp. 28 32, Jan. 2000. [7] J. L. Duarte, M. Hendrix, and M. G. Simoes, Three-Port bidirectional converter for hybrid fuel cell systems, IEEE Trans. Power Electron., vol. 22, no. 2, pp. 480 487, Mar. 2007. [8] M. Marchesoni and C. Vacca, New dc dc converter for energy storage system interfacing in fuel cell hybrid electric vehicles, IEEE Trans. Power Electron., vol. 22, no. 1, pp. 301 308, Jan. 2007. [9] R. W. De Doncker, D. M. Divan, and M. H. Kheraluwala, Power Conversion Apparatus for dc/dc Conversion Using Dual Active Bridge, U.S. Patent 5 027 264, 2005. [10] M. H. Kheraluwala, R. W. Gascoigne, and D. M. Divan, Performance characterization of a high-power dual active bridge dc-to-dc converter, IEEE Trans. Ind. Appl., vol. 28, no. 6, pp. 1294 1031, Nov. 1992. [11] F. Z. Peng, H. Li, and G.-J. Su et al., A new ZVS bidirectional dc dc converter for fuel cell and battery application, IEEE Trans. Power Electron., vol. 19, no. 1, pp. 54 65, Jan. 2004. [12] D. Xu, C. Zhao, and H. Fan, A PWM plus phase-shift control bidirectional dc dc converter, IEEE Trans. Power Electron., vol. 19, no. 3, pp. 666 675, May 2004. [13] S.-K. Han, H.-K. Yoon, and G.-W. Moon et al., A new active clamping zero-voltage switching PWM current-fed half-bridge converter, IEEE Trans. Power Electron., vol. 20, no. 6, pp. 1271 1279, Nov. 2005. Huafeng Xiao was born in Hubei, China, in 1982. He received the B.S. and M.S. degree in electrical engineering from Nanjing University of Aeronautics and Astronautics (NUAA), Nanjing, China, in 2004 and 2007, respectively, where he is currently pursuing the Ph.D. degree in electrical engineering. His main research interests include high frequency soft-switching conversion, and photovoltaic applications. Shaojun Xie (M 05) was born in Hubei, China, in 1968. He received the B.S., M.S., and Ph.D. degrees in electrical engineering from Nanjing University of Aeronautics and Astronautics (NUAA), Nanjing, China, in 1989, 1992, and 1995, respectively. In 1992, he joined the Faculty of Electrical Engineering, Teaching and Research Division, and is currently a Professor at the College of Automation Engineering, NUAA. He has authored more over 50 technical papers in Journals and Conference proceedings. His main research interests include aviation electrical power supply systems and power electronics conversion.