A K-Band Aperture-Coupled Microstrip Leaky-Wave Antenna

1236 IEICE TRANS. ELECTRON., VOL.E82 C, NO.7 JULY 1999 PAPER Special Issue on Microwave and Millimeter-Wave Technology A K-Band Aperture-Coupled Microstrip Leaky-Wave Antenna Tai-Lee CHEN and Yu-De LIN a), Nonmembers SUMMARY Microstrip leaky-wave antenna fed by an aperture-coupled microstrip operating at K-band is presented. Using the aperture-coupled microstrip as a feeding structure can fully exploit the wideband characteristic of the microstrip leakywave antenna. The dimensions of the antenna are obtained from the calculation of the propagation characteristics. Measurement shows a bandwidth of 22% for V SWR < 2 : 1 and a peak power gain of 12 dbi at 22 GHz for one element. Four-element array is developed with a gain of 18.7 dbi and the frequency-scanning feature is exhibited. The waveguide model is verified by measuring the 3-D radiation pattern of the microstrip leaky-wave array. key words: K-band antenna, aperture-coupling, microstrip, leaky-wave 1. Introduction More and more applications have been built on the development of integrated circuit technique at the millimeter-wave frequency band. Antennas designed at K-band have been utilized for direct data distribution system in space-to-ground communication links, satellite-tracking in aeronautical mobile application, commercial mobile satellite communication system, receiver for remote sensing, radar sensor for short range radar applications, and quasi-optical image system, etc. [1]. Leaky-wave theory and applications based on the higher order modes of planar transmission lines have brought about broad interests and discussions recently. Leaky-wave antennas based on the higher order modes of planar transmission lines possess the advantages of higher antenna gain, wider bandwidth, frequency scanning, relaxed requirement of tolerance, etc. [2] [4]. In [2], the microstrip line first higher order leaky mode was excited by an unsymmetrical quasi-tem microstrip line. Matching networks, such as quarter-wave transformers, were required to match the antenna s input impedance. Three arrangements of the feeding structures were proposed and investigated by Lin et al. [3], in one of which the microstrip-to-slotline transition was used. Yet, the radiation pattern would be affected by the feeding circuits on the same side with the antenna. In [4], the slotline feeding method with a CPW-to- Manuscript received December 28, 1998. Manuscript revised March 17, 1999. The authors are with the Institute of Communication Engineering, National Chiao Tung University, Hsinchu, Taiwan, R.O.C. a) E-mail: ydlin@cc.nctu.edu.tw slotline transition applied to the active leaky-wave antenna was developed by Tzuang et al. These transitions and matching networks are narrow band in nature and limit the intrinsic bandwidth of leaky-wave antennas. Patch antennas fed by the microstrip via an aperture have been discussed in [5] and used in many applications. By separating the feeding networks from the antenna with the ground plane, this type of antennas offers two advantages: first, the feeding circuits and the antenna are based on different substrates, and can be optimized respectively; secondly, the ground plane serves to eliminate the spurious radiation from the feeding and the matching network and therefore it would not contaminate the radiation pattern. However, array or stacked design is usually required for high-gain or wideband applications due to its resonant characteristic. In this paper we use the aperture-coupled microstrip to feed the leaky-wave antenna to eliminate the spurious radiation and bandwidth limitation from the feeding circuits. 2. Microstrip First Higher Order Leaky Mode Apart from the antenna applications of the leaky wave resulting from slit cut in waveguide, the NRD guide, the image guide, and dielectric periodic structure, space wave leaks from microstrip have been discussed theoretically [6]. From the viewpoint of traveling-wave antenna, the complex propagation constant should be known in order to determinate the specifications of the antenna. The normalized phase and attenuation constants of the microstrip line first higher order mode shown in Fig. 1are obtained by the spectral domain analysis with appropriate choice of branch cuts and integration contours as in [7]. The space-wave leaky region spans from about 20 to 27 GHz for a substrate of ε r1 =2.2, h 1 =0.508 mm and w=4.3 mm of the microstrip. The width of the microstrip is determined by the desired frequency band in which the space-leakywave occurs; while the choice of the length depends on the effect of the reflection from the end of the microstrip. The current distributions at 22 GHz of the conventional dominant mode and the first higher order leaky mode of the infinite microstrip are shown in Fig. 2. It can be seen that besides the different guided wavelengths and transverse resonance phenomenon in the

CHEN and LIN: A K-BANDAPERTURE-COUPLEDMICROSTRIP LEAKY-WAVE ANTENNA 1237 Fig. 1 Normalized phase constant β/k 0 and attenuation constant α/k 0 of the first higher order mode of the microstrip. w =4.3mm, ε r =2.2, h =0.508 mm. Fig. 2 Comparison of the current distributions of the conventional dominant mode (upper) with the first higher order leaky mode (lower) of the microstrip. w =4.3mm, ε r =2.2, h =0.508 mm, f req = 22 GHz. leaky-mode code, the amplitude of the leaky mode decays exponentially along the microstrip since the power is radiated as the mode is guided down. 3. Design of the K-Band Aperture-Coupled Leaky-Wave Array 3.1Aperture-Coupled Design The microstrip leaky-wave antenna with the feeding structure is depicted in Fig. 3. The top layer contains the leaky-wave microstrip antenna, and the feeding microstrip is placed on the other side of the ground plane, with a coupling aperture between them. Here we use e 2αL to estimate roughly how much power has been radiated from the leaky-wave antenna with length L, where α is the attenuation constant of the microstrip first higher order leaky mode. Less than 5% of the power survives for the frequency below 25 GHz in this Fig. 3 K-band aperture-coupled microstrip leaky-wave antenna. L = 50 mm, w =4.3mm, l sc =1.7mm, l so =1.3 mm, w s = 0.2 mm, ε r1 = 2.2, ε r2 = 10.2, h 1 = 0.508 mm, h 2 = 0.635 mm, w m =0.68 mm, l m = 1 mm. experiment, thus the effect of the reflection caused by the end of the microstrip would be small. It can be observed from the leaky-mode current distribution shown in Fig. 2 that the direction of the current on the centerline of the microstrip is y-directed only, therefore, the feeding structure with a narrow slot arranged along the centerline on the ground plane will not disturb the current distribution so as to excite the leaky mode. The coupling length (l sc in Fig. 3.) of the slot to the top leaky-wave microstrip is chosen to be shorter than λ g /4 for the desired frequency band to prevent the conflict of the fields caused by the slot and the leaky mode of the microstrip. Here, λ g is the guided wavelength of the microstrip first higher order mode. Since the coupling length decides the coupling energy to the leaky mode, the overlapped length can be tuned to match the real part of the input impedance looking into the aperture from the point of view of the circuit. The part of the slot outside the leaky microstrip (l so ), however, can be adjusted to match the imaginary part of the input impedance because of the resonant characteristic of the slot. A 50 Ω open microstrip is used to excite the slot on the other side of the ground plane. The length of the open stub l m (from the center of the aperture to the open end), which serves as the other matching circuit for the imaginary part of input impedance, is selected to be about λ m /4 to obtain maximum coupling to the slot; where λ m is the guided wavelength of the feeding microstrip on the backside of the radiator. Figure 4 shows the moment method simulation of the current distribution from the commercial available IE3D EM package [8]. The guided wave propagates along the feeding line without attenuation and through the coupling aperture; yet the power gradually leaks out along the top layer microstrip leaky-wave antenna. The current distribution of the top layer microstrip is the same as the leaky mode as shown in Fig. 2, so we can see what we excited is the leaky mode instead of the bound mode.

1238 IEICE TRANS. ELECTRON., VOL.E82 C, NO.7 JULY 1999 Fig. 4 The current distribution of the simulating result by the IE3D EM package. Fig. 5 The four-element aperture-coupled microstrip leakywave antenna array using the microstrip T-junction power dividers with modified quarter-wave transformers. The specification is the same as Fig. 3, spacing in elements d = 7.8 mm. 3.2 Array Design The structure of an equally-spaced four-element linear array is shown in Fig. 5. Microstrip T-junction power dividers are adopted to implement the equal-amplitude and equal-phase array. Modified quarter-wavelength transformers were used to match the power dividers to the 50 Ω input port. In general, to arrange the feeding network for array design, some problems should be treated carefully. First, the feeding network had better been separated from the radiator to avoid the interaction of the near fields from both parts. Secondly, the far field from the feeding network might contaminate the desire radiation pattern. Other considerations are the limitation from the properties of the substrate and the area taken by the network layout. Mismatching of the power dividers will lead to resonance and thus the feeding network might act as a radiator. The aperturecoupled feeding method can solve all of the problems and therefore it can exhibit the inherent characteristic of the leaky-wave microstrip array. 3.3 Waveguide Model To estimate the radiation direction and the antenna gain from the microstrip leaky-wave array, the radiation pattern is calculated by the waveguide (cavity) model using the equivalence principle [4]. The aperture electric field under edge of the leaky microstrip for the equivalent magnetic surface current n E aperture = M is E aperture = E 0 e jβx e αx ẑ (1) The far field derived from the Fourier transform of the apertures of the microstrip array is e jk 0r E φ = E 0 cos((k 0 w/2) sin θ cos φ) πr sin(k 0h sin θ sin φ) sin φ ejl(k 0cos θ k x ) 1 AF k 0 cos θ k x (2) where k x = β jα is the guided wavenumber, and k 0 is Fig. 6 The measured return loss of the antenna in Fig. 3. the freespace wavenumber. The array factor AF of the n-element equal-spaced linear array can be expressed as AF = n i=1 exp( jk 0 (n 2i+1) w+d 2 4. Experimental Results 4.1Measured Bandwidth sin θ cos φ) (3) Figure 6 shows the measured return loss of the single element antenna. The bandwidth for V SWR < 2: 1 is 22% (20.6 25.6 GHz), which is in the predicted leaky radiation band as shown in Fig. 1. 4.2 Validity of the Waveguide Model The 3-D patterns derived from the waveguide model compared with the measured results are shown in Figs. 7 and 8, where the optimization of the spacing d for maximum directivity is adopted. Figure 7(a) shows the E φ power gain pattern of the single element calculated by (2) with AF = 1at 22 GHz. The measured pattern with peak power gain of 12 dbi is shown in Fig. 7(b). The 3-dB beamwidth of the elevation angle (θ) and the azimuth angle (ψ) are 22 and 122 respectively, which reveals the cone-like fan-beam property

CHEN and LIN: A K-BAND APERTURE-COUPLED MICROSTRIP LEAKY-WAVE ANTENNA 1239 (a) (a) (b) (b) Fig. 7 (a) The 3-D Eφ power gain pattern of the single element antenna in Fig. 3 calculated by the waveguide model. The contour level is 3-dB. (b) Measured result. Fig. 8 (a) The 3-D Eφ power gain pattern of the four-element array in Fig. 4 calculated by the waveguide model. The contour level is 3-dB. (b) Measured result. of the microstrip leaky-wave antenna. The pattern of the four-element array using the specification in Fig. 5 is shown in Fig. 8(b). The 3-dB beamwidth of the azimuth angle shrinks to 15 as a pencil beam and the peak power gain rises to 18.7 dbi. By comparing with the pattern derived by (2), which is shown in Fig. 8(a), it is seen that the waveguide model can estimate not only the pattern but also the antenna gain approximately. The cross-polarization Eθ is 18 db down less than the peak power of the co-polarization for all measurement throughout this study. smaller dimension provides more freedom for circuit arrangement. The measured backside cross-polarization is over 18 db down from the main peak. Selecting suitable feeding substrate, as in comparison with the result in [9], can eliminate the unwanted fields that are unavoidable for single layer antenna design. 4.3 Selection of the Feeding Substrate In [9], the substrate of the feeding layer is the same as the layer on which the radiators distribute. The wavelength (9.8 mm) and the width (1.6 mm) of the 50 Ω feeding microstrip might be too large to arrange the feeding network under the limitation of the layout area. Besides, the impedance mismatch and the parasitic effects in the feeding network will lead to serious crosspolarization on the backside and a complex design as in [9]. In this paper, a higher dielectric constant substrate of εr2 = 10.2, h2 = 0.635 mm is used as the feeding layer. The width of the 50 Ω microstrip is only 0.68 mm and the guided wavelength is a half of that of the lower dielectric substrate mentioned above. The 4.4 Scanning Characteristic The measured H-plane (x-z plane) Eψ radiation power gain patterns are shown in Fig. 9. To exhibit the frequency-scanning property of the leaky-wave antenna, the operating frequency is varied in the leaky-wave region. When the operating frequency is increased from 22 GHz to 25 GHz, the mainbeam direction scans from an elevation angle of 58 to one of 40, as shown in Fig. 9(a). Figure 9(b) shows the measured H-plane radiation power gain patterns of the four-element array with the space between the elements being 7.8 mm. The Eψ peak power gain is 18.7 dbi at 22 GHz with elevation angle 57. When the frequency is increased to 25 GHz, the peak scans down to an elevation angle of 38. The reflected lobe (θ = 140 ) at 25 GHz (15 db lower than the peak power) is more evident than that of 22 GHz. This is because the attenuation constant at 25 GHz is smaller, which causes larger reflection from the open

1240 IEICE TRANS. ELECTRON., VOL.E82 C, NO.7 JULY 1999 Acknowledgment This work was supported in part by National Science Council under the grants: NSC 87-2213-E-009-108. References (a) (b) Fig. 9 (a) Measured H-plane (φ =90 ) E ψ radiation power gain patterns of the single element shown in Fig. 3 for f =22 (solid line) and 25 GHz (dashed line). (b) of the four-element array shown in Fig. 4. [1] B. Zimmermann and W. Wiesbeck, 24 GHz microwave close-range sensors for industrial measurement applications, Microwave Journal, pp.228 238, May 1996. [2] W. Menzel, A new traveling-wave antenna in microstrip, Archiv fur Electronik und Ubertranungstechnik, vol.33, no.4, pp.137 140, April 1979. [3] Y.-D. Lin, J.-W. Sheen, and C.-K.C. Tzuang, Analysis and design of feeding structures for microstrip leaky wave antenna, IEEE Trans. Microwave Theory & Tech., vol.44, no.9, pp.1540 1547, Sept. 1996. [4] G.-J. Chou and C.-K.C. Tzuang, Oscillator-type activeintegrated antenna: The leaky-mode approach, IEEE Trans. Microwave Theory & Tech., vol.44, no.12, pp.2165 2272, Dec. 1996. [5] D.M. Pozar, Microstrip antenna aperture-coupled to a microstripline, Electronics Letters, vol.21, no.2, pp.49 50, Jan. 1985. [6] A.A. Oliner and K.S. Lee, The nature of the leakage from higher-order modes on microstrip line, 1986 IEEE MTT-S Int. Microwave Symp. Dig., Baltimore, pp.57 60, 1986. [7] Y.-D. Lin and J.-W. Sheen, Mode distinction and radiationefficiency analysis of planar leaky-wave line source, IEEE Trans. Microwave Theory & Tech., vol.45, no.10, pp.1672 1680, Oct. 1997. [8] IE3D electromagnetic simulation and optimization package, ZELAND SOFTWARE, INC. [9] T.-L. Chen and Y.-D. Lin, A K-band aperture-coupled microstrip leaky-wave antenna, 1998 Asia-Pacific Microwave Conference Proceedings, Yokohama, Japan, pp.1491 1494, 1998. end. The larger backside radiation might be due to the resonance of slots. 5. Conclusion Microstrip leaky-wave array fed by aperture-coupled microstrip operating at K-band is presented. The dimensions of the antenna are obtained by the spectral domain analysis. Measurement results are in agreement with the leaky band prediction with a bandwidth of 22% for V SWR < 2 : 1. Tilted beam with high peak power gain (12 dbi) for single element and beamscanning property were observed. Four-element arrays were developed with peak power gain 18.7 dbi at 22 GHz. The validity of using the waveguide model to estimate the 3-D pattern of the microstrip leaky-wave array is demonstrated. Connection with the circuits on the other side of the ground plane reduces the interference caused by the feeding networks, therefore the wideband and frequency-scanning capabilities of the leaky-wave antenna can be effectively exhibited. Tai-Lee Chen received the B.S. degree in mathematics form the National Taiwan University, Taipei, Taiwan, R.O.C., in 1989 and the M.S. degree in communication engineering from the National Chiao Tung University (NCTU), Hsinchu, Taiwan, R.O.C., in 1991, where he is currently working toward the Ph.D. degree. His research interests include analytical and computational electromagnetics, microwave circuits and antenna design.

CHEN and LIN: A K-BANDAPERTURE-COUPLEDMICROSTRIP LEAKY-WAVE ANTENNA 1241 Yu-De Lin received the B.S. degree in electrical engineering from National Taiwan University, Taipei, Taiwan, R.O.C., in 1985, and the M.S. and Ph.D. degrees in electrical engineering from the University of Texas at Austin, in 1987 and in 1990, respectively. In 1990, he joined the faculty of the Department of Communication Engineering, National Chiao Tung University, Hsinchu, Taiwan, R.O.C., where he is currently an Associate Professor. His current research interests include numerical methods in electromagnetics, characterization and design of microwave and millimeter-wave planar circuits, and analysis and design of microwave and millimeter-wave antennas.