REFERENCES 1. G.V. Eleftheriades, A.K. Iyer, and P.C. Kremer, Planar negative refractive index media using periodically L-C loaded transmission lines, IEEE Trans Microw Theory Tech 50 (2002), 2702 2712. 2. C. Caloz and T. Itoh, Application of the transmission line theory of left-handed materials to the realization of a microstrip left-handed transmission line, In the IEEE-APS international symposium, San Antonio, TX, Vol. 2, 2002, pp. 412 415. 3. R.A. Shelby, D.R. Smith, and S. Schultz, Experimental verification of a negative index of refraction, Science 292 (2001), 77 79. 4. A. Sanada, C. Caloz, and T. Itoh, Characteristics of the composite right/left-handed transmission lines, IEEE Microw Wireless Compon Lett 14 (2004), 68 70. 5. C. Caloz and T. Itoh, Novel microwave devices and structures based on the transmission line approach of meta-materials, In the IEEE MTT-S international symposium digest, Philadelphia, PA, Vol. 1, 2003, pp. 195 198. 6. M.A. Antoniades and G.V. Eleftheriades, Compact linear lead/lag metamaterial phase shifters for broadband applications, IEEE Antennas Wireless Propagat Lett 2 (2003), 103 106. 7. Youzhen Wang, Yewen Zhang, Li He, Fuqiang Liu, Hongqiang Li and Hong Chen, Tunable asymmetric composite right-/left-handed transmission line directional coupler controlled by applied voltage, In the Asia Pacific Microwave Conference Proceedings, Suzhou, China December 2005, pp. 170 173. 8. A. Sanada, C. Caloz, and T. Itoh, Novel zeroth-order resonance in composite right/left-handed transmission line resonators, In the Asia Pacific Microwave Conference Proceedings, Seoul, Korea 2003, pp. 1588 1591. 9. A. Sanada, M. Kimura, I. Awai, C. Caloz, and T. Itoh, A planar zeroth-order resonator antenna using a left-handed transmission line, In the 34th EMC2004, pp. 1341 1344. 2007 Wiley Periodicals, Inc. BROADBAND INTEGRATED DTV ANTENNA FOR USB DONGLE APPLICATION Wei-Yu Li, 1 Kin-Lu Wong, 1 and Saou-Wen Su 2 1 Department of Electrical Engineering, National Sun Yat-Sen University, Kaohsiung 804, Taiwan 2 Technology Research and Development Center, Lite-on Technology Corp., Taipei 114, Taiwan Received 29 September 2006 ABSTRACT: A novel broadband planar antenna integrated in the USB dongle for digital television (DTV) signal reception is presented. The planar antenna is printed on a dielectric substrate and electrically connected to the system ground plane of the USB dongle. When in operation, the planar antenna is swung upward to be perpendicular to the system ground plane and can provide a wide operating bandwidth (2.5:1 VSWR) of larger than 50% centered at about 630 MHz, allowing it to cover the DTV signal reception in the 470 806-MHz band. While not in operation, the planar antenna can be swung downward and attached onto the housing of the USB dongle, keeping the aesthetic appearance of the device. Detailed design considerations of the proposed antenna are described, and obtained experimental and simulation results are presented and discussed. 2007 Wiley Periodicals, Inc. Microwave Opt Technol Lett 49: 1018 1021, 2007; Published online in Wiley InterScience (www.interscience.wiley.com). DOI 10.1002/mop.22329 Key words: mobile antennas; DTV antennas, broadband antennas, integrated antennas, USB dongle 1. INTRODUCTION The universal series bus (USB) dongles [1] embedded with antennas for achieving wireless communications are becoming very attractive for application in the laptop computers. For this kind of perspective application, promising internal multiband antenna for mobile communications for the USB dongle has recently been reported [2, 3]. Since other possible functions for the USB dongle, such as the digital television (DTV) [4 6] signal reception, are expected to be very attractive for wireless users, we demonstrate in this paper a novel broadband planar antenna integrated in the USB dongle for DTV signal reception in the 470 806 MHz band [7]. The proposed planar antenna is of a simple, long inverted-u shape with an open gap, and is easy to fabricate by printing on a dielectric substrate. By properly selecting the position of the open gap, the proposed planar antenna incorporating the system ground plane of the USB dongle can generate two adjacent resonant modes to form into a wide operating band of larger than 50% to cover the 470 806 MHz band for DTV signal reception. Moreover, owing to its planar structure [8], the proposed antenna can be attached onto the housing of the USB dongle when not in operation, keeping the aesthetic appearance of the device. Details of the design considerations of the proposed integrated planar DTV antenna are described. A parametric study on analyzing the effects of various antenna parameters on the performances of the antenna is also conducted. The proposed antenna is also constructed and tested, and obtained experimental and simulated results are presented and discussed. 2. PROPOSED INTEGRATED DTV ANTENNA Figure 1(a) shows the geometry of the proposed broadband integrated planar DTV antenna for application in the USB dongle. Note that the geometry shown in Figure 1(a) is for the operation condition in which the proposed antenna is perpendicular to the system ground plane of the USB dongle. When not in operation, the proposed antenna can be bent to be parallel to the system ground plane of the USB dongle [Fig. 1(b)] and attached onto the housing of the USB dongle (see the example of a promising USB dongle embedded with the DTV tuner module and proposed antenna shown in Fig. 2). The proposed antenna has a long inverted-u strip with an open gap of length g printed on a 0.8-mm thick FR4 substrate of size 120 30 mm 2. Owing to the presence of the open gap, the inverted-u strip is separated into two portions. One is a long straight strip of length h and uniform width w and is denoted as strip 1 in this study. The other is an inverted-l strip of uniform width 10 mm and is denoted as strip 2 here. One end of strip 2 is electrically connected to the system ground plane of size 110 30 mm 2 at point C shown in Figure 1(a). The system ground plane in the study is printed on a 0.8-mm thick FR4 substrate, which can be considered as the system circuit board of the USB dongle. Also, the chosen dimensions of the system ground plane here are reasonable ones for general USB dongles. Across one end of strip 1 and the system ground plane, there is a 1-mm feed gap. For testing the antenna, a 50- mini coaxial line is used, with its central conductor connected to strip 1 at point A and its outer grounding sheath connected to the system ground plane at point B shown in Figure 1. By incorporating the system ground plane as part of the resonant path, the proposed antenna can first be operated as a halfwavelength resonant loop structure, which begins from point A, then through the inverted-u strip formed by strips 1 and 2, to point C and further to point B through the short side edge of the system ground plane. This resonant loop structure has a mean length of 1018 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 49, No. 5, May 2007 DOI 10.1002/mop
Figure 2 Example of a promising USB dongle integrated with the proposed antenna. [Color figure can be viewed in the online issue, which is available at www.interscience.wiley.com] modes, and the detailed effects will be discussed with the aid of Figure 6. Figure 1 (a) Geometry of the proposed broadband integrated planar DTV antenna for USB dongle application. (b) Geometry of the proposed DTV antenna when not in operation. [Color figure can be viewed in the online issue, which is available at www.interscience.wiley.com] about 270 mm in this design, which leads to the excitation of a resonant mode at about 500 MHz for the proposed antenna. Then, by properly selecting the position of the open gap, a second resonant mode controlled by strip 1 and the system ground plane can be excited. This second resonant mode is related to a halfwavelength dipole structure formed by two asymmetric dipole arms of strip 1 and the system ground plane. Thus, by adjusting the length h of strip 1, this second resonant mode can be controlled. In this study, this second resonant mode is designed to occur at about 700 MHz. In addition, the length g of the open gap is found to greatly affect the impedance matching of the antenna s two resonant modes. By choosing a proper length of the open gap (1 mm in this study), good impedance matching of the antenna s two resonant modes can be obtained, and a wide operating band formed by the two resonant modes can be achieved for the proposed antenna to cover the DTV signal reception in the 470 806 MHz band. More detailed effects of the length g of the open gap and the length h of strip 1 will be discussed with the aid of Figure 5 in the next section. The width w of strip 1 is also found to have some effects on the impedance matching of the antenna s two resonant 3. RESULTS AND DISCUSSION A preferred prototype of the proposed planar DTV antenna shown in Figure 1 with g 1 mm, h 80 mm, and w 10 mm was first constructed and studied. Figure 3 shows the measured and simulated return loss for the constructed prototype. The simulated results are obtained using Ansoft simulation software HFSS (High Frequency Structure Simulator) [9], and good agreement between the measurement and simulation is obtained. From the measured results, it is clearly seen that two adjacent resonant modes are generated, which form a wide operating band of about 52% (2.5:1 VSWR bandwidth) centered at about the desired center frequency of 638 MHz to cover the DTV band of 470 806 MHz. Note that the bandwidth definition of 2.5:1 VSWR (about 7.3-dB return loss) is used here, which is generally acceptable for DTV signal reception in practical applications. To analyze the two excited resonant modes, Figure 4 shows the measured return loss for the constructed prototype studied in Figure 3 Measured and simulated return loss for the proposed antenna with g 1 mm, h 80 mm, and w 10 mm. [Color figure can be viewed in the online issue, which is available at www.interscience.wiley.com] DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 49, No. 5, May 2007 1019
Figure 4 Measured return loss for the proposed antenna, the antenna without strip 2, and the antenna without the open gap (g 0). The antenna parameters are the same as studied in Figure 3. [Color figure can be viewed in the online issue, which is available at www.interscience.wiley.com] Figure 3, the antenna without strip 2, and the antenna without the open gap (g 0). It is clearly seen that, when strip 2 is not present, the antenna s first resonant mode disappears, with the second resonant successfully excited only. On the other hand, when the open gap is not present (g 0), the antenna s second resonant mode disappears and only the first resonant mode is excited. The obtained results are expected as described in Section 2. To analyze further, the antenna s radiation efficiency contributed from various portions of the proposed antenna including the system ground plane is studied by using Zeland simulation software IE3D [10]. The simulated results at 500 and 700 MHz are listed in Table 1. From the results at 500 MHz, the antenna s radiation efficiency are mainly contributed from strips 1 and 2, which indicates that the antenna s first resonant mode is mainly related to the half-wavelength loop structure as explained in Section 2. On the other hand, from the results at 700 MHz, the antenna s second resonant mode is mainly contributed from strip 1 and the system ground plane, which serve as the two asymmetric arms of the half-wavelength dipole structure. An experimental study for analyzing the effects of the parameters h and g on the impedance matching of the two excited resonant modes was also conducted. Figure 5 shows the measured return loss for the antenna with three different values of h and g. Note that the total length of h and g is fixed to be 81 mm in this study. From the results, it is seen that the resonant frequency of the antenna s first resonant mode is almost the same for the three different cases; however, the impedance matching is degraded with an increase in the length g of the open gap. This behavior is largely because the larger value of g leads to larger destruction in the antenna s resonant loop structure, hence resulting in the degradation in the impedance matching of the antenna s first resonant Figure 5 Measured return loss for the proposed antenna as a function of h (the length of strip 1) with the total length of h and g fixed to be 81 mm; other parameters are the same as studied in Figure 3. [Color figure can be viewed in the online issue, which is available at www.interscience.wiley. com] mode. On the other hand, it is seen that the resonant frequency of the antenna s second resonant mode is shifted to higher frequencies when the length h is smaller. This agrees with the expectation that the antenna s second resonant mode is related to the halfwavelength dipole structure with strip 1 and the system ground plane as the two asymmetric arms. In this case, the smaller length in h will lead to a smaller total length of the dipole structure, and thus the resonant frequency will be shifted to higher frequencies. Effects of the width w of strip 1 on the impedance matching of the antenna were also studied. Figure 6 shows the measured return loss for the width w varied from 5 to 15 mm, with other parameters the same as those in Figure 3. Generally, the resonant frequencies of the two excited resonant modes are about the same. However, by selecting a proper width (w 10 mm in this study), improved impedance matching over the desired DTV band can be obtained. The obtained results indicate that the width w is also an important factor for achieving improved impedance matching over the desired DTV band for the proposed antenna. Radiation characteristics of the proposed antenna were also studied. Since our anechoic chamber cannot operate at low fre- TABLE 1 Simulated Radiation Efficiencies (Obtained From Zeland IE3D) for Different Portions of the Proposed Antenna and the System Ground Plane at 500 and 700 MHz Efficiency at 500 MHz (%) Efficiency at 700 MHz (%) System ground plane 9 56 Strip 1 19 33 Strip 2 72 11 Total efficiency 100 100 Figure 6 Measured return loss for the proposed antenna as a function of w (the width of strip 1); other parameters are the same as studied in Figure 3. [Color figure can be viewed in the online issue, which is available at www.interscience.wiley.com] 1020 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 49, No. 5, May 2007 DOI 10.1002/mop
Figure 9 Simulated antenna gain and radiation efficiency for the proposed antenna studied in Figure 3. [Color figure can be viewed in the online issue, which is available at www.interscience.wiley.com] Figure 7 Simulated radiation patterns at 500 MHz for the proposed antenna studied in Figure 3. [Color figure can be viewed in the online issue, which is available at www.interscience.wiley.com] quencies such as those in the UHF band, the radiation characteristics of the proposed antenna were studied using Ansoft simulation software HFSS, which is expected to provide reliable results for the proposed antenna. Figure 7 shows the simulated radiation pattern at 500 MHz. In this case, monopole-like radiation pattern patterns with good omnidirectional radiation are observed. The corresponding results at 700 MHz are plotted in Figure 8. From the results, the obtained omnidirectional radiation characteristic is still good. Figure 9 shows the simulated antenna gain and radiation efficiency of the proposed antenna studied in Figure 3. The antenna gain is found to vary from about 0 to 1.9 dbi over the DTV band, while the radiation efficiency for frequencies over the band is all better than 60%. 4. CONCLUSION A novel broadband integrated planar DTV antenna for USB dongle application has been proposed. The antenna is with a simple structure and is easy to fabricate with a low cost. In addition, the antenna can generate two adjacent resonant modes to provide a wide operating bandwidth of larger than 50% to cover the DTV band in the 470 806 MHz. The proposed antenna is designed to be perpendicular to the system ground plane of the USB dongle in the operation condition. However, when not in the operation condition, the planar structure of the proposed antenna makes it very promising to be swung downward and attached onto the surface of the dongle housing to achieve an aesthetic appearance of the device. The proposed antenna has been successfully fabricated and tested. Detailed design considerations for the proposed antenna have been described. Over the 470 806 MHz band for DTV signal reception, good radiation characteristics have also been obtained for the proposed antenna. REFERENCES 1. Wikipedia, http://en.wikipedia.org/wiki/usb. 2. W.C. Su and K.L. Wong, Internal PIFAs for UMTS/WLAN/WiMAX multi-network operation for a USB dongle, Microwave Opt Technol Lett 48 (2006), 2249 2253. 3. K.L. Wong and C.H. Kao, Internal GSM/DCS/PCS antenna for USB dongle application, Microwave Opt Technol Lett 48 (2006), 2408 2412. 4. http://www.fcc.gov/dtv/, Digital Television, Major Initiatives of Federal Communications Commission. 5. C.M. Su, L.C. Chou, C.I. Lin, and K.L. Wong, Internal DTV receiving antenna for laptop application, Microwave Opt Technol Lett 44 (2005), 4 6. 6. K.L. Wong, Y.W. Chi, B. Chen, and S. Yang, Internal DTV antenna for folder-type mobile phone, Microwave Opt Technol Lett 48 (2006), 1015 1019. 7. http://www.ntia.doc.gov/osmhome/allochrt.html, U.S. Frequency Allocations, Office of Spectrum Management, National Telecommunications and Information Administration, USA. 8. K.L. Wong, Planar antennas for wireless communications, Wiley, New York, 2003. 9. http://www.ansoft.com/products/hf/hfss/, Ansoft Corporation HFSS. 10. http://www.zeland.com/, Zeland Software. 2007 Wiley Periodicals, Inc. Figure 8 Simulated radiation patterns at 700 MHz for the proposed antenna studied in Figure 3. [Color figure can be viewed in the online issue, which is available at www.interscience.wiley.com] DOI 10.1002/mop MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 49, No. 5, May 2007 1021
EFFECT OF SNR OF INPUT SIGNAL ON THE ACCURACY OF A RATIOMETRIC WAVELENGTH MEASUREMENT SYSTEM Ginu Rajan, Qian Wang, Gerald Farrell, Yuliya Semenova, and Pengfei Wang Applied Optoelectronics Centre, School of Electronics and Communication Engineering, Dublin Institute of Technology, Kevin Street, Dublin-8, Ireland Received 29 September 2006 ABSTRACT: The impact of a change in signal-to-noise ratio (SNR) of the input signal on the accuracy of a ratiometric wavelength measurement system is studied. The variation of the output ratio of the system due to the change in SNR of the input signal is modeled and the experimental investigation has been carried out, and both indicate that the accuracy is influenced significantly by variations in the SNR of the input signal. The demonstration example shows a wavelength shift of 0.055 nm when the SNR of the input signal varies from 47 db to 42 db (the measurable range is from 1500 nm to 1600 nm and the edge filter used has a slope of 0.15 db/nm). 2007 Wiley Periodicals, Inc. Microwave Opt Technol Lett 49: 1022 1024, 2007; Published online in Wiley Inter- Science (www.interscience.wiley.com). DOI 10.1002/mop.22328 Key words: ratiometric system; edge filter; signal-to-noise ratio (SNR) 1. INTRODUCTION Wavelength measurement is involved in many optical systems such as dense wavelength division multiplexing, and optical sensing based on fiber Bragg gratings (FBGs) [1, 2] where an accurate measurement of wavelength is needed. Among different wavelength measurement schemes, a ratiometric detection scheme [3 6] employing an edge filter has a simple configuration that uses the transition region of the filter s transmission response and converts wavelength measurement to a signal intensity measurement. The different edge filters used are bulk thin film filters [3], biconical fiber filters [4], fiber gratings [5], macrobend singlemode fiber [6], and so on. In principle a ratiometric scheme should allow for the measurement of the wavelength of an input signal without a dependency on other parameters of the input signal such as signal power or signal-to-noise ratio (SNR). In practice, the input signal has a limited SNR, and it has been proven in a previous investigation [7] that the SNR limits the slope of the edge filter and measurement range of the system. In this article the accuracy of a ratiometric system is investigated theoretically and experimentally, taking into account the effect of the SNR of the input signal, which indicates that variations in the SNR have a significant influence on the system s accuracy. 2. ANALYSIS OF THE IMPACT OF SNR ON RATIOMETRIC SYSTEM The schematic configuration of the ratiometric wavelength measurement system based on a macrobending fiber filter is shown in Figure 1. The input signal is split into two signals. One passes through a reference arm and the other passes through the bending fiber. The macrobending fiber acts as an edge filter as in Ref. [6]. Two photodetectors are placed at the ends of both arms. The system effectively operates as a discriminator, where the ratio of the power levels reaching the photodetectors is wavelength dependent. The wavelength of the input signal can be determined, assuming a suitable calibration has taken place, using the ratio of the electrical outputs of the two photodetectors [7], which is Figure 1 Schematic of a fiber bend loss ratiometric wavelength measurement system R 0 10log 10 S 1 0 I 0 T f d (1) S 2 0 0 I 0 d where S 1 ( ) and S 2 ( ) are the transmission responses of the arms of the splitter. T f ( ) is the transmission response of the edge filter and I 0 is the spectrum of the input signal. In practice, the light from a tunable laser or reflected by an FBG element has a limited SNR. Assuming for convenience a source with a power of 0 dbm at the peak wavelength, then as in Ref. [7], the spectrum of such an input signal can be modeled as 10 log 10 I 0 log 10 10 exp 4ln2 0 2 2, 0 0 S Rand Rs, 0 S is the SNR for the source in db. To describe the random fluctuations in the noise floor of the optical source, the term Rand Rs is used, where Rand is a random number (between 1 and 1) and Rs is a parameter in db which dictates the peak fluctuation in the SNR and is dependent on the nature of the source. Ù is a parameter which is determined by the noise level and can be determined for a source with a given SNR from the relationship 10 log 10 exp 4ln2 2 0 2 S. Using the above equations a system with an edge filter with a discrimination range of 25 db and a 3 db coupler that splits the input signal is modeled for different SNR values, for a wavelength range 1500 1600 nm. The modeled results are shown in Figure 2 and it can be seen that different SNRs produce a discrimination characteristic which is not the linear ideal. In practice, when the SNR of the input signal changes, the measured wavelength will be an error where the original calibration curve designed for a different SNR is used. Figure 3 shows a numerical example of the influence of SNR on output ratio for a fixed wavelength. Assume the wavelength of the input signal is 1520 nm and the slope of the edge filter is 0.25 db/nm. When the SNR decreases from 50 to 45 db, the calculated ratio variation is 0.0042 db, which would leads to a wavelength error of 0.017 nm. Thus it is seen that for a system, SNR changes will have a significant impact on the wavelength accuracy of the system. (2) 1022 MICROWAVE AND OPTICAL TECHNOLOGY LETTERS / Vol. 49, No. 5, May 2007 DOI 10.1002/mop
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