Design Consideration of Half-Bridge LLC Resonant Converter

Transcription

1 Design Consideration of Halfbridge LLC Resonant Converter 13 JPE 71 Design Consideration of HalfBridge LLC Resonant Converter HangSeok Choi airchild Korea Seiconductor, Korea ABSTRACT LLC resonant converters display any advantages over the conventional LC series resonant converter such as narrow frequency variation over wide range of load and input variation and zero voltage switching even under no load conditions. This paper presents analysis and design consideration for the half bridge LLC resonant converter. Using the fundaental approxiation, the gain equation is obtained, where the leakage inductance in the transforer secondary side is also considered. Based on the gain equation, the prtical design procedure is investigated to optiize the resonant network for a given input/output specifications. The design procedure is verified through an experiental prototype of the 115W halfbridge LLC resonant converter.. Keywords: Halfbridge resonant converter, zero voltage switching (ZVS) 1. Introduction The Conventional PWM technique processes power by controlling the duty cycle and interrupting the power flow. All the switching devices are hardswitched with abrupt changes of currents and voltages, which results in severe switching losses and noises. Meanwhile, the resonant technique process power in a sinusoidal for and the switching devices are softly coutated. Therefore, the switching losses and noises can be draatically reduced. or this reason, resonant converters have drawn a lot of attentions in various applications [13]. Aong any resonant converters, the halfbridge LLCtype resonant converter has been the ost popular topology for any applications since this topology has any advantages over other topologies; it can regulate the output over wide line Manuscript received July. 18, 006; revised Sep. 1, 006. Corresponding Author: hschoi@fairchildsei.co.kr Tel: , ax: Power Conversion Tea, airchild Korea Seiconductor and load variations with a relatively sall variation of switching frequency, it can hieve zero voltage switching (ZVS) over the entire operating range, and all essential parasitic eleents, including junction capitances of all seiconductor devices and the leakage inductance of the transforer, are utilized to hieve softswitching. While uch research has been done on the LLC resonant converter topology ever since its introduction in the 1990s [4], ost of the research has been focused on the steady state analysis rather than prtical design consideration. In [5], the low noise features were ainly investigated and no design procedure was studied. Reference [6] discussed the above resonance operation in buck ode only, where LLC topology was introduced as LCL type series resonant converter. In [7], detailed analysis was done on the buck ode operation as well as on the boost ode operation, but no design consideration was given. References [81] investigated LLC topology in detail using fundaental approxiation, but siplified the AC equivalent circuit by ignoring the leakage inductance

2 14 Journal of Power Electronics, Vol. 7, No. 1, January 007 in the secondary side. In general, the agnetic coponents of the LLC resonant converter are ipleented with one core by utilizing the leakage inductance as the resonant inductor. Consequently, the leakage inductance exists not only in the priary side but also in the secondary side. Since the leakage inductance in the secondary side affect the gain equation, ignoring the leakage inductance in the secondary side results in an incorrect design. This paper presents design consideration for the half bridge LLC resonant converter. Using the fundaental approxiation, the gain equation is obtained, where the leakage inductance in the transforer secondary side is also considered. Based on the gain equation, the prtical design procedure is investigated to optiize the resonant network for given input/output specifications. The design procedure is verified through an experiental prototype converter of the 115W halfbridge LLC resonant converter.. Operation Principle and undaental Approxiation appears in the resonant network. The filtering tion of the resonant network allows us to use the classical fundaental approxiation to obtain the voltage gain of the resonant converter, which assues that only the fundaental coponent of the squarewave voltage input to the resonant network contributes to the power transfer to the output. The current is lagging the voltage applied to the resonant network (that is, the fundaental coponent of the square wave applied by the halfbridge tote pole), which enables the MOSETs to be turned on with zero voltage. Q1 Q I ds I p L lkp I L Np:Ns Cr ig. 1 A scheatic of halfbridge LLC resonant converter I D R o V O The priary side stage of LLC resonant converter can be built as a fullbridge or halfbridge type and the output stage can be ipleented as a fullbridge or center tapped rectifier configuration with capitive output filter. ig. 1 shows the halfbridge ipleentation of the LLC resonant converter with fullbridge output rectifier, where L is the agnetizing inductance and L lkp and are the leakage inductances in the priary and secondary, respectively. ig. shows the typical wavefors of the LLC resonant converter. Operation of the LLC resonant converter is siilar to that of the conventional LC series resonant converter. The only difference is that the value of the agnetizing inductance is relatively sall. Thus the resonance between L L lkp and C r affect the converter operation. Since the agnetizing inductor is relatively sall, there exists considerable agnetizing current (I ) as shown in ig.. The halfbridge tote pole coposed of Q 1 and Q applies a square wave voltage ( ) to the resonant network. Since the resonant network has the effect of filtering the higher haronic voltages, essentially, a sinusoidal current I ds I D V gs1 V gs I p I ig. Typical wavefors of halfbridge LLC resonant converter It is iportant to note that the equivalent load resistance

3 Design Consideration of Halfbridge LLC Resonant Converter 15 shown in the priary side is different fro tual load resistance. ig. 3 shows how this equivalent load resistance is derived. The priary side circuit is repled by a sinusoidal current source, I and a square wave of voltage, V RI appears at the input to the rectifier. Considering the transforer turns the ration (nn p /N s ), the equivalent load resistance shown in the priary is obtained as R 8n R (1) o I I V RI pk I V o I o I V O pk I Io sin( wt) 4Vo V RI VRI sin( wt) V RI VRI 8 V 8 R R I I o Io R o ig. 3 Equivalent Load resistance R C r C r L lkp L 8n nn p /N s R R L lkp o L V RO N p :N s n R o o V O R o nv RI ig. 4 AC equivalent circuit for the LLC resonant converter With the equivalent load resistance obtained in (1), the charteristics of the LLC resonant converter can be derived. Using the AC equivalent circuit of ig. 4, the voltage gain is obtained as nv M V ω LR C O r in ω ω jω L n Llks R ωo ωp (1 ) ( ) (1 ) where R 8n R L L L L L L n L 1 1 ωo, ωp LC LC,, //( ) o p lkp r lkp lks r r p r As can be seen in (), there are two resonant frequencies. One is deterined by L r and C r while the other is deterined by L p and C r. In tual transforer, L p and L r can be easured in the priary side with the secondary side winding open circuited and short circuited, respectively. Iportant feature that should be observed in () is that the gain is fixed at resonant frequency (w o ) regardless of the load variation, which is given as ω ωo L L n L L L L lks p r Without considering the leakage inductance in the transforer secondary side, the gain in (3) becoes unity. In the previous research, the leakage inductance in the transforer secondary side was ignored to siplify the gain equation [71]. However, as observed, there exists considerable error when ignoring the leakage inductance in the transforer secondary side, which generally results in nonoptiized design. By assuing that L lkp n, the gain in () can be siplified as ω k ( ) nv ωp k 1 O (4) M Vin ω ω ( k 1) ω j( ) (1 ) Q (1 ) ω ω k 1 ω o o p () (3) where L k L (5) lkp

4 16 Journal of Power Electronics, Vol. 7, No. 1, January 007 Lr/ Cr Q (6) R The gain at the resonant frequency (w o ) is also siplified as ω ωo L n L L lks p k 1 L L L k p r By using the gain at the resonant frequency of (7) as a virtual additional gain of the transforer, the AC equivalent circuit of LLC resonant converter of ig. 4 can be siplified in ters of L p and L r as shown in ig. 5. C r C r L L L n L Llkp L // Llkp L L L L r L lkp L p L r L r lkp //( lks ) p lkp n:1 1: p V RI r Lp L L R V O nv RI R o (7) ig. 5 Siplified AC equivalent circuit for LLC resonant converter The equation (4) is plotted in ig. 6 for different Q values with k5 and f o 100kHz. As observed in ig. 6, the LLC resonant converter shows nearly load independent charteristics when the switching frequency is around the resonant frequency. This is a distinctive advantage of the LLCtype resonant converter over conventional seriesresonant converters. Therefore, it is natural to operate the converter around the resonant frequency to iniize the switching frequency variation at light load conditions. Gain f p LLC resonant Converter f o L L M L Q 1 Q 0.8 Q 0.6 Q 0.4 Q freq (khz ) ig. 6 Typical gain curves of the LLC resonant converter The operation range of the LLC resonant converter is deterined by the available peak voltage gain. The frequency where the peak gain is obtained exists between f p and f o as shown in ig. 6. As Q decreases (as load decreases), the peak gain frequency oves to f p and higher peak gain is obtained. Meanwhile, the peak gain frequency oves to f o and the peak gain drops as Q increases (as load increases). Thus, the full load condition should be the worst case for the resonant network design. Another iportant ftor that deterines the peak gain is the ratio between L and L lkp which is defined as k in (5). Even though the peak gain at a given condition can be obtained by using the gain in (4), it is difficult to express the peak gain in explicit for. Moreover, the gain obtained fro (4) has soe error at frequencies below the resonant frequency (f o ) due to the fundaental approxiation. In order to siplify the analysis and design, the peak gains are obtained using siulation tool and depicted in ig. 7, which shows how the gain varies with Q for different k values. It appears that higher peak gain can be obtained by reducing k or Q values. With a given resonant frequency (f o ), decreasing k or Q eans reducing the agnetizing inductance, which results in increased circulating current. Accordingly, there is a tradeoff between the available gain range and conduction loss. lkp

5 Design Consideration of Halfbridge LLC Resonant Converter 17.4 I p I (I) f s < f o..0 I D I O Peak Gain k1.5 k1.75 k k.5 k3 I p I (II) f s > f o k4 k5 k7 k9 I D I O Q ig. 7 Peak gain versus Q for different k values ig. 8 Wavefors of current in the transforer priary side and secondary sides for difference operation odes 3. Design Procedure Based on the previous analysis, the prtical design procedure is presented in this section. It discusses optiizing the resonant network for given input/output specifications. 3.1 Operation ode While the conventional LC series resonant converter always operates at a frequency above the resonant frequency, the LLC resonant converter can operate at frequency below or above the resonance frequency. ig. 8 shows the wavefors of the currents in the transforer priary side and secondary side. As can be observed, operation below the resonant frequency (case I) allows the soft coutation of the rectifier diodes in the secondary side while the circulating current is relatively large. Meanwhile, operation above the resonant frequency (case II) allows the circulating current to be iniized, but the rectifier diodes are not softly coutated. Thus, below resonance operation is preferred for high output voltage application where reverse recovery loss in the rectifier diode is severe. On the other hand, above resonance operation can show better efficiency for application where the output voltage is low and schottky diodes are available for the secondary side rectifiers since the conduction loss is iniize 3. Maxiu Gain The axiu gain should be deterined considering the input voltage variation. Since the peak gain takes ple when the converter operates at the boundary of zero voltage switching (ZVS) and zero current switching (ZCS) ode, ZVS condition is lost at the axiu gain condition. Therefore, soe argin is required when deterining the axiu gain. Based on the axiu gain, the proper Q and k values can be obtained fro ig. 7. While higher gain is obtained with sall k, too sall k value results in poor coupling of the transforer. It is typical to set k to be 5~10, which results in a gain of 1.1~1. at the resonant frequency. The value of k affects the losses of the converter. The ajor portion of the conduction loss is caused by the agnetizing current whose peak value is given by: I nv T (8) L 4 pk o s The value of k also affects the switching loss. Since the turnon switching loss is reoved by zero voltage switching operation, the turnoff switching loss is doinant. The turn off switching loss is proportional to the turnoff current, which is the sae as the peak agnetizing current of (8). Therefore, agnetizing current should be iniized for high efficiency.

6 18 Journal of Power Electronics, Vol. 7, No. 1, January 007 Vin0 ~ 70V C DL 0u Vcc u U4005 Core : EER3541 Np5T (0.1 30, Litz wire) Ns1Ns7T ( , Litz wire) Sectional winding Lp800uH (Measured with secondary side open) Lr00uH (Measured with secondary side short) AN7770 QP13N50C Q1 D1 YP010DN Vo 5V/ 4.6A Cr 18n Q QP13N50C R sense L lkp 107uH L 713uH Np Ns Ns Ro D YP010DN KA431 C R ig. 9 Scheatic of the prototype converter 4. Experiental Verification In order to show validity of the previous analysis and design consideration, an experiental prototype converter of the 115W halfbridge LLC resonant converter has been built and tested. The scheatic of the converter and circuit coponents are shown in ig. 9. The input voltage is 0V~70V and the output is 5V/4.6A. In ters of DC voltage, the input voltage is 60~380V considering holdup tie. The ratio (k) between L and L lkp is deterined as 6.5, which results in the gain at the resonant frequency as ω ω o k (9) k As observed in the previous analysis, there is a tradeoff between the available gain range and conduction loss. Since the input voltage varies over wide range, if the converter is designed to operate only below resonance frequencies, the excessive circulating current can deteriorate the efficiency. Thus, the converter is designed to operate above resonance at high input voltage conditions and below resonance at low input voltage to iniize the conduction loss caused by circulating current. The iniu gain at full load is deterined as 1.0. With the iniu gain, the transforer turn ratio is obtained as ax Np Min Vin / 1 380/ n 7.4 N ( V V ) (5 0.7) s o (10) where V is the diode forward voltage drop. The axiu gain to cover the input voltage variation is 380/ With 10% argin, axiu gain of 1.6 is required. ro the gain curves in ig. 10, Q is obtained as 0.4. By selecting the resonant frequency as 85kHz, the resonant network is deterined as L 713uH, L lkp 107uH (L p 800uH, L r 00uH) and C r 18n. The transforer is ipleented with a sectional winding ethod to increase the leakage inductance as depicted in ig. 9. Peak Gain Q ig. 10 Design gain curves k3 k4 k5 k6.5 k9 ig. 11 and 1 show the operation wavefors for full

7 Design Consideration of Halfbridge LLC Resonant Converter 19 load condition with input voltage of 0V and 70V, respectively. ig. 13 and 14 show the operation wavefors for no load condition with input voltage of 0V and 70V, respectively. As can be seen, ZVS is hieved for entire input/output range. ig. 15 shows the easured efficiency. As expected, efficiency decreases for low input voltage condition due to the increased circulating current. ig. 16 shows the easured switching frequency with load and input voltage variation. ig. 17 shows the gain curves for different load conditions. As can be seen, the switching frequency variation is well atched with the gain curves of ig. 17 when the switching frequency is close to the resonant frequency. ig. 13 Operation wavefors : 0put and no load C1: gate drive signal (0V/div), C3: Drain voltage (00V/div) C4: Drain current (1A/div) ig. 11 Operation wavefors : 0put and full load C1: gate drive signal (0V/div), C3: Drain voltage (00V/div) C4: Drain current (1A/div) ig. 14 Operation wavefors : 70put and no load C1: gate drive signal (0V/div), C3: Drain voltage (00V/div) C4: Drain current (1A/div) Efficiency 94.00% 9.00% 90.00% Eff (%) 88.00% 86.00% 84.00% 0V 50V 70V 8.00% ig. 1 Operation wavefors : 70put and full load C1: gate drive signal (0V/div), C3: Drain voltage (00V/div) C4: Drain current (1A/div) 80.00% 0.0W 0.0W 40.0W 60.0W 80.0W 100.0W 10.0W Output Power ig. 15 Measured efficiency

8 0 Journal of Power Electronics, Vol. 7, No. 1, January 007 Switching freq (khz) Gain ig. 16 Measured switching frequency variation LLC resonant Converter Vin380Vdc (70V) requency 0V 50V 70V 0 0.0W 0.0W 40.0W 60.0W 80.0W 100.0W 10.0W Output Power freq (khz) Vin300Vdc (0V) 100% load 80% load 60% load 40% load 0% load Vin335Vdc (50V) ig. 17 Gain curves and operating range 5. Conclusions This paper has presented design consideration for the LLC resonant converter utilizing the leakage inductance and agnetizing inductance of the transforer as resonant coponents. The leakage inductance in the transforer secondary side was also considered in the gain equation. The design procedure was verified through experiental results. References [1] Robert L. Steigerwald, A Coparison of Halfbridge resonant converter topologies, IEEE Transtions on Power Electronics, Vol. 3, No., April [] A.. Witulski and R. W. Erickson, Design of the series resonant converter for iniu stress, IEEE Transtions on Aerosp. Electron. Syst., Vol. AES, pp , July 1986 [3] Y. G. Kang, A. K. Upadhyay, D. L. Stephens, Analysis and design of a halfbridge parallel resonant converter operating above resonance, IEEE Transtions on Industry Applications Vol. 7, MarchApril 1991 pp [4] Yasuhito urukawa, Kouichi Morita, Taketoshi Yoshikawa A High Efficiency 150W DC/DC Converter, Intelec 1994, pp [5] Koichi Morita Novel Ultra Lownoise Soft switchode Power Supply, Intelec 1998, pp.1151 [6] Ashoka K. S. Bhat, "Analysis and Design of LCLType Series Resonant Converter," IEEE Transtions on Industrial Electronics, Vol. 41, No. 1, eb [7] J.. Lazar and R. Martineli "Steadystate analysis of the LLC series resonant converter, APEC 001, [8] Yan Liang, Wenduo Liu, Bing Lu, van Wyk, J.D, " Design of integrated passive coponent for a 1 MHz 1 kw halfbridge LLC resonant converter," IAS 005, pp. 38 [9] B. Yang,.C. Lee, M. Concannon,"Over current protection ethods for LLC resonant converter," APEC 003, pp [10] Yilei Gu, Zhengyu Lu, Lijun Hang, Zhaoing Qian, Guisong Huang, "Threelevel LLC series resonant DC/DC converter," IEEE Transtions on Power Electronics Vol.0, July 005, pp [11] Bo Yang, Lee,.C, A.J Zhang, Guisong Huang, "LLC resonant converter for front end DC/DC conversion," APEC 00. pp [1] Bing Lu, Wenduo Liu, Yan Liang, red C. Lee, Jobus D. Van Wyk, Optial design ethology for LLC Resonant Converter, APEC 006. pp HangSeok Choi Hangseok Choi received the B.S., M.S. and Ph.D degrees in electrical engineering fro Seoul National University, in 1996, 1999 and 00, respectively. He is currently working for airchild Seiconductor in Bucheon, Korea as a syste and application engineer. His research interests include high frequency power conversion, and odeling and control of converters. He has published 14 papers in IEEE conferences and transtions and 10 application notes in airchild seiconductor.