Circuit description BTR18 by Peter Solf DK1HE. 1. Receiveer unit: 1.1 Receiver input
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1 Circuit description BTR18 by Peter Solf DK1HE 1. Receiveer unit: 1.1 Receiver input The received signal coming from the antenna socket Bu4 passes through the transmitter low-pass filter L2/L3 and reaches the high point of the RX- filter Fi1/C4 via the coupling capacitor C1. In order to achieve a high circuit quality and thus good preselection, the coupling to the antenna as well as the inductive decoupling of the received signal to the following mixer input IC1 is done loosely. The PA transistor (T14) is inactive in receive mode and has no attenuating effect on the antenna signal. We need the protective circuit T6/D2/C3 for transmit operation. In receive operation, the switching transistor T6 is blocked and the protective diode D2 is also in blocking operation via R5 with +6V at the cathode. Intermodulation effects at high reception voltages are thus safely avoided. If the BTR18 is switched to transmission, T6 and D2 become conductive. The very high HF voltage coming from the low-pass filter and coupled in via C1 is largely short-circuited via the now conducting D2 and C3. The HF is further attenuated to RX level via automatic regulation of the control stage (see description control stage) and then made audible as a monitor signal like a received signal via the further reception path. 1
2 1.2 VFO tage at its source is set to approx. 1 Vpp which is sufficient for triggering a frequency counter optionally connected via C43. The HF voltage at the divider resistor (R18) is about 300 mvpp and is fed to the receiver mixer IC1 via C7 and to the transmitter mixer via R19/44. The oscillator frequency required for mixing the input frequency to the intermediate frequency is generated in a modified clapp circuit with T9. This circuit variant is characterized by a particularly high short- and longterm frequency constancy. Due to the large capacitance values of C38/C39, a change in the dynamic input capacitance of T6 only has a minimal effect on the generated frequency. With the antiserial capacitance diodes D3/D4, the circuit can be tuned by the required amount of the respective CW band segment (VCO Voltage Controlled Oscillator). L1 is mounted on a T50-6 toroidal core, which in combination with the Styroflex capacitors C38/C39 with negative temperature coefficient and the NP0 capacitor C36 forms a temperature-stable oscillating circuit. The beginning of the band can be fine-tuned with C37. In order to avoid feedback effects of the following stages on the VCO, they are coupled via the FET voltage follower T10. With P3 the HF output vol- 2
3 1.3 RX mixer, gain-control and quartz filter The symmetrical outputs (pin 4/5) of the receiving mixer IC1 are followed by a resonant circuit (Fi2/C9) tuned to the intermediate frequency with inductive coupling. The choosen transformation ratio of the two windings transforms the mixer output resistance (3K Ohm) to an impedance of 200 Ohm at the coupling winding, which corresponds to the input-side filter termination of the subsequent Cohn quartz filter (Q1 to Q4). The filter termination on the output side is formed by the input resistance of the IF post-amplifier T7 operating in gate circuit. It is also about 200 Ohm. In conjunction with the IF resonant circuit (Fi3/C15) acting as the working resistance of T7 with inductive decoupling to the product detector IC2, this results in a step gain of about 20 db. The bandwidth of the quartz filter is mainly determined by the size of the capacitors. With the 150pF capacitors we use, we achieve a bandwidth of about 350 Hz. If a larger bandwidth is desired, the capacitors C9-C14 can be reduced in size or increased for even narrower filters. The complete gain of the receiving section can be adjusted to the respective reception situation by means of HF manual control. The gain is adjusted by changing the total collector current in the differential amplifiers of the Gilbert cell contained in IC1. T4 and T5 form a current bank, i.e. a current impressed via R1 or R2 into the reference diode (T4) appears 1:1 (current mirror circuit) as collector current in T5, which influences the gain of the mixer cell inversely proportional via the bias generation in IC1. With P1 the amplification of the circuit so can be adjusted. For an optimal function of the current bank it is important that the transistors involved have identical parameters. This requirement is achieved by using SMD types from the same belt section for T4,T5. In transmit mode, current is set via P2/T3 independent of the manual control setting. 6V reduces the transistor T3 by means of the voltage divider P2 and thus influences the current I1. P2 defines the listening tone strength in transmit mode. 3
4 1.4 Productdetector and Audioamplifier In the product detector IC2, the amplified IF signal is mixed with the BFO to the AF level. The internal oscillator working as a BFO oscillates with Q5 in Colpitts circuit by about 650Hz offset from the center frequency of the quartz filter. The frequency can be fine-tuned using C18. The AF signal at the balanced outputs Pin4/5 of IC2 is cleared of IF residues with C22/C25 and fed symmetrically via the coupling capacitors C23/C24 to the inputs (Pin2/3) of the connected audio amplifier IC3, where it is amplified to headphone volume. If the PK4 keyer is installed, the jumper (J1) is plugged in. The programming process output in Morse code when programming the PK4 keyer is switched through to the NF part and can thus be followed acoustically. 4
5 2. Transmitter: 2.1 Keying the transmitter constant has elapsed, that means the rising edge of the transmitter keying voltage is soft keyed. When the keying is finished, the drain of T15 changes to +Ub potential. The gate of T16 follows the voltage jump again with a time constant of about 5mSec via the series connection of R33/R34 since the C77 charged with reversed polarity must be reloaded into the old state. The output voltage of the keying stage at R33 (+U TX) therefore continuously reaches ground potential after the time constant has elapsed. That means the decay edge of the transmitter keying voltage is also soft keyed. From the transmitter push-button voltage (+U TX), a further push-button voltage (+6V TX) for supplying the transmitter mixer (IC4) as well as for generating the gate bias voltage for the PA transistor (T14) is obtained with the aid of the R20/D5/T11 limiter circuit. the PK4 keyer is installed and key is pushed to transmit, T15 becomes conductive and switches to ground potential via R33/R34. If no PK4keyer is installed, the keydown of an external keyer or a hub key through the direct connection to R33/R34 causes them to be connected to ground potential. The gate of T16 (P-channel Mosfet) follows the voltage jump via R34 with a time constant of about 5 Milliseconds, since the previously charged C77 has to discharge via R34. The output voltage of the keying stage at R33 (+U TX) therefore continuously reaches its maximum value = Ub after the time 5
6 2.2 TX Mixer, Impedance converter, Driver, PA step gain of about 26dB. The base voltage divider R22/R23 and the series resistor R21 stabilize the operating point of the Darlington stage at a collector current of about 30mA in T13. The HF voltage present on the secondary side of Tr1 is fed to the gate of the RF power mosfet T14. The gate series resistor R28 prevents parasitic oscillations in the VHF/UHF range. By means of P4, the drain current angle and thus the HF output power of the stage can be adjusted via a variable DC bias voltage. The 1:4 output transformer (Tr2) transforms the dynamic output resistance from T14 (about 12 Ohm at Ub=10V; Pout=4Watt) to the 50 Ohm level. C57 is used for frequency compensation of Tr2 on the higher bands. DThe transmit frequency is formed by mixing the VFO signal with a quartz frequency that corresponds to the center frequency of the RX crystal filter in the transmitter mixer (IC4). The crystal oscillator oscillates with Q6 in Colpitts circuit of the IC-internal oscillator stage. The frequency can be fine-tuned by means of C47. The symmetrical outputs (pin 4.5) of IC4 are followed by a 2-circuit bandpass filter (Fi4,C51,Fi5,C53,C54) which is capacitively coupled to C52 and filters out the useful transmit frequency from the output spectrum of the mixer. The supply voltage of IC4 is +6V TX. The selected transmit signal is decoupled from the voltage divider (C53,C54) of the band filter secondary circuit and fed to the impedance converter (T12). Thanks to the high-impedance input resistance of T12 and the loose inductive coupling of the primary circuit to the transmitter mixer, the filter can offer high operating qualities of its individual circuits and the resulting good selection of the wanted signal The low impedance output of T12 is followed galvanically by the driver with T13. The stage is linearly countercoupled via R25. The working resistance is formed by the 1:1 transformer (Tr1) which is loaded with R27 (220 Ohm) on the secondary side. In conjunction with R25, this results in a broadband 6
7 2.3 SWR probe und display The 50 Ohm secondary side of Tr2 is followed by a 5-pole low-pass filter with an additional attenuation pole formed by L2 and C63 on the 2nd harmonic of the operating frequency. This configuration allows sufficient attenuation values of the harmonics to be achieved with little filter effort. Between the TX output filter and the antenna socket (Bu4) there is a SWR measuring device which provides important information about the respective antenna matching, especially for portable operation. Tr3 in combination with D6/D7 supplies proportional directional voltages to the forward and backward flow, which are supplied to the bases of T17 and T18 respectively. R47/R50 serve as working resistors for the two rectifier diodes. C70/C71 eliminate HF residues. In the collector circuit of T17/T18 there is a light emitting diode (D8,D9) which signals the ratio of power running forward to power running backward by its luminosity. The emitter resistors R48/R49 define the maximum LED current at a given TX output power for an optically linear display. 7
8 3. Other parts of the circuit: D11 is activated (the green LED continues to burn); a yellow mixed colour is produced. At 10% battery capacity (empty battery + reserve) U Batt/2 falls below the threshold of Urot (+4.8V), i.e. Pin1 of IC6 changes to low and switches off the green LED in D11; only the red diode system lights up and signals AKKU LEER. The voltage regulator (IC7) supplies all voltage-relevant stages in the device with a stable +6V operating voltage. The built-in battery can be charged via the charging socket (Bu3); F1 (1A medium delay) protects the battery from excessive charge/discharge currents. When the battery is removed, the device can be supplied with external voltage (9 to 12V) via Bu3. D10 serves as reverse polarity protection; F1 is not effective! The OPV s IC6-1/IC6-2, operating as comparators, are used to monitor the current state of charge of the battery installed in the device (8 pcs NiMH 1.2V= 9.6V). Via the divider chain R36/R37/R38, the switching points for 50% battery capacity or 10% (empty) are obtained from the stable +6V reference voltage supplied by IC7. The divider voltage values are based on the discharge curve of NiMH ENELOOP AA cells with 1900mAh. Since the reference voltage is only 6V, but the maximum cell voltage is considerably higher (11 Volt), the divider resistors are dimensioned so that exactly half of the switching point voltage value is set. For the same reason, the battery voltage to be measured is halved via the voltage divider R39/R40. The outputs of the comparators are connected via R41 or R42 with a Duo-LED D11 (red,green). When the battery is fully charged (100%), U Batt/2 is above the switching points Uyelow and Ured. This results in Pin1 of IC6 goes to high and D11 glows green. At 50% battery capacity U Batt/2 falls below the threshold of Ugelb (+5.08V) with the result that Pin7 changes to high and the red LED in 8
9 Basic information about the receive branch of the BTR18 using the example of the 40m band (using adapted parts of an essay by Paul Harden, NA5N in FI s Workshop Primer in translation and editing by Ingo, DK3RED and Uwe, DL8SAI) The BTR18 RX is a single superhet RX with an intermediate frequency of 4.9 MHz. Gilbert cell mixers of type NE/SA602 are used as mixers. The NE602 is a balanced mixer that mixes the signal of our VFO with the RF signal at 7MHz. The output signal of the mixer contains desired and unwanted frequencies. The desired frequency in our example would be the intermediate frequency 4.9MHz. This occurs when the desired signal at 7.0MHz is mixed with the VFO 2.1MHz (7-2.1=4.9). In practice, however, our VFO generates harmonic oscillations (e.g. 4MHz, 6MHz, 8MHz,10MHz). Each of these harmonics is also mixed with the input frequencies and thus generates a number of other unwanted frequencies. On a spectrum analyzer, the output spectrum at terminals 4 and 5 looks so jammed that one is surprised that this receiver works at all. The required IF frequency is rarely the dominant frequency. This is the case with most mixers, not just the NE602. If we look at our example on a spectrum analyzer, we can see the following frequencies at the output of the mixer: 11.9 MHz Sum HF oscillator (7 MHz +4.9 MHz) 2.1 MHz VFO frequency 2.8 MHz Mixed product HF - 2nd harmonic of the VFO frequency (7MHz -4.2MHz) 4.2 MHz 2nd harmonic of the VFO frequency 4,9 MHz the desired IF (difference HF - VFO) 6.3 MHz 3rd harmonic of the VFO frequency 9.1 MHz Sum HF+VFO (7MHz + 2.1MHz) Such mixed products can be detected up to the 50 MHz range. Our goal is to receive a CW station with a bandwidth of 500Hz or less cleanly. However, the calculated spectrum clearly shows us that the desired signal is only a small part of the total power coming from the mixer. From this we can see that we have to filter out everything except the small 500 Hz segment we are interested in and also amplify the -80 dbm signal to +10 dbm so that it is clearly audible as an AF signal. The fact that the signal we are interested in is always at 4.9 MHz (the IF) helps us here, no matter what reception frequency we have just set. This is an advantage of the superhet circuit. Since we receive extremely weak signals in the µv range in shortwave amateur radio, we have to amplify them extremely to make them audible at the headphone output. The amplifiers we use for this are usually very broadband. This means that they amplify everything that is offered to them for amplification. In order to process the 500 Hz narrow signal on the 4.9 MHz IF in such a way that it is the only signal that can be heard at the headphone output, we have to convert these broadband amplifiers into narrowband amplifiers. In the BTR18, this is achieved by a combination of a low-pass filter, 2 bandpass filters and a quartz filter, with which selectivity is first achieved by suppressing unwanted components and then the desired amplification factor by amplifiers with correspondingly high amplification. The first preselection takes place in the low-pass filter of the transmitter. The cut-off frequency is just above 8 MHz, so that signals > 8 MHz coming from the antenna are only transmitted very attenuated. Via C1, the HF < 8 MHz is loosely coupled to the FI1 filter, which makes a further selection. FI1 is an LC resonant circuit, also known as a bandpass. At its resonant frequency of 7.02 MHz, the signal has an energy maximum. All other frequencies, however, are attenuated. Its passband curve corresponds to a narrow bell curve ; the higher the quality of the resonant circuit, the narrower the passband. With an operating quality Q of about 100 the bandwidth is about 70 khz (b = f0/q) which helps the following mixer to stay in the linear range because all incoming signals except those in the 7MHz range are already strongly attenuated. The mixer is followed by another bandpass, Fi2, which is resonant on the intermediate frequency. In contrast to many other NE602 based QRP devices we use the balanced output of the BTR18, which brings about 6dB gain at the mixer gain. Fi2 attenuates the unwanted mix products above and below the IF, which is good for the far-away selection of the quartz filter. At the same time, the high 9
10 output impedance of the mixer (3kOhm) is adjusted to the low input impedance of the Cohn filter. The quartz filter (Cohn filter, ladder filter) As we know, the equivalent circuit of a quartz crystal is an LC series resonant circuit. The frequency determining components are Lm and Cm, which are often referred to as dynamic parameters. Rs is a series resistor and embodies the occurring losses. The static parallel capacitance Cp results from the capacitance of the connecting electrodes as well as the holding and stray capacitances within the quartz oscillator housing. A quartz crystal has both a series and a parallel resonance. The distance between fs and fp for filters should preferably be above 3 khz. Since the main task of an IF filter is to achieve a narrow passband with the lowest possible attenuation, it is possible to deduce why quartz filters work on the series resonance of the crystals. With series resonance, XLm = XCm applies, i.e. inductive and capacitive reactance compensate each other, so that only the loss resistance Rs is effective. This is an important feature for filters, since Rs leads to an attenuation of the signals passing through the filter. This is called insertion loss and is theoretically about 1 db per crystal. In practice, however, an insertion loss of approximately 4-6 db must be expected for the 4-pole quartz filters in QRP devices (and goodbye, antenna gain). If possible, you should therefore choose crystals with a loss resistance Rs below 100 Ohm. Rs is therefore included in most catalogues. Another important parameter of a quartz is its tolerance. Even with four identical crystals, the exact resonance frequencies differ slightly from each other. The quartz crystals for the filter of the BTR18 are therefore used on +/- 20 Hz paired low profile quartz crystals, which have a considerably higher quality and lower loss resistances due to their internal construction. components is achieved. Secondly, you only get a clear signal from one station on one side of the beat zero if you place the BFO frequency exactly on the steep slope. Without the suppression of the upper sideband, the signal would be heard above and below the beat zero, which would always give the 50:50 possibility of calling the other station at the wrong frequency. Together with the crystals Q1 - Q4 the capacitors C9 - C14 form - C48 such a ladder filter. Their capacitance values are normally experimentally, or by rather complex calculations. With the values given in the construction folder, we get an IF bandwidth of about 350Hz. If you want to make the filter wider, you can try smaller capacitor values. The signal coming from the quartz filter is amplified in the IF amplifier transistor T7 and fed to the product detector via the filter Fi3. Many QRP devices use the NE602 as a product detector. It works analog to the NE602 in the IF mixer described above. The input signal is the 4.9 MHz IF, the integrated oscillator, which represents the BFO, is adjusted to 4.9 MHz minus 650 Hz, resulting in a difference signal of 650 Hz at the output. Without this offset of the BFOs around 650 Hz would result in an output signal of zero Hz. With the offset, the CW signal is heard as a 650 Hz tone if the received station is exactly in the middle of the passband of the IF filter. At the output of the product detector we find both the AF signal and the IF of 4.9 MHz. The IF signal is blocked by filters in the AF stage and derived from ground via C22/C25. In the NF amplifier, the selected signal is amplified to headphone volume. The most common filter is the ladder filter, in which the crystals are connected in series. (The name is based on the ladder like arrangement of the components in the circuit diagram). The function of this filter is based on an essential peculiarity of the crystals: in the frequency range between series and parallel resonance (fs - fp) they behave like a high inductance. By connecting the crystals with capacitors, so that XC compensates just XL, a very steep edge is created above the series resonant frequency, which strongly attenuates the upper sideband of a signal, while the lower sideband is let through. This filter is therefore often referred to as an LSB filter. It has two advantages for the QRP receiver. First of all, the bandwidth is reduced by almost 50%, i.e. a high selectivity with relatively few 10
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