國立宜蘭大學電子工程學系 ( 研究所 ) 碩士論文. Department of Electronic Engineering. National Ilan University. Master Thesis

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1 國立宜蘭大學電子工程學系 ( 研究所 ) 碩士論文 Department of Electronic Engineering National Ilan University Master Thesis 用於超寬頻之平面帶拒開槽天線分析與設計 Design and analysis of planar band-notched slot antennas for UWB applications 指導教授 : 邱建文博士 Chien-Wen Chiu, Ph. D. 研究生 : 李家珊撰 Chia-Shan Li 中華民國九十七年七月

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5 誌謝 首先誠摯的感謝指導教授邱建文副教授, 兩年的相處, 深深覺得老師非常的替學生著想, 還記得, 當我剛進入宜大還未清楚確定研究的方向時, 老師問了我 將來是否有要繼續攻讀博士亦或是畢業後就進入職場就業, 這兩者對兩年的碩士班來說, 所需要學習的加強的不太一樣, 一個要栽培的是富有微波 電磁較深入理論的博士, 另一個則是具備基本微波知識 熟悉軟硬體應用的工程師, 這些對於一個研究生來說是很重要的起點, 就是這樣的一段話讓我了解到老師的與眾不同, 兩年來老師對我未來所選擇的方向, 悉心的教導使我得以對微波以及天線設計領域有了深入的了解, 也對此更產生了興趣, 在碩士研究生的生活中, 不時的花費寶貴的時間與學生討論並指點我研究的方向, 使我在這看似短短的兩年中獲益匪淺, 衷心的感謝邱副教授的指導才讓本研究得以探討 其次要感謝口試委員鄧聖明教授與吳俊德教授, 在百忙之中可抽空指導並且給予寶貴的意見, 讓學生了解研究是永無止盡的, 使學生能洞悉往後更該加強學習的地方 接著要感謝一直跟我互相鼓勵扶持的好友佳芬 御廷 秀容, 同窗維昌 國志 翊博 宏宇 育生 偉晟 履安 嘉宏 秉恩 玉麟, 格致大樓的研究生活點滴, 相約午餐晚餐消夜的友誼, 碩一時一起修課 一起通宵熬夜趕報告 一起討論課業衝刺期中期末考, 大家秉持著一起努力修習學分一起順利畢業的無私心, 那股甘苦與共的熱血至今我還記憶猶新, 如今大家也都一同朝向人生另一個旅途邁進 對於去年剛畢業的智超學長還有同實驗室努力不懈的玉麟 白皙可愛的靜美學妹 實力不容小視的俞任學弟 有實驗室燈塔之稱的智翔學弟, 實驗室有你們的歡笑, 研究路上有你們的陪伴與協助, 讓我的無線通訊實驗室充滿著朝氣, 這一切都成為了我求學生崖中最珍貴的回憶 回首兩年的研究生崖, 不僅讓我在微波 天線領域的課業上學習到很多, 在心智上成長更多, 在面對研究上, 縱使不斷失敗也不能氣餒, 因為研究不是一下子就會有成果的, 沒有經過先前的失敗也不會有之後的成功, 我知道唯有不間斷的努力前進才是戰勝阻礙的唯一方法, 沒有完成不了的事, 只有提早放棄的失敗 人走過的每一步每一腳 i

6 印都是會留下痕跡的, 當你回頭時, 就會看見 這段話藉由兩年的經歷, 我深深感受到 最後, 我想感謝我摯愛的父母親, 因為有他們的開導, 我才能前往研究所這條路 ; 因為有他們的鼓勵, 我才能展開繼續努力的熱忱 ; 因為有他們的支持, 我才能不顧一切的學習所學 ; 因為有他們的關心, 我才能健健康康地成長, 有家人的陪伴讓我學會更堅強的面對一切的挑戰, 謹以此文獻給我的父親奕楨, 母親月娥, 姐姐婉瑜以及弟弟尚學, 願你們與我分享這份榮耀 家珊 28 年 6 月 18 日於宜大 感謝國家高速網路與計算中心提供 Ansoft HFSS 軟體資源, 使 本研究得以順利進行 ii

7 摘要 本論文旨在設計使用於超寬頻通訊系統之共面波導饋入平面槽孔天線 為使超寬頻系統可用於 3.1GHz~1.6 GHz 頻段, 首先提出利用圓形匹配殘段連接共面波導饋入之平面橢圓形槽孔天線設計, 並說明橢圓槽孔天線是利用多模態諧振觀念達到超寬頻能力之原理, 本論文除了量測驗證阻抗頻寬外, 所量測之輻射場型具有全向性, 証實此天線可提供手持式通訊設備使用 為了避免對現行 5~6 GHz 之無線區域網路系統產生干擾, 本論文接下來設計具有 5~6 GHz 帶拒功能之平面橢圓槽孔天線, 此帶拒平面天線是利用於圓形匹配殘段上切出四分之一波長開路之槽隙或於槽孔上內嵌入半波長寄生帶線來對 5~6 GHz 產生帶拒功能, 模擬與實驗量測結果顯示除產生帶拒外仍保有超寬頻性, 輻射場型仍具全向性 最後, 為了讓天線可以使用於手持行動通訊設備上, 本論文設計一個摺疊式尺寸縮小之改良式弧狀槽孔天線, 其體積減少一半但仍保有超寬頻特性, 實驗結果除了高頻下輻射場型有零點外, 此縮小化之改良式平面天線仍然可達超寬頻系統之頻寬要求 iii

8 Abstract In this thesis, various planar slot antennas fed by CPW are proposed for UWB communication-system applications. First, an elliptical slot antenna with a circular stub fed by CPW is presented and discussed why it has ultra-wide bandwidth based on multiple resonance concept. The antenna is fabricated and experimental results show that its impedance bandwidth can cover frequencies from 3.1 to 1.6GHz, which meets system requirements. The radiation pattern also confirms its omni-directional pattern features, which is suitable for portable device or laptop computer application. Then, a band-notched elliptical slot antenna is proposed to overcome interference with some already existing standards such as the IEEE82.11a and HyperLan/2A systems. This planar slot antenna using open-end slits on the tuning stub or a pair of parasitic strips in the aperture is studied and designed to reject this particular sub-band within 5-6GHz. Measured results show that this antenna not only retains ultra-wide bandwidth but also has band-rejection capability. Also, its radiation pattern still preserved omni-directional feature which meets our system requirements. Finally, a compact and novel folded slot antenna, which its size is shrunken to a half of the original curved-shape structure, is proposed for practical UWB applications. From experimental results, it is found that it still preserved UWB characteristics although its radiation pattern seems not perfect for ultra-wideband transmission. iv

9 Table of Contents 誌謝...i 摘要...iii Abstract...iv Table of Contents...v List of Figures...vii Chapter 1 Introduction Background and Motivation Literature Survey Chapter Outlines...6 Chapter 2 Design and Analysis of UWB Slot Antenna Slot Antenna for Broadband Applications Slot antenna fed by CPW/microstrip lines Rectangular /elliptical/circular slot antenna Multiple resonators concept Design of an Elliptical Slot Antenna Fed by CPW The lower edge of the impedance bandwidth Impedance matching by a tuning stub Design parameter analysis Simulation and Measurement Results Voltage standing wave ratio (VSWR) and return loss Radiation pattern and gain Transfer Function and Time Domain Analysis Normalize transfer function...19 v

10 Guassian pulse for mask regulation Time Domain analysis Simulation results and measurement Summary...24 Chapter 3 Band-Notched UWB Slot Antenna Fed by CPW Geometry and Band-notch design UWB slot antenna band-notched using open-end slits UWB slot antenna band-notched using parasitic strips Simulation and Experimental Results Return Loss and voltage standing wave ratio Radiation pattern Gain Summary...44 Chapter 4 Folded UWB Antenna Edge-Fed Folded-CPW Design for an edge-fed transmission line Simulation and experimental Results The Novel Folded UWB Antenna Antenna configuration Summary...68 Chapter 5 Conclusions...83 References...84 vi

11 List of Figures Fig.1.1 FCC indoor emission spectrum mask....7 Fig. 2.1 Geometry of a CPW-Fed rectangular slot antenna using a folk-shape stub...25 Fig. 2.2 Geometry of a CPW-fed printed elliptical slot antennas using a U-shape stub Fig. 2.3 Field distributions on the slot region...27 Fig. 2.4 The equivalent solid cylinder Fig. 2.5 Configuration of an elliptical aperture antenna...28 Fig. 2.6(a) Geometry of the proposed UWB antenna...29 Fig. 2.6(b) its photograph Fig. 2.7 S 11 of the proposed UWB antenna with different value A....3 Fig. 2.8 Smith chart for the input impedance....3 Fig. 2.9 S 11 of the proposed UWB antenna with different value a...31 Fig. 2.1 S 11 of the proposed UWB antenna with different value t...31 Fig Measured and simulated VSWR of the proposed UWB antenna...32 Fig. 2.12(a) Measured and simulated S 11 of the proposed UWB antenna...32 Fig. 2.12(b) Phases of the return loss for the proposed UWB antenna Fig Measured radiation patterns of the proposed antenna on yz-plane at (a) 3.5GHz, (b) 5GHz, (c) 6.5GHz, and (d) 9.5GHz...34 Fig Measured radiation patterns of the proposed antenna on xz-plane at (a) 3.5GHz, (b) 5GHz, (c) 6.5GHz, and (d) 9.5GHz...35 Fig Measured radiation patterns of the proposed antenna on xy-plane at (a) 3.5GHz, (b) 5GHz, (c) 6.5GHz, and (d) 9.5GHz...36 Fig Measured total peak gain for the proposed UWB antenna...37 Figure 2.17 Arrangement for measuring the normalized antenna transfer functions vii

12 Figure 2.18 spectrum of the input pulse Fig Transmitted and received signal waveform in the time domain...38 Fig. 3.1 Geometry of the band-notched design using the open-end slits. (a) Top layer view (b) cross-sectional view (c) its photograph...45 Fig. 3.2 Geometry of the band-notched design using the parasitic strips. (a) Top layer view (b) cross-sectional view (c) the photograph Fig 3.3 Field distributions on the slot...48 Fig. 3.4 Effects of varying S 1 of the band-notched design using the open-end slits Fig. 3.5 Effects of varying r 1 of the band-notched design using the open-end slits Fig. 3.6 Effects of varying S 2 of the band-notched design using the open-end slits Fig. 3.7 Effects of varying S 1 of the band-notched design using the parasitic strips....5 Fig. 3.8 Effects of varying S2 of the band-notched design using the parasitic strips...5 Fig. 3.9 VSWR for the band-notched design using open-end slits...51 Fig. 3.1 Return loss for the band-notched design using open-end slits...51 Fig phase of return loss for the band-notched design using open-end slits...52 Fig VSWR for the band-notched design using parasitic strips...52 Fig Return loss for the band-notched design using parasitic strips Fig Phase of the return for the band-notched design using parasitic strips Fig Measured radiation patterns of the band-notched design using the open-end slits on yz-plane at (a) 3GHz, (b) 3.5GHz, (c) 4.5GHz, and (d) 9.5GHz Fig Measured radiation patterns of the band-notched design using the open-end slits on xz-plane at (a) 3GHz, (b) 3.5GHz, (c) 4.5GHz, and (d) 9.5GHz Fig Measured radiation patterns of the band-notched design using the open-end slits on yx-plane at (a) 3GHz, (b) 3.5GHz, (c) 4.5GHz, and (d) 9.5GHz Fig Measured radiation patterns of the band-notched design using the parasitic strips on yz-plane at (a) 3.5GHz, (b) 4.5GHz, (c) 6GHz, and (d) 9.5GHz viii

13 Fig Measured radiation patterns of the band-notched design using the parasitic strips on xz-plane at (a) 3.5GHz, (b) 4.5GHz, (c) 6GHz, and (d) 9.5GHz Fig. 3.2 Measured radiation patterns of the band-notched design using the parasitic strips on yx-plane at (a) 3.5GHz, (b) 4.5GHz, (c) 6GHz, and (d) 9.5GHz Fig Measured peak gain...6 Fig. 4.1 Geometry of an edge-fed folded CPW. (a) Top layer view (b) cross-sectional view (c) photograph Fig. 4.2 Measured characteristic impedance of the edge-fed waveguide...7 Fig. 4.3 Measured and simulated insertion loss of the edge-fed waveguide...7 Fig. 4.4 UWB antenna with a pair of symmetrically curved slot fed by CPW...71 Fig.4.5 simulation results for the originally unfolded UWB antenna compared with afterfolded antenna Fig. 4.6 Field distributions on the slot region at (a) 3GHz, (b) 4.5GHz, (c) 7GHz, (d) 9GHz.73 Fig. 4.8 Geometry of the novel folded UWB antenna. (a) Top layer view (b) cross-sectional view (c) photograph...76 Fig. 4.9 Measured and simulated VSWR of the folded UWB antenna...77 Fig. 4.1 Measured and simulated return loss of the folded UWB antenna...77 Fig Measured and simulated phase of the folded UWB antenna...78 Fig Measured radiation pattern of the novel folded-uwb antenna on E-plane ( xz-plane ) at (a) 3GHz, (b) 4.5GHz, (c) 7GHz, and (d) 9GHz...79 Fig Measured radiation pattern of the novel folded-uwb antenna on H-plane ( yz-plane ) at (a) 3GHz, (b) 4.5GHz, (c) 7GHz, and (d) 9GHz...8 Fig Measured radiation pattern of the novel folded-uwb antenna on xy-plane at (a) 3GHz, (b) 4.5GHz, (c) 7GHz, and (d) 9GHz Fig Measured peak gain of the novel folded UWB antenna...82 ix

14 Chapter 1 Introduction 1.1. Background and Motivation For the people in the 21 century, communication between each person is more urgent and frequent than those in the 2 century. Transmission speed in high data rate is the most compelling aspect from a user s point of view and also from a commercial manufacture s position. High data rates can enable new applications or business model that will be possible in the future. Now, data transferring speed of the mobile communication over 5 Mbps has been demonstrated, and the potential for much higher speed over wireless communication will be expected. As we know, the more transferring data rate we want, the more transmission bandwidth we need. So, ultra-wideband transmission has recently significant attention in both academia and industry for application in wireless communications. UWB system has many benefits, such as high data rate, availability of low-cost CMOS transceiver, low transmit power, multipath immunity, and so on [1]. The name Ultra Wideband is an extremely general term to describe a particular technology. Most people would see UWB as a new technology. Actually, the first UWB radio, by definition, is a pulse-based transmission system. It had its origins in the Spark-Gap transmission design of Hertz and Marconic in the late 189s [2]. In 1893, Heinrich Hertz used a spark discharge to produce electromagnetic waves. Then, Marconic developed wave generator based on a pulse-based Spark-Gap and arc discharge between carbon electrode in In other words, the first wireless communication system was based on UWB. However, due to technology limitation, strong emission and interference to narrowband radio system (continuous wave modulation), Spark-Gap radios were forbidden in most application. Until recently, the interest in UWB was sparked since FCC issued a Report and Order allowing 1

15 its commercial development[i]. In February 24, the FCC allocated the GHz spectrum for unlicensed use [3]. This enabled the use and marketing of products which incorporate UWB technology. Ultra wideband is defined as any communication technology that occupies a fractional bandwidth W/f c 2%, where w is the transmission bandwidth and f c is the band center, or more than 5 MHz of absolute bandwidth [4]. It is found that UWB system occupies ultra-wide bandwidth. To avoid interference with other existing narrowband wireless-communication systems, the power spectral density of a modulated UWB signal must satisfy a spectra mask subject to a regulating report of Part 15,2g rules [5]. The spectral mask for indoor applications specified by FCC in the United States is shown in Figure 1.1. Operation at such a wide bandwidth is limited in low power (-41.3dBm) that enables peaceful coexistence with narrowband system. These specifications give communication engineers opportunities and challenges to design available UWB system including RF circuit and antenna. For the antenna designers, all of the fundamental parameters, such as impedance matching, radiation patterns, gain, and radiation efficiency, must be considered in design antenna, including ultra wideband. However, there are some additional challenges for UWB. By definition, an UWB antenna must be operable over the entire GHz frequency range. Therefore, the UWB antenna must achieve a bandwidth spanned over 7.5 GHz. Another consideration that must be taken into account is group delay. Group delay is given by the derivative of the unrapped phase of transfer function of an antenna. If the phase is linear throughout the frequency, the group delay will be constant for the frequency range. This is an important characteristic because it helps to indicate how well a UWB pulse will be transmitted and to what degree it may distorted or dispersed. In addition, a nearly Omni-directional radiation pattern is also important for UWB system. Conductor and dielectric losses should be minimized in order to maximize radiation efficiency. High radiation efficiency is urgent for an ultra wideband because the transmit power spectral 2

16 density is excessively low for UWB system. To sum up, the UWB antenna requirements can be summarized in the following table [6]. VSWR Bandwidth Radiation Efficiency Phase Radiation Pattern Directivity and Gain Half Power Beamwidth Physical Profile GHz High(>7%) Nearly linear, constant group delay Omnidirectional Low Wide(>6%) Small, Compact, Planar Another issue is that UWB transmitter should not cause any electromagnetic interference (EMI) on nearby communication systems such as GPS and wireless LAN (WLAN) system. So, UWB antennas are also necessary for the rejection of an interference with existing wireless networking technologies such as IEEE 82.11a in the U.S. ( GHz, GHz) and HIPERLAN/2 in Europe ( GHz, GHz)[7]. Therefore, UWB antennas with a band-notched feature in WLAN-band (5-6GHz) are desired and important in the future Literature Survey Traditionally, there are many known antenna topologies that are said to achieve broadband characteristics, such as horn antenna, helix antenna, spiral antenna, log-periodic antenna, biconical antenna, monoconical antenna, discone antenna, sleeve antenna, bow-tie antenna and traveling-wave antenna. Their theories and techniques can be referred to Balanis s textbook [8]. Recently, some new planar wideband antenna emerged in the 3

17 literatures, such as planar wideband monopole, planar bow-tie, or planar slot antenna [9-15]. From the view point of theory and technique, those antennas mentioned above can be divided to some categories. These theories or techniques have been developed in past to design antenna with matched characteristics of wideband impedance. First, traveling wave antennas are inherently wideband and have been extensively used. Self complementary concept is second method that can provide the constant input impedance irrespective of frequency provided that the size of the ground plane for slot segment of the antenna is large and an appropriate self-complementary feed can be designed. However, its drawbacks are that it cannot be printed on a dielectric substrate as well as its input impedance is 18Ω, not 5Ω. The third kind technique for designing wideband antenna is to use the concept of multiple resonant modes. Log-periodic antenna, microstrip patch with parasitic elements, and planar wide-side antenna are examples of this category. Another technique to broaden the bandwidth of antenna for dipole geometry leads to a broader bandwidth. Biconical antenna, bow-tie antenna, or planar wideband monopole antennas are examples of this kinds, discone monoconical antenna are also its relatives [8]. The drawback of these types of antenna is that the antenna polarization as a function of frequency changes. Some antenna topologies mentioned above can achieve ultra-wideband but some mentioned can not support for UWB application. Recently, various planar antennas have been proposed for UWB system application. Planar wideband monopole and its relatives obtain many researchers interest. In the past decade, planar monopole has been investigated and proposed as an UWB antenna. In 1998, Agrawall proposed three kinds of ultra-wideband planar monopole antenna which yielded an impedance bandwidth of GHz for VSWR 2. After that, many researchers applied planar monopole to design UWB antenna with band-notched function [9]. Cho and Park proposed a miniature UWB planar monopole antenna with 5GHz band-rejection filter function and discussed its time-domain characteristic [7]. Lee also presented a wideband 4

18 planar monopole antenna with dual-band-notched characteristics [1]. Liang reported a printed annular monopole antenna by CPW feeding [11]. Luo proposed a dual band-notched UWB monopole antenna with an angular CPW-feeding structure [12]. Among the planar UWB antenna designs in the recent literature, the slot-type antenna is one of the most promising candidates for UWB applications [13-15]. The advantages of slot antennas include wide-bandwidth, low cost and planar in the PCB process. In 21, Prof. Wong demonstrated that a new design of printed wide-slot antenna fed by microstrip line with a fork-like tuning stub in the back plane can enhance impedance bandwidth [13]. Then, square slot planar antenna fed by coplanar waveguide with different form of stubs is reported in their research to broaden the bandwidth [14]. In 24, Chair et. al. presented the first UWB rectangular-slot antenna fed by coplanar waveguide [15]. From that time to now, many publishes based on planar slot fed by different transmission lines and stubs can be found in the open literatures [16-18]. In 25, Behad employed a fictitious short circuit concept to design a wide-band slot antenna. Its drawback is that it has large volume [19]. Lin et. al. proposed two designs based on printed elliptical and circular slot to design UWB antenna [17]. They applied the concept of multiple-resonators to explain why the slot aperture can achieve UWB characteristics. At the same year, Ma et. al. designed his slot antenna with tapered ring fed by coplanar waveguide [18]. Similarly, Sun proposed his UWB slot antenna by using a pair of symmetry curved radiating slot fed by coplanar waveguide [2]. About UWB antenna with stub-band rejection capabilities, in 26, Lin designed an UWB rectangular aperture antenna with band-notched characteristic using the isolated slit, the open-end slits, and the parasitic strips, irrespectively [17]. Adboh introduced a tuning slit on the feeding tuning stub to achieve band-rejection function in their elliptical slot antenna [21]. Recently, for shrinking the geometry size, Ding applied multiple-ring-resonators printed inside the inner patch to achieve multiple notch frequency bands [22]. The notched frequency band centered at 3.65GHz and 5.75GHz while retaining UWB from 3.1GHz-1.6GHz [23]. 5

19 1.3. Chapter Outlines In this thesis, chapter 2 will generalize the methodology of designing an elliptical slot antenna for UWB applications. The concept of multiple resonators will be introduced to broaden the impedance bandwidth for the elliptical slot antenna fed by a coplanar waveguide with different turning stub. The design parameters which affect antenna performance will be studied. Since the dimension of the slot perimeter is directly related to the lowest edge frequency of impedance bandwidth. Design criterion of an UWB antenna will be further investigated in this chapter. Chapter 3 will present an UWB elliptical slot antenna with band-notched characteristic. Two kinds of band-notched structure will be introduced and applied to reject the undesired band. One is a pair of open-end slits that are embedded into the circular stub. The other is a pair of parasitic strips that are inserted in the aperture region. They will be demonstrated to achieve the rejection function. Also, the practical prototypes will be manufactured to perform experiment to verify the simulation results. In this thesis, the primary application of our design was focused on being capable of integration with IC for portable electronic devices. Therefore, the antenna is required to be physically compact and low profile and preferably planar. Thus, Chapter 4 will be focused on shrinking the geometry of UWB antenna. Here, a compact planar antenna that originally consists of a pair of symmetrically curved radiating-slot fed by coplanar waveguide will be proposed and studied to achieve UWB requirement. If this pair of planar curve slot antenna is symmetrically folded by the center feeding line, the size can be shrunk to a half of original geometry size. The impedance bandwidth, radiation pattern, and radiation gains of this shrunken UWB antenna will be investigated and further measurement will be performance to verify its UWB characteristic. 6

20 -3-4 Normalized Spectrum (db) spectrum of the input pulse FCC indoor emission mask Frequency (GHz) Fig.1.1 FCC indoor emission spectrum mask. 7

21 Chapter 2 Design and Analysis of UWB Slot Antenna This chapter will be devoted to design an elliptical slot antenna fed by coplanar waveguide. By plotting the mode-patterns of electric fields on the slot region, multiple resonance concept will be described and included to explain why it can achieve ultra-wide bandwidth. Then, design methodology for the elliptical slot antenna will be introduced and parametric analysis will be studied to complete a practical design. An EM simulation tool, HFSS, will be invoked to simulate its response. Furthermore, practical structure will be fabricated and experiment will be performed to verify simulation results. Finally, time domain analysis will be devoted to the pulse-based wave transmission for UWB applications Slot Antenna for Broadband Applications Slot antenna fed by CPW/microstrip lines A conventional coplanar waveguide (CPW) fabricated on a dielectric substrate was first demonstrated by C. P. Wen in 1969 for microwave signal transmission. As shown in Figure 2.1, a coplanar waveguide on a dielectric substrate is composed of printed slot-lines with a third strip conductor centered in the slot region, and the dielectric structure can support even or odd quasi-tem mode of propagation. Traditionally, the electric field of micro-strip transmission line comes into existence among the dielectric structures but the electric field of coplanar waveguide transmission line existed mostly on the gap of coplanar waveguide transmission line. The existence of the dielectric structure affects the micro-strip transmission line more than coplanar waveguide transmission line [14]. Hence, coplanar waveguide transmission line has lower dielectric loss and less dispersion than micro-strip line. The CPW 8

22 offers several advantages over conventional micro-strip line [14], such as easy fabrication, easy shunt, as well as easy series surface mounting of active and passive devices. Also, it eliminates the need for wraparound and via holes, and fourth, it reduces radiation loss. The slot antenna fed by CPW has recently received much attention and is applicable to wireless communications systems. The advantages of CPW-fed wide slot antennas include wide bandwidth performance, low cost in the printed circuit board (PCB) process, and easy integration with monolithic microwave integrated circuits (MMICs). A few methods have been used to increase the bandwidth of CPW-fed slot antennas, including the use of a wide rectangular slot [18] or a bow-tie slot. Other broadband designs such as using a patch element loaded in a circular slot or the hybrid slot have also been used to obtain a dual-resonance response. To achieve ultra wideband, the bandwidth enhancement is the main purpose of these slot antenna designs. The approaches can be categorized into two kinds. One is to manipulate the field distribution in the slot with a tapered shape or with a feeding scheme to generate multiple resonances of close bands [17]. The other is to use a widened slot (or aperture, precisely speaking) and a fork-like stub for excitation such that a broad bandwidth can be achieved [15]. The latter approach has significant progress on the bandwidth enhancement and has reached the UWB bandwidth requirement recently. However, the design of using fork-like stub requires relatively large aperture and contains many parameters for the complex geometry. In addition, it is difficult to modify the designed antenna for the band-rejection function, a desirable feature in the UWB system Rectangular /elliptical/circular slot antenna The slot antennas fed by CPW feeding technique, only one side of the substrate requires to be processed. Both radiating elements and ground planes are on the same side of the 9

23 substrate, most of the electromagnetic wave travels in the slots on the surface of the substrate, and less energy is lost in the substrate. Modified slot antennas provides a possibility for a wider impedance matching bandwidth, such as the rectangular slot [15], the circular or the elliptical slot [16], particularly for wide apertures embedded with a tuning stub. Therefore, slot antennas are suitable constructions for applications in ultra-wideband communications. Two different aperture designs of UWB antennas from the published papers are presented here to showcase the state-of-the-art. Chair and Prof. Lee et. al. demonstrated a rectangular aperture antenna which is capable of achieving ultra wideband, the first time in 24 [15]. Shown in Fig. 2.1 is the configuration of their CPW-fed UWB antenna fabricated on a mm 3 Rogers RO43 substrate with dielectric constant of They proposed a stub which its arm length is close to a half wavelength. The ultra wideband feature is demonstrated from the return loss obtained by both simulation and measurement. The return loss is less than -1 db from 2.79 GHz to 9.48 GHz, which corresponds to an impedance bandwidth of 11%. From their results shown in his figure 2 in their published, it is found that the wide bandwidths are due to the multiple resonances introduced by the combination of the rectangular slot and the U-shaped stub. Radiation patterns displayed in his results showed good beamwidths and polarizations. Their slot antenna radiates bidirectionally, and the 3 db beamwidths at 9 GHz are 1 and 46 in the E-plane and H-plane [15]. Prof. Li et. al. presented another kind of slot antenna fed by CPW [16]. They proposed a slot antenna with elliptical shape or circular shape. Their elliptical slot antenna has a ground plane of 4 38 mm 2 and another kind of circular slot antenna has a size of mm 2. Their elliptical antenna shown in Fig.2.2 is printed on one side of a rectangular FR4 substrate with thickness of 1.5 mm and relative permittivity of 4.7. They constructed a central signal strip with a feeding CPW tapered with a slant angle θ=15 degrees for a length t to connect with the 1

24 U-shaped tuning stub which is symmetrically positioned with the elliptical/circular slot. Simulation and measurement results in their paper showed that the bandwidth of return loss which is less or equal to -1 db is from 3.1 GHz to 1.6 GHz for elliptical slot antenna and from 3.75 GHz to 1.3GHz for circular slot antenna. Table II is some summaries of comparison between the rectangular, the elliptical and the circular slot antenna. Rectangular slot antenna Elliptical slot antenna Circular slot antenna Bandwidth (S 11-1 db) GHz Radiation Patterns Bi-directionally Peak Gain Variation/Minimum peak gain (Simulation) Average gain 3.5dBi±1.6dBi Slot range length/width 32.2mm/21.1mm GHz Omni-directional 6dBi/2dBi 3.8mm/2mm GHz Not available 5.8dBi/2.1dBi 26.6mm/26.6mm Table II summaries of comparison between rectangular, elliptical and circular slot antennas Multiple resonators concept The concept of multiple-resonators can be employed to explain why a slot aperture can achieve UWB characteristics. There are many methods mentioned and developed in the past to design antennas with wide impedance bandwidth [21]. Multiple resonances concept is one kind of famous approach. Multiple-resonant radiating structures, such as Log-periodic antennas, microstrip patches with parasitic elements or slotted antenna with various slits are well-known third examples for increasing impedance bandwidth. Here, the combination of aperture slot antennas and those suitable tuning stubs are found that multiple resonances can be excited here for ultra-wideband application. Shown in Fig. 2.3 are electric field distributions on the slot region at different resonating 11

25 frequencies, 3.5GHz, 5GHz, 6.5GHz and 9.5 GHz, respectively. Those frequencies where resonances occurred are chosen to plot field distributions in order to explain the concept of multiple resonances. After studying and surveying over these figures, it is found that many resonance modes happened at different frequencies. Also shown in Fig. 2.3 with arrow-type are the directions of magnetic current distributions on the slot regions. The concept of the magnetic current can help us to explain why this antenna can be viewed as a multi-modes resonator. In view of the symmetry Maxwell s equations, it is convenient to consider E n as the equivalent magnetic current flowing on the slot region, where n is a unit-vector normal to the aperture. Due to the elliptical boundary condition, the magnetic currents will form resonant modes on the aperture region. Current resonating can be applied to explain why the aperture can excite multiple resonance modes. If the field distributions between 5 and 6 GHz are explored in detailed, it is found that the electric fields around the tuning stub are strong and gradually decay to the ground edge. On the other side, the magnetic current circulates around between the circular stub and ground conductor edge. If a parasitic conductor strip is embedded on the slot region, the resonant magnetic current will be interrupted to form band-rejection characteristic. They will be discussed and applied in the next chapter Design of an Elliptical Slot Antenna Fed by CPW The lower edge of the impedance bandwidth To design a planar monopole antenna, the approximate lower frequency corresponding to VSWR=2 can be determined by an approximate formula [24]. Figure 2.4 shows the cylinder monopole antenna. Here, L and r are the height and radius of the cylinder. Refer to the textbook by C. K. Balanis, the antenna length of the monopole is lightly smaller than 12

26 quarter-wavelength, as given by [8] L =.24λF (1) where F = ( L / r) /(1 + L / r). From these equations, the wavelength is obtained λ = ( L + r) /.24. (2) Thus, the resonant frequency is given by c f = = = λ L+ r L+ r (3) Here, f is in GHz, L and r are in millimeters. The area of an elliptical slot is greatly concerned with frequency for its lower edge of the impedance bandwidth. Shown in Fig.2.5 is a configuration of an elliptical aperture antenna. First, the area of the elliptical slot must be decided according the given lowest resonant frequency. A planar monopole antenna radiated from the metal plane but the slot antenna radiated from slot region. In fact, the electric current of the former resonated on the conductor but the equivalent magnetic current of the later resonated in the slot region. Because they look much similar, the lowest resonant frequency of the slot antenna can be derived from above-mentioned empirical formula [24]. As shown in Fig.2.5, the major and minor axis of the elliptical slot are A and B, respectively. The lowest resonating frequency corresponding to VSWR=2 can be approximately calculated by equating its area to that of an equivalent cylindrical monopole as below: 2π rl = π AB (4) The long axis radius A is chosen to be L=1.25A and the short axis radius B is r=.4b from (eq.4). According to the empirical formula (3), it can be revised to 13

27 where f l 3 C = 1.25A+.4 B f l is the frequency (in gigahertz) corresponding to the lower edge of the -1 db (5) impedance bandwidth, A and B are in millimeters. Element factor C for elliptical slot is equal to.22 while the scope of the axial ratio (A/B) is 1.45 to 1.5 and the relative dielectric constant of the substrate is 4.3. The configuration of an ultra-wide band slot antenna is shown in Figure 2.6 (a). This UWB antenna is fed by a coplanar waveguide transmission line with an elliptical tuning stub for impedance matching. It is fabricated on a rectangular FR4 substrate with substrate thickness h = 1.6 mm, relative dielectric constant ε = 4.3, loss tangent=.2, and the r ground plane size is mm 2. Its photograph is shown in Fig.2.6 (b). For achieving the allocated and desired impedance bandwidth, the size of the elliptical aperture must be carefully designed. In the beginning, w=1.45mm was calculated by TXline to obtain a 5ohm CPW transmission line if g=.25mm. To offer an UWB operating bandwidth from 3.1 to 1.6 GHz, the ratio of major axis to minor axis (A/B) of the ellipse is initially set as an approximate 1.47 according to our experience. Then, B/A is about.68. If f l =3.1GHz, the calculation procedure for A and B is listed as follows: 3 C fl = = = = ( GHz) L+ r 1.25 A+.4 B 1.25A A A A = = = ( mm) f B =.68A = 9.52 ( mm) l Here, the dimensions of design parameters are initially set to A=14mm, B=9.52mm, a=5mm, b=5.65mm, t=.6mm, w=1.4mm, and g=.25mm. Why is the axis ratio set as 1.47? Simulation results can help to explain it. Shown in Fig. 2.7 are simulation results of the return loss with long axis length A as a parameter. For different value of A, A=12.8 mm, A=13.2 mm, A=13.6 mm, A=14 mm, A=14.4 mm, the corresponding lower frequency is 4.12 GHz, 4.6 GHz, 3.35 GHz, 3.11 GHz and 3.2 GHz, 14

28 respectively. The final length which corresponds to the demanded lower edge of the impedance bandwidth is 14 mm, i.e., the elliptical slot has a circumference of mm Impedance matching by a tuning stub Impedance matching is essential to antenna design. To prevent the reflections power back to the source, the antenna load must be matched exactly to the source impedance over a desired bandwidth. Impedance matching is to make the output impedance Z s of a source equal to complex conjugate of the input impedance Z L of the load in order to maximize the power transfer and minimize reflections from the load. It is very important for high speed or microwave circuit. However, impedance matching (here, also called broadband matching) to minimize reflections and maximize power transfer over a (relatively) large bandwidth is not easy to realize. Since the UWB system applies low-power CMOS devices, its output-power spectral-mask is defined to allow a spectral density of dbm/mhz, which is lower than any other communication systems. To obtain sufficient power the antenna matching is required to satisfy VSWR< 2 or return loss < -1dBm, which is hard to achieve for ultra wide-band antenna. Shown in Fig.2.8 is its impedance shown in Smith chart. It is found that the CPW transmission line is used to capacitively excite the input signal into the aperture at the lower band below 3GHz. A length t is applied to connect with the tuning stub which is symmetrically placed in the elliptical slot. This elliptical tuning stub and the elliptical slot exhibit the multiple resonance modes to achieve ultra-wide bandwidth Design parameter analysis For the slot structure with a tuning stub, the dimensions and feeding depth of the tuning stub have a great effect upon the impedance matching. To make use of an excitement between 15

29 the stub and slot segment can achieve impedance matching for the ultra wideband. Multiple resonances can be excited by the tuning stub embedded inside the elliptical slot to enhance the wideband. The tuning stub must be designed in detail! After the area and circumference size of the elliptical slot is decided in previous section, the tuning stub shown in Fig.2.6 can be adjusted to achieve good impedance matching. The axis lengths, say a and b, will affect the area and then capacitance of this tuning stub and finally have influences on the input impedance. Impedance matching for broad bandwidth is very sensitive to the change of the length t on this stub. Also, the radius of the elliptical stub not only affects the impedance performance but also have influence on the lower edge of the impedance bandwidth. For optimization matching, parametric study must be performed. Shown in the Fig.2.9 are return loss curves with different axis a as a parameter. As shown in this figure, when a =5 mm, the -1dB of return loss is consistent with the bandwidth form 3.1 to 1.6 GHz. The tuning stub is extended into the aperture for impedance matching. The distance between this tuning stub and CPW transmission line is t, which makes the input impedance sensitive. Therefore, the lower resonant frequency will be affected. For the different value of t, t=1. mm, t=.8 mm, t=.6 mm, t=.4 mm, t=.2 mm, the corresponding lower frequency affected by the feeding depth t is shown in Fig Choosing t as.6mm meets our requirements, i.e., the lowest operating frequency is 3.1 GHz. Finally, the dimensions of the practical UWB slot antenna are A=14mm, B=9.52mm, a=5mm, b=5.65mm, t=.6mm, w=1.4mm, and g=.25mm Simulation and Measurement Results The antenna shown in Fig. 2.6(b) was manufactured on the FR4 substrate by a PCB mark plotter. The measurement was performed by HP8363-PNA network analyzer. SMA connector 16

30 was also included in the simulation by a 3D EM-simulator, FHSS, which is based on the finite element method Voltage standing wave ratio (VSWR) and return loss Perhaps the most obvious metric for antenna performance is the antenna s resonant frequency and impedance bandwidth. Input matching can be described either by VSWR or return loss S 11. Generally speaking, the bandwidth is frequently defined as the frequency range over which VSWR<2 or S 11 <-1dB. Since FR4-substrate is lossy at those frequencies, roughly say higher than 5GHz, one need to be careful to understand how this influences on antenna performance. As a matter of effect, VSWR is not easy to exhibit the information of substrate loss or radiation loss. By comparison, return loss is more effective to realize the information about loss but VSWR is generally applied in UWB design. Shown in Fig.2.11 is measured VSWR for the UWB antenna shown in Fig.2.6. Low VSWR over the required frequency band is found to be between 3.1GHz to 1.6GHz. Fig.2.12 shows return loss for this antenna, here, (a) shows magnitude response and (b) phase response. Solid (red) line shown in this figure is measured return loss and dashed (blue) line is simulation results by HFSS. The resonant frequencies of the simulation and measurement results are in good agreement over the matching frequency band from 3GHz to 1GHz. The resonant frequencies are about located on the 3.5GHz, 5GHz, 6.5GHz and 9.5GHz. The wide bandwidths over 3GHz to 1GHz are due to the multiple resonances generated by the combination of the elliptical slot and the tuning stub. The lowest frequency of the simulation results has some discrepancy compared with the measurement. It is supposed that mesh segmentation numbers in the simulation tool between fed-cpw and tuning stub are not sufficient. It causes some numerical error since fields are sensitive on that region. 17

31 Radiation pattern and gain All of the UWB antennas in this thesis are characterized in the NTUT s anechoic chamber in order to determine radiation patterns and maximum gain values. Anechoic chamber patterns were measured with co-polarization and cross-polarization at the different -resonating frequencies. There are taken at 3.5GHz, 5GHz, 6.5GHz, and 9.5GHz, respectively. Shown in Fig.2.13 are measured patterns on the yz-plane (H-plane). The co-polarization field patterns are generated due to the x-direction field on the slot. It shows nearly omni-directional at theses frequencies. The cross-polarization field pattern at 3.5GHz is generated due to y-direction field on the slot. It leads to a bidirectional pattern in the H-plane (yz-plane). At 6.5GHz, The cross-polarization pattern can be modeled as the radiation from two small y-polarized dipoles out of phase, leading to the nulls in the y-axis (due to the element factor) and x-axis (due to the array factor) with relative maximums in between [17]. Shown in Fig.2.14 are measured patterns on the xz-plane (E-plane) at 3.5GHz, 5GHz, 6.5GHz, and 9.5GHz, respectively. The co-polarization fields on the xz-plane are generated due to x-direction field on the slot. It is considered as a fundamental resonating mode of the aperture antenna. This mode is like a dipole lying on the PCB plane if reviewing the field distribution in Fig.2.3. It leads to a nearly directional pattern in the E-plane. Since the fields is out of phase on the slot, the cross-polarization fields are cancelled each other on the far field region as shown in Fig Note that the co-polarization field patterns are no longer bidirectional. The field distribution contains multiple higher order modes and makes the peak of E-plane patterns shift slightly from the z-axis, as shown in Fig.2.14 (d). Generally speaking, UWB planar antennas are placed along the horizontal plane. So, the horizontal-cut pattern is also important in real application. Shown in Fig.2.15 are measured radiations on the xy-plane at different azimuth angles. The co-polarization fields have nulls, especially at higher resonating frequencies since multiple resonating modes are excited. Thus, 18

32 the antenna becomes more directive and exhibits a different mode of radiation. Shown in Fig.2.16 are measured maximum antenna gains for this UWB antenna. The antenna gain is measured in the anechoic chamber using a standard double ridge horn antenna (EMCO 3115). It is seen that the gain of the UWB antenna increases approximately linear, but some variation, over the entire bandwidth from the band 3GHz until 1GHz and reaches the maximum at 1GHz. The maximum gain occurs at the front and back of the antenna, with 2.11 db of gain at the front and db at the back Transfer Function and Time Domain Analysis Normalize transfer function In narrowband wireless communication system, antenna and front-end filter are generally designed in the frequency domain since the fractional bandwidth of transmission is small. However, it is not true for UWB system. Basically, UWB system is typically pulse-based transmission system. It does not need front-end filter but antenna acts like a filter for the generated UWB signal. The antenna only allows those signal component radiated to be passed. The lower and much higher frequency components of short pulse are reflected back. So, antennas are a critical element in the signal flow of UWB systems. Eventually, the basic effect of antennas is that they produce the derivative of the transmitted or received pulse waveform in the received antenna. In the time domain, it can be often approximated as a temporal differentiators/integrators or spectral/spatial filters both in the transmitter and receiver. From the system view of the wireless communication, if the magnitude of transfer function of a transmitting/receiving antenna system show in Fig.2.17 is not flat, or phase response is not linear, some problems will happen. When a very short time-domain pulse or 19

33 impulse (implying large bandwidth) is used to excite the antenna system, ringing effect will be the first problem. This ringing effect will spread the pulse and distort the pulse. Finally, time domain resolution of pulse trains will be decreased. That is to say, any distortion of the received signal in the frequency domain (which is a filter function response) will cause distortion of the transmitted pulse shape, therefore increasing the complexity of the detection mechanism at the receiver. So, whether the antenna transfer function is flat or phase response is linear is an important issue of concern from which we can judge to what extent the spectra of the pulse will be modified by the antenna. For UWB application, the magnitude of the transfer function of antenna system should be flat as possible in the operating band. The phase response must be linear and then group delay is required to be constant over the entire band as well. Various literatures have been devoted to evaluating the antenna transfer function [25]-[27]. A common way to describe system is by means of an impulse response in the time domain for ultra-wideband problem. The impulse response in the time domain is equivalent to the transfer function in the frequency domain. If the signal which inputs into antenna system is a short-pulse (ultra-wide bandwidth), the input impedance Z a at the antenna terminal is really a frequency function. So, it would be logical and easier to integrate the frequency-dependent term Z a in the impulse response. When normalizing the voltages and electric field to the characteristic impedance Z c, the normalized impulse response (IR) of N, TX h and h N, RX can be defined as [25] h h NTX, NRX, = = Z Z c a Z Z a c τ τ TX f g RX f g h h TX RX (1) Here, τ 2Za Za 2Zc =, f =, τ = Z + Z Z Z +Z TX g RX c a c c a 2

34 If the transmitting and receiving antenna are the same, the impulse responses of transmitting and receiving antennas will be same, h N, TX = h N, RX. In reference [25], the received voltage V ( t) after the received antenna, measured with a Z = 5Ω oscilloscope rec c at the receiving antenna can be related to the input applied voltage Vs ( t) before the transmitting antenna by 1 d R Vrec() t = hn, TX (,, ) Vs () t hn, RX (,, ) ( ) 2πRC τθφ dτ τθφ δτ C (6) where R is the distance between the virtual sources of the antennas. Converting (6) to the Vrec ( f) frequency domain by Fourier transform and using the definition S 21 ( f) = V ( f), we have [25] R jλ j2π f C S21 ( f) = HNTX, ( f, θφ, ) HNRX, ( f, θφ, ) e (7) 4π R s Here, hntx, ( f, θ, φ ) and hn, RX( f, θ, φ) are defined as the dimensionless normalized antenna transfer function of the transmitting and receiving antenna and normalized by factor 4 π / λ, λ is the waveleng th. The system transfer, which is in essence the transmission scattering parameter S21( f ) of a two-port network, can be measured in an anechoic chamber with a pair of proposed DUT antenna serving as the transmitting and receiving antenna. Also, it can be obtained from eq.(7) if the normalized antenna transfer function of the AUT is known. Detail about how to evaluate the normalized transfer function theoretically can be referred to [25] Guassian pulse for mask regulation As mentioned in previous chapter and Fig.1.1, FCC in the U.S. has specified spread mask for indoor and outdoor application. This emission-limit masks show in Fig.1.1 (blue line) regulate the spectra of 1-18GHz and effective isotropic radiated power (EIRR) levels in this 21

35 spectra. The source pulse radiated from transmitter must comply with this regulation. On the other hand, as considering the modulation approach in practice case, UWB system can utilize the UWB band in a variety of ways. For instance, multi-band schemes and single-band schemes have been proposed for UWB radio system. For the consideration of the emission control, source pulses must be decided to meet the requirement mentioned above. It must avoid possible interference with each channel and other system. Due to unique temporal and spectra properties, a family of Rayleigh (differentiated) Guassian pulse V ( ) n t or V% n ( ω) is widely used as the source pulse in UWB system [26], i.e., 2 2 t ωσ σ n 2 n ω ( ω) % σπ dn Vn() t = e, V ( ) = j e n dt (8) where the pulse parameter σ stands for the time when the Guassian pulse voltage decay to Vs ( σ ) = e 1. It will influence on the bandwidth of UWB signals. This single source, which usually has a very short duration, is shaped and varied so that its spectrum occupies as wide as possible bandwidth within the UWB regulation. This source pulse will be suitable for single-band scheme application. On the other hand, modulated Guassian monocycle of second order is suitable for the multiband scheme [28]. Here, since the available UWB is divided into several sub-band with bandwidth 5MHz, each of modulated source pulse is shaped to occupy only one sub-band to avoid the interference between each other. The modulated Guassian monocycle can be described as here f c is the carrier frequency. 2 d 2 VS ( t) = cos 2πf ct g ( t) 2 o (9) dt 2 2π t σ g ( t) = Ae (1) o Time Domain analysis 22

36 To illustrate how the pulses are distorted by a transmitting-receiving antenna system, time domain analysis must be done and time domain characteristic of antenna system must be investigated. From above definition, the source pulse spectrum Vs ( f ) can be obtained by Fourier transform if the input source Vs ( t) is assigned. This pulse spectrum is then multiplied by antenna system transfer function in (7) and then inverse Fourier transform is performed to obtain the required time domain response V rec ( t). The output waveform at the receiving antenna terminal can be expressed as V ( t) = F V ( f) S ( f,, ) Π( f) 1 { } 21, θφ rec s system (11) where Π ( f ) represent an ideal bandpass filter from 1 to 18 GHz. After obtaining the output waveform, we can discuss the similarity between the transmitting pulse and the receiving pulse. A well-defined parameter named fidelity is then proposed to describe the capability of pulse detection of an antenna [ 25] V () t V ( t+ τθφ,, ) dt Fidelity( θφ, ) = MAX τ V () t dt V (, t θφ, ) dt s rec 2 2 s rec. (12) Simulation results and measurement In this thesis, the source pulse at the transmitting antenna is assumed to be a fifth derivative of a Guassian function for single-band scheme application 2 t π π π 2 4 σ V S ( t) = A 3 6( ) t + ( ) t e, (12) σ σ here, A=.1 and σ =.175 μs. The normalized spectrum V ( f ) is shown in Fig.2.18 and it s proves to comply within the required FCC indoor emission mask limits. Simulation results for the transmitted wave and received waveform are shown in Fig It shows that received waveform is similar with the transmitted waveform. 23

37 2.5. Summary In this chapter, we have described a planar elliptical slot antenna with a circular tuning stub fed by coplanar waveguide. Design methodology and parameter analysis have been established for antenna designer to design a UWB antenna. Multiple resonance concepts have been included to illustrate why it has an ultra-wide impedance bandwidth. Measurement has also performed to verify the simulation results. The experimental results show a good agreement with the simulation results. The measured data indicates this planar antenna can achieve the system requirements for UWB applications. 24

38 Fig. 2.1 Geometry of a CPW-Fed rectangular slot antenna using a folk-shape stub. Fig. 2.2 Geometry of a CPW-fed printed elliptical slot antennas using a U-shape stub. 25

39 (a) 3.5GHz (b) 5GHz 26

40 (c) 6.5GHz (d) 9.5GHz Fig. 2.3 Field distributions on the slot region. 27

41 Fig. 2.4 The equivalent solid cylinder. Fig. 2.5 Configuration of an elliptical aperture antenna. 28

42 2A tuning stub a b 2B t g w g x y coplanar waveguide 5Ω SMA gronnd plane g w g gronnd plane y z coplanar waveguide substrate h Fig. 2.6(a) Geometry of the proposed UWB antenna. Fig. 2.6(b) its photograph. 29

43 Return Loss (db) A = 12.8 mm A = 13.2 mm A = 13.6 mm A = 14. mm A = 14.4 mm Frequency(GHz) 11 Fig. 2.7 S 11 of the proposed UWB antenna with different value A. Smith Plot 1 Name Freq Ang Mag RX m i m Fig. 2.8 Smith chart for the input impedance. 3

44 -5 Return Loss (db) a = 5.4 mm a = 5.2 mm a = 5. mm a = 4.8 mm a = 4.6 mm Frequency(GHz) 11 Fig. 2.9 S 11 of the proposed UWB antenna with different value a Return Loss (db) t=1. mm t=.8 mm t=.6 mm t=.4 mm t=.2 mm Frequency(GHz) 11 Fig. 2.1 S 11 of the proposed UWB antenna with different value t. 31

45 1 9 8 Measured VSWR Simulated VSWR 7 VSWR Frequency ( GHz ) Fig Measured and simulated VSWR of the proposed UWB antenna Return Loss (db) Measured S 11 Simulated S Frequency(GHz) 11 Fig. 2.12(a) Measured and simulated S 11 of the proposed UWB antenna. 32

46 Phase (degree) Measured phase Simulated phase Frequency(GHz) 11 Fig. 2.12(b) Phases of the return loss for the proposed UWB antenna. 33

47 YZ-plane 3.5GHz co-pol 3.5GHz cross-pol YZ-plane 5GHz co-pol 5GHz cross-pol (a) (b) YZ-plane 6.5GHz co-pol 6.5GHz cross-pol YZ-plane 9.5GHz co-pol 9.5GHz cross-pol (c) (d) Fig Measured radiation patterns of the proposed antenna on yz-plane at (a) 3.5GHz, (b) 5GHz, (c) 6.5GHz, and (d) 9.5GHz. 34

48 XZ-plane 3.5GHz co-pol 3.5GHz cross-pol XZ-plane 5GHz co-pol 5GHz cross-pol (a) (b) XZ-plane 6.5GHz co-pol 6.5GHz cross-pol XZ-plane 9.5GHz co-pol 9.5GHz cross-pol (c) (d) Fig Measured radiation patterns of the proposed antenna on xz-plane at (a) 3.5GHz, (b) 5GHz, (c) 6.5GHz, and (d) 9.5GHz. 35

49 YX-plane 3.5GHz co-pol 3.5GHz cross-pol YX-plane 5GHz co-pol 5GHz cross-pol (a) (b) YX-plane 6.5GHz co-pol 6.5GHz cross-pol YX-plane 9.5GHz co-pol 9.5GHz cross-pol (c) (d) Fig Measured radiation patterns of the proposed antenna on xy-plane at (a) 3.5GHz, (b) 5GHz, (c) 6.5GHz, and (d) 9.5GHz. 36

50 8 7 6 Gain ( dbi ) Frequency ( GHz ) 1 Fig Measured total peak gain for the proposed UWB antenna. Figure 2.17 Arrangement for measuring the normalized antenna transfer functions. 37

51 -3-4 Normalized Spectrum (db) spectrum of the input pulse FCC indoor emission mask Frequency (GHz) Figure 2.18 spectrum of the input pulse. Signal strength ( V/m ) Transmitted signal Received signal Time (ns) Fig Transmitted and received signal waveform in the time domain. 38

52 Chapter 3 Band-Notched UWB Slot Antenna Fed by CPW To overcome electromagnetic interference on nearby existing WLAN in 5-6GHz bands, UWB antenna with a band-notched function is desired and must be developed. This chapter will present an UWB elliptical slot antennas with band-notched characteristics. Here, two kinds of band-notched structures will be introduced and applied to reject undesired sub-band. One is a pair of open-end slits that are embedded into the circular stub. The other is a pair of parasitic strips that are inserted in the aperture region. They will be investigated in order to achieve band rejection function. Finally, their practical prototypes will be fabricated to perform the experiment to verify simulation results Geometry and Band-notch design UWB slot antenna band-notched using open-end slits Shown in Fig.3.1 are (a) top layer view of a band-rejection UWB slot antenna and (b) its cross-section view. Practical structure is manufactured and its photograph is shown in Fig.3.1(c) with a SMA connector. The UWB antenna consists of an elliptical slot which is the same as that in the previous chapter. This UWB antenna is fed by a CPW transmission line with a tuning stub for impedance matching. Two quarter-wavelength slits at 5.5GHz were formed by etching on this tuning stub. It is also fabricated on the same plane, where has only one layer of substrate of a thickness 1.6 mm and relative permittivity ε r =4.3. Why these two slits can form notch behavior? Shown in Fig.3.3 (a) are field distributions before the stub was cut to form slits but Fig.3.3 (b) show field distributions after cutting to form slits. The field near the region A is strong means that electric current density on the stub 39

53 near the region A is much high. Cutting out some conductor to form slits on the stub can deteriorate the current near the region A. One end of the quarter-wavelength slit is open but the other end is shorted and connected to fed-cpw transmission line. It is found that inserting these slits causes anti-resonate at 5-6GHz band so that the resonating field was deteriorated and impedance matching was damaged. This is why it has band-rejection feature. Position, gap widths and length of the slits must be suitably selected to control desired rejection band. Design parameters S 1, r 1, and S 2 shown in Fig 3.1(a) must be optimally designed. In order to design a notch-band, design parameters must be analyzed by the simulation tool, HFSS. Here, the distance S 1, S 2 and radius r 1 are the most importance parameters which affect the antenna performance so that they need to be further investigated. The simulated VSWR curves of the slot antenna for various S 1 are illustrated in Fig.3.4. It is seen that the curves with different S 1 have similar behavior on notch function. The optimal distance is S 1 =1. mm because central resonating frequency can be located at 5.5GHz. Fig.3.5 shows the return loss curves for different r 1 and Fig.3.6 plotted for different S 2. The optimal radius r 1 =4.6mm and the optimal S 2 =.9mm can achieve what we want. To sum up, if the slit length of one side is adjusted to be about a quarter-wavelength at the center frequency of the desired band, a narrow bandwidth can be notched out while it still keeps good impedance matching for the rest of ultra-wide bandwidth. Finally, total antenna size is mm 3. The geometry size of the designed parameters of the antenna band-notched by open-end slits are A=14mm, B=9.52mm, a=3.9mm, b=4.485mm, r 1 =4.6mm, r 2 =5.4mm, S 1 =1.mm, S 2 =.9mm, t=.6mm, w=1.4mm and g=.25mm. The antenna is fabricated on a 1.6 mm FR4 epoxy substrate with relative permittivity ε r =4.3 and loss tangent tanδ=.2. A ground plane which size is 33 mm 36 mm was circled around the etched aperture. The ellipse-shape radiator is fed by a CPW transmission line terminated with a SMA connector. Since the antenna and the feeding structure are implemented on the same plane, manufacturing for the antenna was very easy 4

54 and extremely low cost UWB slot antenna band-notched using parasitic strips Fig. 3.2(a) shows the geometry and configuration of another UWB band-notched antenna using parasitic strips. Fig. 3.2(b) shows the photograph of this practically manufactured structure. Inside the elliptical aperture shown in Fig.3.3(c), a pair of parasitic strips is inserted into suitable aperture-region to disturb the field distribution where resonates at the designed notch-band. Shown in Fig.3.6 are fields distributions after a pair of strips are inserted on the aperture. Compared with Fig.3.3(a) where field is strong in region B, it is found that the strong field was changed to be at the edge region near the ground plane. While the total length of this pairs of parasitic strip is adjusted to be about a half-wavelength at the center frequency of the desired notch-band, resonance will occur at the strip but not at the aperture. Then, a notched frequency band at 5.5GHz can thus be created to reduce the interference with the signal from WLAN when UWB system is in power-on mode. However, second-harmonic resonance also slightly occurred at 11GHz, as shown in Fig.3.7. The widths and length of the parasitic strips must be suitably adjusted to control the desired rejection band. Design parameters S 1, and S 2 shown in Fig 3.2(a) must be optimally designed by the simulation tool. The simulated return loss curves of the slot antenna for various S 1 are illustrated in Fig.3.7. It is seen that the curves with different S 1 have similar behavior on notch function. The optimal distance is S 1 =11.4 mm because central resonating frequency can be located at 5.5GHz. In like manner, the optimal radius S 2 is about 8.9mm and then it can achieve what we want, as shown in Fig Finally, for the design structure, the dimensions of the antenna band-notched using the parasitic strips are A=14mm, B=9.52mm, a=4.8mm, b=5.28mm, S 1 =11.4mm, S 2 =8.9mm, t=1.17mm, w=1.4mm and g=.25mm, as shown in Fig

55 3.2. Simulation and Experimental Results Return Loss and voltage standing wave ratio Shown in Fig.3.9 is measured and simulated VSWR for the UWB antenna with open slits. It is found that the proposed band-notched antenna exhibits a notch frequency band centered at 5.4 GHz. Meanwhile, except the notch frequency band, the antenna worked over the required frequency band between 3.1GHz to 1.6GHz with VSWR<2, covering almost the entire frequency band for a UWB system. Fig.3.1 shows the magnitude response of the return loss for this antenna, Fig 3.11 show its phase response. It still has linear-phase response which is important for UWB application. Compared to the original design, this band-notch UWB antenna successfully blocks out the 5-6GHz band and still has good impedance matching at other frequency bands. Shown in Fig.3.12 is measured and simulated VSWR for the UWB antenna with parasitic strips. It is also found that the proposed band notched antenna exhibits a notch frequency band centered at 5.5GHz. Investigating the input impedance, it is found that minimum resistance (singular impedance) is made to form rejection. This is the design concept. However, the pair of these thin parasitic strips generated resonance so that it induces some radiation. Therefore, the VSWR drops a little. It means that it can not form complete band- rejection function. However, the resonant frequencies of the simulation and measurement results are in good agreement. Shown in Fig 3.13 are the measured and simulation return loss, Fig 3.14 shows it phase response Radiation pattern 42

56 For UWB applications, designed antenna is still required to have omni-directional radiation pattern. The requirement is confirmed by measured pattern. Shown in Fig.3.15 are measured patterns on the yz-plane (H-plane) for UWB antenna with open-end slits. The patterns in the H-plane do not change much outside the band-rejection band. The-cross polarization still leads to a bidirectional pattern in the H-plane at 9.5GHz. Shown in Fig.3.16 are measured patterns on the xz-plane (E-plane) at 3.5GHz, 5GHz, 6.5GHz, and 9.5GHz, respectively. The co-polarization still has directional pattern in the E-plane due to dipole-like behavior. Shown in Fig.3.17 are measured radiations on the xy-plane at different azimuth angles. The co-polarization fields have nulls, especially at higher resonating frequencies since multiple resonating modes are excited. Thus, the antenna becomes more directive and exhibits a different mode of radiation. Shown in Fig are measured patterns for UWB antenna using parasitic strips to achieve notch function. It is found that their behaviors are similar to those patterns using open-slits Gain The measured gain of the designed antenna is presented in Fig This figure reveals that the gain (in db) for the band 3GHz until 1GHz is between 3to 8 dbi for this antenna outside the rejected band. The gain in the rejected band is found to be as low as -3dB. Half power input to transmitter UWB antenna are rejected to avoid interference with existing WLAN system. 43

57 3.3. Summary An UWB elliptical slot antenna embedded with open-end slit on the tuning stub or parasitic strips on the aperture for achieving band-notch characteristics has been proposed in this chapter. Field distributions near the tuning stub were included to explain why these structures can achieve band-notch characteristics in the sub-band 5-6GHz. Experimental results have also confirmed band-rejection capability for the proposed antenna at the desired band. The measured radiation pattern shows that these antennas are still preserved nearly omni-direction radiation features in the E-plane. 44

58 Fig. 3.1 Geometry of the band-notched design using the open-end slits. (a) Top layer view (b) cross-sectional view (c) its photograph. (c) 45

59 2A S 1 S 2 tuning stub a b 2B t x g w g y coplanar waveguide 5Ω SMA gronnd plane (a) g w g gronnd plane y z substrate coplanar waveguide (b) h (c) Fig. 3.2 Geometry of the band-notched design using the parasitic strips. (a) Top layer view (b) cross-sectional view (c) the photograph. 46

60 (a) Field distributions before the slits are cut out. (b) Field distributions after the slits are cut out. 47

61 (c) Field distributions after a pair of parasitic strips is inserted. Fig 3.3 Field distributions on the slot S 1 =.6 mm S 1 = 1. mm S 1 = 1.4 mm S 1 = 1.8 mm VSWR Frequency(GHz) 11 Fig. 3.4 Effects of varying S 1 of the band-notched design using the open-end slits. 48

62 r 1 = 4.4 mm r 1 = 4.6 mm r 1 = 4.8 mm r 1 = 5. mm VSWR Frequency(GHz) 11 Fig. 3.5 Effects of varying r 1 of the band-notched design using the open-end slits S 2 =.5 mm S 2 =.7 mm S 2 =.9 mm S 2 = 1.1 mm VSWR Frequency(GHz) 11 Fig. 3.6 Effects of varying S 2 of the band-notched design using the open-end slits. 49

63 S 1 = 1.6 mm S 1 = 11. mm S 1 = 11.4 mm S 1 = 11.8 mm VSWR Frequency(GHz) 11 Fig. 3.7 Effects of varying S 1 of the band-notched design using the parasitic strips S 2 = 8.1 mm S 2 = 8.5 mm S 2 = 8.9 mm S 2 = 9.3 mm VSWR Frequency(GHz) 11 Fig. 3.8 Effects of varying S 2 of the band-notched design using the parasitic strips. 5

64 1 9 8 Measured VSWR Simulated VSWR 7 VSWR Frequency ( GHz ) 11 Fig. 3.9 VSWR for the band-notched design using open-end slits. Return Loss (db) Measured S 11 Simulated S Frequency(GHz) 11 Fig. 3.1 Return loss for the band-notched design using open-end slits. 51

65 Phase (degree) Measured phase Simulated phase Frequency(GHz) 11 Fig phase of return loss for the band-notched design using open-end slits Measured VSWR Simulated VSWR 7 VSWR Frequency ( GHz ) 11 Fig VSWR for the band-notched design using parasitic strips. 52

66 -5 Return Loss (db) Measured S 11 Simulated S Frequency(GHz) 11 Fig Return loss for the band-notched design using parasitic strips. Phase (degree) Measured phase Simulated phase Frequency(GHz) 11 Fig Phase of the return for the band-notched design using parasitic strips. 53

67 YZ-plane 3GHz co-pol 3GHz cross-pol YZ-plane 3.5GHz co-pol 3.5GHz cross-pol (a) (b) YZ-plane 4.5GHz co-pol 4.5GHz cross-pol YZ-plane 9.5GHz co-pol 9.5GHz cross-pol (c) (d) Fig Measured radiation patterns of the band-notched design using the open-end slits on yz-plane at (a) 3GHz, (b) 3.5GHz, (c) 4.5GHz, and (d) 9.5GHz. 54

68 XZ-plane 3GHz co-pol 3GHz cross-pol XZ-plane 3.5GHz co-pol 3.5GHz cross-pol (a) (b) XZ-plane 4.5GHz co-pol 4.5GHz cross-pol XZ-plane 9.5GHz co-pol 9.5GHz cross-pol (c) (d) Fig Measured radiation patterns of the band-notched design using the open-end slits on xz-plane at (a) 3GHz, (b) 3.5GHz, (c) 4.5GHz, and (d) 9.5GHz. 55

69 YX-plane 3GHz co-pol 3GHz cross-pol YX-plane 3.5GHz co-pol 3.5GHz cross-pol (a) (b) YX-plane 4.5GHz co-pol 4.5GHz cross-pol YX-plane 9.5GHz co-pol 9.5GHz cross-pol (c) (d) Fig Measured radiation patterns of the band-notched design using the open-end slits on yx-plane at (a) 3GHz, (b) 3.5GHz, (c) 4.5GHz, and (d) 9.5GHz. 56

70 YZ-plane 3.5GHz co-pol 3.5GHz cross-pol YZ-plane 4.5GHz co-pol 4.5GHz cross-pol (a) (b) YZ-plane 6GHz co-pol 6GHz cross-pol YZ-plane 9.5GHz co-pol 9.5GHz cross-pol (c) (d) Fig Measured radiation patterns of the band-notched design using the parasitic strips on yz-plane at (a) 3.5GHz, (b) 4.5GHz, (c) 6GHz, and (d) 9.5GHz. 57

71 XZ-plane 3.5GHz co-pol 3.5GHz cross-pol XZ-plane 4.5GHz co-pol 4.5GHz cross-pol (a) (b) XZ-plane 6GHz co-pol 6GHz cross-pol XZ-plane 9.5GHz co-pol 9.5GHz cross-pol (c) (d) Fig Measured radiation patterns of the band-notched design using the parasitic strips on xz-plane at (a) 3.5GHz, (b) 4.5GHz, (c) 6GHz, and (d) 9.5GHz. 58

72 YX-plane 3.5GHz co-pol 3.5GHz cross-pol YX-plane 4.5GHz co-pol 4.5GHz cross-pol (a) (b) YX-plane 6GHz co-pol 6GHz cross-pol YX-plane 9.5GHz co-pol 9.5GHz cross-pol (c) (d) Fig. 3.2 Measured radiation patterns of the band-notched design using the parasitic strips on yx-plane at (a) 3.5GHz, (b) 4.5GHz, (c) 6GHz, and (d) 9.5GHz. 59

73 Gain ( dbi ) the original UWB antenna with open-end slits with parasitic strips Frequency ( GHz ) 1 Fig Measured peak gain. 6

74 Chapter 4 Folded UWB Antenna Generally speaking, an UWB system will be devoted to high-speed data transmission applications, such as laptop computers or portable video devices. The computer-design engineers and industrial designers don t consider antenna in the early development so the space allocated for antenna is usually very small, and antenna s location is very bad. For example, the embedded antennas of laptop computer are usually rectangular so it has a large length to width ratio when they are placed in the laptop LCD [27]. It is usually very thin. Therefore, the antenna for a laptop computer is much smaller in volume than a cell-phone antenna. In like manners, UWB antenna will be applied and embedded into laptop or portable digital device for high-speed video-data transmission. Therefore, UWB antenna must be low profile, planar and easy fabrication. Although there are many UWB antennas emerged in the publishing, size reduction is still an important issue in the practical application. Size reduction will be our focus in the future. In this chapter, a pair of taper-curred shape slot antenna fed by CPW was chosen as an original design example. Then, it will be folded symmetrically for size reduction. The performance of this folded UWB antenna will be investigated in detail. The impedance bandwidth, radiation pattern, antenna gain will be explored for UWB applications Edge-Fed Folded-CPW Design for an edge-fed transmission line Since size reduction is an important issue for UWB system, it will be studied in this chapter. To shrink geometry size of a slot antenna fed by CPW, folded type will be a considerable candidate. If this slot antenna is symmetrically folded from the center line, the fed transmission line, say CPW, is also necessary to be folded. This folded coplanar 61

75 waveguide is called edge-fed folded CPW. It is shown in Fig.4.1. A traditional coplanar waveguide in essence needs an adequate ground plane on the left and right sides of the center feed line in order to work normally; on the contrary, the large ground planes for the edge-fed folded CPW are placed on the top and bottom of the printed-circuit board, and the center feeding line is on the side of it (the thickness of the circuit board, as shown in Fig4-8). Thus, folding the CPW-fed antenna symmetrically can reduce a half of the antenna size. Obviously, the dimension of the antenna can be lower down significantly if using the novel edge-fed folded CPW. To design a 5ohm edge-fed transmission line, TXLINE shown in below is a helpful tool. The electrical parameters of the edge-fed coplanar waveguide can be calculated using the CPW Ground provided by this software, that is to say, the electrical property of an edge-fed CPW is here supposed to be the same as the CPW with back grounding but the size is just a half. Considering the fabricated capability on slot width w by using a circuit board plotter (PC board plotter), the gap width is selected to be g =.3mm. Then, the TXLINE software shown below was invoked to design CPW transmission line with a back ground plane. The parameters: dielectric = FR4, dielectric constant = 4.3, height = 1.6mm, conductor = copper, conductivity = S/m, thickness=.36mm were chosen. If the center line width was selected to be w=1.6mm, the characteristic impedance is about Ohm 5 Ohm. 62

76 After designing the 5-Ohm CPW transmission line with a back ground plane, the design parameters h, g, and w will be included in the design of folded antenna using this edge-fed folded CPW. Shown in Fig.4.1 (a) is a folded CPW transmission line connected with SMA connectors. Fig.4.1 (b) is its cross section view. Here, the center conductor is arranged on the side wall with width w=1.6mm. The two slots with slot width g=.3mm are arranged the corners of the right-angles. Top and bottom plates are set as ground planes soldered to the outer conductor of the SMA fed by a coaxial cable. The photograph of the edge-fed folded CPW is shown in Fig. 4.1(c). It was fabricated on a 1.6mm FR4 substrate. For the sake of antennas-size reduction, the folded-slot UWB antenna will be fed by this kind of folded-cpw transmission line in the next section. Is the folded CPW really a 5-ohm transmission line? It must be verified. The characteristic impedance and insertion loss of the transmission line are very important. They must be investigated and its characteristic impedance will be extracted step by step. To measure the characteristic impedance Zs, we can refer to W. C. Johnson formula [28]: Z ( real) = Z open 1 R X ( 1 Ro ) (4.1) 2 2 o o Xo 63

77 Z ( real) = Z short 1 R X ( 1 Rs ) (4.2) 2 2 s s Xs W. C. Johnson Formula Zs = Zopen Zshort (4.3) R o :the real part of an open circuit. X o :the imaginary part of an open circuit. R s :the real part of a short circuit. X s :the imaginary part of a short circuit. Z ( ) open real :the real part of input impedance of an open circuit. Zshort ( real ):the real part of input impedance of a short circuit. Here, we measure the return loss of a length-width 4mm edge-fed transmission line by open and short terminators and then calculate the real part and imaginary part of the input impedance in case of terminated at the open and short conditions. Because the real part is dominant, the characteristic impedance can be easily calculated by the formula (eq. 4.3) Simulation and experimental Results The characteristics impedance can be obtained by W. C. Johnson Formula where return loss must be measured in advance. The measurement was performed by Agilent E571B Network Analyzer. To measure the return loss of a 4mm edge-fed CPW terminated at the open and short conditions, open and short terminator were connected with SMA connectors on one end and another end connected to E571B. Then, the characteristic impedance can be easily calculated from the formula (eq. 4.3). As shown in Fig. 4.2, the measured characteristics impedance of the edge-fed waveguide is nearly 5 ohm. The maximum and minimum of the impedance are about 52.9 and 47.3 ohm. It s variation is about only 5±2.8 ohm. Therefore, the edge-fed folded CPW has a good performance to transmit a microwave 64

78 signal. As shown in Fig. 4.1(c), its insertion loss can be measured if the two SMA connectors were connected to coaxial cables of the network analyzer. Fig 4.3 shows the performance of a section of folded CPW transmission line where its length L =4mm and its ground width W=3.3mm. Because the loss tangent of FR4 dielectric substrate is large, the insertion loss curve of the simulated result attenuates 1.7dB at 8.5GHz and 2.9 db at 8.5GHz in the measured result. Since only 1mm length of the edge-fed CPW line was included in the new-type folded antenna mentioned in the next section, its power loss will be small. Simultaneously, the edge-fed transmission line can be treated as a part of the folded antenna in the next section The Novel Folded UWB Antenna Antenna configuration In the beginning, a compact planar antenna that originally consists of a pair of symmetrically curved radiating-slot fed by coplanar waveguide is proposed and studied to achieve UWB requirement. Shown in Fig4.4 is its top-view pattern of the curved slot antenna. The proposed antenna is designed on a rectangular FR4 substrate with a thickness 1.6 mm, a relative permittivity 4.4 and a loss tangent.2, and size of mm 2. It was originally proposed by Prof. Sun in NTUT [2]. Its simulation result blue cure by HFSS was shown in Fig.4.5. UWB operation is successfully demonstrated to satisfy with system requirements before folding. Also shown in this figure, the simulation results for the structure where this pair of planar-curved slot antenna is symmetrically folded by the center feeding line. To be surprised, it also has an ultra-wide impedance-bandwidth. Shown in Fig4.6 are field distributions at different resonating frequencies. They show that this antenna can generate 65

79 multiple resonating modes. The current distributions of the folded slot UWB antenna at different resonating frequencies are presented in Fig The current is mainly distributed along the edge of the slot for all of the four different frequencies. The size can be shrunk to a half of original geometry size if this pair of planar curve slot is symmetrically folded by the center feeding line. The geometry of this novel folded slot UWB antenna is shown in Figure 4.8. The proposed antenna is also fabricated on a rectangular FR4 substrate and its outer-ground size of mm 2. Using HFSS to analyze and revise the folded antenna shown in Fig 4.8, impedance bandwidth and return loss of this shrunken UWB antenna will be investigated and further measurement will be performed to verify whether it can reach UWB system requirement for industrial applications. Position, aperture widths, and curved degree of the slots shown in Fig.4.8 must be suitably selected to control its performance. Design parameters r 1, r 2, R 1, and R 2 can be optimized by the simulation tool. In order to preserve broadband performance, design parameters must be analyzed. Here, fixing r 2 and R 2, the distance R 1 and radius r 1 will be the most important parameters which affect the antenna performance and need to be further investigated. To sum up, if the aperture length of one side is adjusted to be about a half-wavelength at the center frequency of the desired band, a narrow bandwidth can be notched out while still keeping good impedance matching for the rest of ultra-wide bandwidth. Finally, the proposed antenna is fabricated on a rectangular FR4 substrate with a thickness 1.6 mm, and size of mm 2. The geometry sizes for this taper-curved slot antenna folded with two half-ellipse-shaped plane are R 1 r 1 ( mm 2 ) and R 2 r 2 ( mm 2 ), respectively. The dimensions of the edge-fed CPW are length 11 mm and width 1.6 mm, the same as the substrate s high, and the gap distance g of the CPW is.3 mm. Its photograph is shown in Fig.4.8(c). Experiment was performed by a HP8363-PNA network analyzer. HFSS was invoked to 66

80 simulate its performance. SMA connector was also included in the simulation by HFSS. Shown in Fig. 4.9 and Fig. 4.1 are measured results compared with simulated results. The former shows voltage standing wave ratio and the later shows return loss. Shown in Fig.4.11 are phase responses of the return loss. Obviously, comparing the measured and simulated results, we can find that the wide bandwidth of the folded-slot antenna starts from 2.87GHz to 11GHz. Its measured bandwidth is more than 8.13GHz with VSWR < 2 (return loss less than -1dB). The peak resonant points are at 3.11GHz, 4.43GHz, 7.1GHz, 8.92GHz, and 1.31GHz. The simulated results shows a 8GHz bandwidth which starts from 3GHz to 11GHz with VSWR < 2 ( return loss less than -1dB ), and the measured peak resonant points are at 3.23GHz 3.98GHz 6.35GHz 8.68GHz and 1.13GHz. The difficulty on manufacturing this antenna results in some unavoidable error about resonating frequencies, but the behavior of the simulated and measured results are similar. The radiation patterns of the novel folded-slot antenna were measured in an anechoic chamber and gain are obtained by an EMCO 3115 double-ridged horn antenna. The measured frequency is start from 3GHz to 1 GHz, 5MHz a step, and then the antenna are measured at both vertical and horizontal polarization in XY, XZ, YZ planes. Fig shows the measured E plane (xz-plane) radiation patterns at 3GHz, 4.5GHz, 7GHz and 9 GHz for the novel folded-slot antenna, respectively. From the results, it is noticed that the E plane pattern exhibits some null at 9GHz. Fig shows the measured H-plane (yz-plane) radiation patterns. From the results, when it radiates at 7GHz or 9GHz, there are some nulls in the radiation pattern due to this is slightly weak electronic field or maybe spurious mode of folded CPW excited. The measured radiation patterns in E plane and H plane are nearly omni-directional at the lower bands, says 3GHz and 4.5GHz, and nulls appear at higher frequency bands. To conclude, pattern seems asymmetrical due to folding. Shown in Fig 4.14 are radiation pattern on the xy-cut. 67

81 Shown in Fig.4.15 are measured maximum antenna gains of the novel folded slot antenna. The antenna gain is measured in the anechoic chamber using a standard double ridge horn antenna (EMCO 3115). It is seen that the gain of the UWB antenna increases approximately linear, but some variation, over the entire bandwidth from the band 3GHz until 1GHz and reaches the maximum at 1GHz. The maximum gain is about 9dBi but the minimum gain is about 3dBi. Finally, Table Ⅲ listed average gains on the xy-plane for comparison Summary A novel folded-planar antenna with curved-shape slots fed by a folded CPW transmission line has been proposed in this chapter for UWB system applications. The folded coplanar waveguide investigated by measurement confirms is still a 5 ohm transmission line. The experimental results compared with the simulation results also demonstrated this folded-slot antenna still has UWB characteristics but radiation pattern shows some nulls on some cuts. Supposed some higher order modes of the folded CPW were excited to cause radiation. It must be further discussed or studied in the future. 68

82 (c) Fig. 4.1 Geometry of an edge-fed folded CPW. (a) Top layer view (b) cross-sectional view (c) photograph. 69

83 7 Characteristic Impedance Z o (ohm) Measured by a 4mm edge-fed folded CPW Frequency(GHz) Fig. 4.2 Measured characteristic impedance of the edge-fed waveguide. -5 Insertion Loss (db) Simulated S 21 Measured S Frequency(GHz) 11 Fig. 4.3 Measured and simulated insertion loss of the edge-fed waveguide. 7

84 Fig. 4.4 UWB antenna with a pair of symmetrically curved slot fed by CPW Return Loss (db) the folded design the original design Frequency(GHz) 11 Fig.4.5 simulation results for the originally unfolded UWB antenna compared with after- folded antenna. 71

85 (a) 3GHz (b) 4.5GHz 72

86 (c) 7GHz (d) 9GHz Fig. 4.6 Field distributions on the slot region at (a) 3GHz, (b) 4.5GHz, (c) 7GHz, (d) 9GHz. 73

87 (a) 3GHz (b) 4.5GHz 74

88 (c) 7GHz (d) 9GHz Fig. 4.7 Current distributions on the antenna at (a) 3GHz, (b) 4.5GHz, (c) 7GHz, (d) 9GHz. 75

89 (c) Fig. 4.8 Geometry of the novel folded UWB antenna. (a) Top layer view (b) cross-sectional view (c) photograph. 76

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