ISSN Vol.05,Issue.08, August-2017, Pages:

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1 ISSN Vol.05,Issue.08, August-2017, Pages: High Voltage Application using Flying Capacitor Based Hybrid LLC Converters S. MALATHI 1, C. HIMA BINDU 2, T. RANGA 3 1 PG Scholar, Dept of EEE(PED), St.Mark Educational Institution Society Group of Institutions, Anantapuramu, AP, India. 2 Assistant Professor, Dept of EEE, St.Mark Educational Institution Society Group of Institutions, Anantapuramu, AP, India. 3 Assistant Professor, Dept of EEE, St.Mark Educational Institution Society Group of Institutions, Anantapuramu, AP, India. Abstract: A propelled hybrid LLC arrangement full converter with incorporated flying capacitor cell is proposed in this paper to empower the high stride down transformation in the high info voltage applications.the inalienable flying capacitor branch in the essential side can adequately divide the essential switch voltage stretch contrasted and the half bridge LLC converters. Furthermore, the information voltage can be shared similarly and consequently between the two arrangement half bridge modules without extra adjust circuit or control systems due to the implicit flying capacitor cell. Additionally, the innate delicate exchanging execution amid wide load extend that exists in the LLC converters is as yet kept to decrease the exchanging misfortunes, which guarantees the high efficiency. Moreover, the proposed converter can be reached out to further reduction the switch voltage worry by utilizing stacked association. Finally, a 500V~640V input 48V -output 1kW prototype is built and tested to verify the effectiveness of the proposed converter. The results prove that the proposed converter is an excellent candidate for the high input voltage and high step-down DC/DC conversion systems. Keywords: LLC Converters, Flying Capacitor, DC/DC Converter. I. INTRODUCTION High voltage DC/DC converters have been widely employed in the three-phase communication power supply systems, DC transmission for large offshore wind farm and DC distribution as the interface between the DC transmission and distribution system or energy storage components [1]-[4]. However, high input voltage and high step-down DC/DC converters are still challenging in the power electronics community due to the technological limitations of semiconductors with high blocking voltage. Apart from the pulse-width modulation (PWM) converters, such as the phase-shifted converters [5]-[8], LLC series resonant converters are the popular candidates because they can achieve soft switching performance for all power devices from light to full loads. By adjusting the switching frequency, the controlled constant output voltage can be achieved with a wide input voltage variation [9]-[11]. However, in the high voltage applications, such as the secondary conversion stage following a three-phase 380V AC/DC converter, where the DC bus voltage is approximately 600V-800V, the primary switches of the conventional half-bridge LLC (HB LLC) converter, as shown in Fig. 1 (a), suffer from relatively high voltage stresses. In this case, high-voltage IGBTs are required. However, compared with MOSFETs, the switching frequency of IGBTs is limited. As a result, the high power density requirement cannot be fulfilled. In order to solve this problem, the input-seriesoutput-parallel technique provides a selectable solution to sustain the high input voltage and large output current requirements [12]-[14]. However, how to balance the voltage and current during each converter module is a big concern. Generally, additional control loop is required, which increases the control complexity and impacts the system response. Three-level LLC series resonant converters are another optimized solution to reduce the high voltage stress on the primary switches [15]. By employing the three-level configuration in the primary side, the voltage stress across each power switch is effectively clamped to only half of the input voltage. At the same time, the advantages of the conventional HB LLC series resonant converters, such as the soft switching performance, remain unchanged. However, the converter proposed in [15] can clamp the switch voltage stress in the leading leg to half of the input voltage, but that in the lagging leg still suffers from the high input voltage. The three-level LLC converter in [16] just employs one resonant tank, but additional diodes are required. Furthermore, the asymmetrical turn-off gate signals for the power switches make the modulation a little complex. The half-bridge based three-level LLC converter with asymmetrical turn-off gate signals in [16] is extended to fullbridge based three-level LLC converter for high power and wide input voltage application in [17]. Moreover, three-level resonant converter with double LLC resonant tanks is proposed for high input voltage applications [18]. In this converter, two resonant inductors and the transformer with multiple windings are necessary, which makes the topology a little complex. And the potential voltage unbalance may occur due to the mismatch of the resonance capacitors and transformer windings [19]. A part from the above-mentioned three-level configurations that have been applied to LLC 2017 IJIT. All rights reserved.

2 converters, the three-level flying-capacitor circuit, shown in Fig. 1(b), is another simple but effective solution for high voltage converter applications. By adapting the three-level flying-capacitor structure, the voltage stress of all the switches is effectively clamped to half of the high bus voltage [20]-[22]. However, the voltage unbalance, switching losses and output voltage regulation are the main issues associated with three-level flying-capacitor circuit [23]-[25]. S. MALATHI, C. HIMA BINDU, T. RANGA amount. Resonance in the LLC tank continues but no external energy is provided for the resonant tank There are two resonant frequencies for the proposed hybrid LLC converter. One is due to Lr and Cr, while the other is caused by Lm, Lr and Cr, which can be expressed as follows: II. PROPOSED METHOD As the proposed hybrid LLC converter consists of two half-bridge modules connected in series, and a flying capacitor is utilized to achieve the voltage balance between the two input capacitors. An LLC series resonant tank is connected across one switch, resulting in a square wave tank, whose amplitude is half of the high input voltage. A full-wave diode rectifier is employed as an example in the secondary side As shown in Fig1(d), S1, S2, S3 and S4 are the primaryside power switches; CS1, CS2, CS3 and CS4 are their parallel capacitors. CSS is the flying capacitor, and Lss is the inserted series inductor to limit the current surge of CSS, which consists of the circuit equivalent parasitic inductor and the additional inductor. In the LLC resonant tank, Cr is the series resonant capacitor; Lr is the series resonant inductor, which is composed of the equivalent leakage inductance of the transformer and an additional inductor; and Lm is transformer magnetizing inductance. Do1 and Do2 are the output rectifier diodes. By varying the switching frequency fs, there are three frequency ranges, namely fm<fsfr. Due to LLC characteristics [9], the converter can achieve ZVS for the primary switches in all three ranges, and achieve ZCS for the secondary diodes in the range of fm<fs<fs<fs Stage 1 [t0 - t1]. At t0, switches S1 and S3 are turned on with ZVS. The magnetizing current ilm linearly increases while the resonant current ilr increases sinusoidally. The difference between ilm and ilr flows through the transformer primary winding, enabling the rectifier diode Do1 to conduct. The transformer primary side voltage is clamped to the output voltage NVout. Therefore, the resonance occurs between only the leakage inductor Lr and the capacitor Cr, sourced by voltage(vc1 - NVout). The flying capacitor Css is paralleled with the input capacitor C1. At t0, VCss>VC1 since Css is smaller than C1 and icss is twice of ic1 in the last stage. Therefore, there is resonant charging current from Css to C1, which flows in the direction is3, the opposite direction to is1 (the positive current through each switch is downwards). So in stage 1, is3 is slightly larger than is1. The sum of the voltage of C1 and C2 is clamped to the input voltage Vin. Fig 1. Main waveforms of proposed converter (fm<fs). The transformer has a turn ratio of n=n1/n2. In the presented converter, switches S1 and S3 are driven with a constant 50% duty cycle simultaneously, while the switches S2 and S4 are both complementary to S1 and S3. There are two operational modes for the proposed hybrid LLC converter. When Css is paralleled with C1, C1 mainly provides energy for the LLC resonant tank, and C2 is charged by the same amount. The resonant tank obtains its energy from both C1 and Css. In the other mode, when Css is in parallel with C2, C2 provides the supplement energy for Css, and C1 is charged by the same

3 High Voltage Application using Flying Capacitor Based Hybrid LLC Converters charged and Cs2 and Cs4 are discharged. When the voltage across switches S2 and S4 drops to zero, a freewheeling current flows through the bypass diodes of the switches, which creates the ZVS conditions for switches S2 and S4. In stage 3, the flying-capacitor branch provides the freewheeling path for ZVS of switch S4. By selecting the appropriate dead time interval, when switches S2 and S4 are turned on, the voltage of the switches S2 and S4 is zero and ilr is positive, the switches can be turned on with ZVS. Stage 4 [t3 - t4]: At t3, switches S2 and S4 are turned on with zero switch voltage. The operation principle in stage 4 is similar to that in stage 1 Stage 5 [t4 - t5]: The operation principle in stage 5 is similar to that in stage 2, and the corresponding current relationship is: Stage 6 [t5 - t6]: The operation principle in stage 6 is similar to that in stage 3. When the proposed hybrid LLC converter operates at fs=fr or in the range of fs>fr, stages 2 and 5 do not occur. In the range of fs>f, the reverse recovery problem occurs in the secondary diodes. Fig 2. (Stage 1) [t0~t1], (b) (Stage 2) [t1~t2], (c) (Stage 3) [t2~t3], (d) (Stage 4) [t3~t4], (e) (Stage 5) [t4~t5], and (f) (Stage 6) [t5~t6]. Therefore, the total electric charge in C1 and C2 is constant. In this way, the electric charge that charges to C2 should always be equal to the one that discharges from C1. For stage 1-3, assuming that ic1 is the discharging current of C1 and ic2 is the charging current of C2, the following can be obtained. III. SIMULATION RESULTS The experimental results of above design example are shown in Fig. 3 to Fig. 4, which are in the full load condition. The input voltage and the voltages of the input capacitors C1 and C2 are shown in Fig.3. The voltage of each capacitor is 300V, which is only half of the input voltage, which illustrates that voltage balance is maintained due to the inherent voltage sharing mechanism. Also, the current is1 and is3 though switch S1 and S3 can be expressed as follows Stage 2 [t1 - t2]: At t1, the magnetizing current ilm equals to the resonant current ilr. The secondary current drops to zero naturally, assuming the stray inductance in the secondary is small enough to be ignored. Consequently, ZCS is achieved for the rectifier diode Do1. The transformer is no longer clamped by the output voltage Vout, and the magnetizing inductor Lm also participates in resonance. Since the resonant frequency is much higher than the switching frequency, ilr can be assumed constant The currents is1 and is3 through switches S1 and S3 can be expressed as Stage 3 [t2 - t3]: At t2, switches S1 and S3 are turned off. During the dead time period, since the resonant current ilr lags the gate signals of the power switches and maintains a constant value, the junction capacitors Cs1 and Cs3 are Fig 3. Experimental results of Vin, VC1 and VC2. The AC voltage waveforms of the flying capacitor and its resonant inductor are plotted in Fig.4 (a), and their DC offset voltages are correspondingly 300V and 0V. The voltage waveform of the resonant capacitor, the current waveform of the LLC resonant tank and the current waveform of the flying capacitor are shown in Fig.4 (b). Standard resonance occurs in the LLC resonant tank when the switching frequency is 100kHz (fs=fr), and the current through the resonant switch capacitor is half of the LLC resonant current and the same in each phase. This is in agreement with the above-mentioned analysis.

4 S. MALATHI, C. HIMA BINDU, T. RANGA Fig 4. Experimental results of flying capacitor. ZVS and ZCS waveforms at fs=100khz are plotted in Fig5.3. From Fig.5.3 (a)-(d), the switch voltage stress is about 300V, which is half of the 600V input voltage. This means the low voltage rated power switches can be employed. Also, the drain-source voltages of the switches fall to zero before the turn-on gate signal is applied, which indicates that ZVS is achieved for all the switches. In Fig.5 (e), the current through the secondary diode falls to zero before it turns off. This means there is no diode reverse-recovery problems for the secondary diodes. Fig 5. ZVS and ZCS waveforms of proposed converter (fs=100khz). The ZVS and ZCS waveform sat fs=87khz (when fs <fr) are given in Fig.5.4. The experiment results are still similar to those in Fig.5. This indicates that the proposed converter can achieve excellent circuit performance during wide load range as the conventional LLC converters.

5 High Voltage Application using Flying Capacitor Based Hybrid LLC Converters To illustrate the performance of the resonant inductor Lss, an experiment without Lss has been carried out and the current waveform of the flying capacitor is plotted in Fig. 7. In Fig. 5.6, the maximum value of the current icss is almost 15A, which is twice of that in Fig. where the resonant inductor Lss is added. In other words, the resonant inductor Lss effectively reduces the impulse current through the flying capacitor. Fig 8. Waveforms of icss without Lss. Fig 6. ZVS and ZCS waveforms of proposed converter (fs=87khz): (a) waveforms of switch S1, (b) waveforms of switch S2, (c) waveforms of switch S3., (d) waveforms of switch S4. (e) waveforms of secondary diode Do1. The secondary waveforms at fs=110khz (when fs >fr) are given in Fig.5. In Fig.4, the secondary current is forced to zero and the impulse voltage on the diode caused is almost 150V, which indicates that the diodes suffer from reverserecovery problem. Fig 9. simulink diagram. Fig 7. Secondary waveforms of proposed converter. Fig 10. Output Waveform of Capacitor.

6 IV. CONCLUSION In this thesis, the new input voltage auto-balanced hybrid LLC series resonant converters with flying capacitors have been proposed, which have the following clear advantages. Firstly, input voltage balance is achieved and the voltage of each switch is clamped and minimized, which enables the converters to operate with a high input voltage. Secondly, ZVS operation for the power switches and ZCS operation for the rectifier diodes are achieved during a wide input voltage and load conditions, which ensures high conversion efficiency. Thirdly, high and adjustable step-down voltage conversion is achieved due to the combination of the flyingcapacitor cell and the LLC series resonant converter. It can be concluded that the proposed hybrid LLC converter is an excellent candidate for the high input voltage, high step-down and high efficiency DC/DC systems. V. REFERENCES [1] G. P. Adam, S. J. Finney, K. Bell and B. W. Williams, New Breed of Network Fault-Tolerant Voltage-Source- Converter HVDC Transmission System, IEEE Trans. Power Electron., vol. 28, no.1, pp , [2] Y. Gu, W. Li, X. He, Frequency Coordinating Virtual Impedance for Autonomous Power Management of DC Microgrid, IEEE Trans. Power Electron., vol. 30, no.4, pp , Apr [3] P. Zhan,C. Li, J. Wen, et al, Research on hybrid multiterminal high-voltage DC technology for offshore wind farm integration, Journal of Modern Power Systems and Clean Energy, vol. 1, no, 1, pp , [4] Y. Gu, X. Xin, W. Li, X. He, Mode-Adaptive Decentralized Control for Renewable DC Microgrid With Enhanced Reliability and Flexibility, IEEE Trans. Power Electron., vol. 29, no.9, pp , Sep [5] J. Ke, X. Ruan and F. Liu, "An improved ZVS PWM three-level converter," IEEE Trans. Ind. Electron., vol. 54, no. 1, pp , 2007 [6] W. Li, P. Li, H. Yang and X. He, Three Level Forward- Flyback Phase Shift ZVS Converter with Integrated Series- Connected Coupled Inductors, IEEE Trans. Power Electron., vol. 27, no.6, pp , Jun [7] X. Ruan, D. Xu, L. Zhou, B. Li and Q. Chen, Zero- Voltage-Switching PWM Three-Level Converter With Two Clamping Diodes, IEEE Trans. Ind. Electron., vol. 49, no. 4, pp , Aug [8] W. Li, J. Qun, M. Ye, C. Li, Y. Deng, X. He, Modular Multilevel DC/DC Converters with Phase Shift Control Scheme for High Voltage DC-Based Systems, IEEE Trans. Power Electron., vol. 30, no.1, pp , Jan [9] B. Lu, W. Liu, Y. Liang, F. C. Lee and J.D. V. Wyk, "Optimal Design Methodology for LLC Resonant Converter," in Proc. IEEE Appl. Power Electron. Conf. Expo., pp , March [10] X. Xie, J. Zhang, C. Zhao, Z. Zhao and Z. Qian, Analysis and Optimization of LLC Resonant Converter With a Novel Over-Current Protection Circuit, IEEE Trans. Power Electron., vol. 22, no.2, pp , [11] R. Beiranvand, B. Rashidian, M. R. Zolghadri and S. M. H. Alavi, Using LLC Resonant Converter for Designing S. MALATHI, C. HIMA BINDU, T. RANGA Wide-Range Voltage Source, IEEE Trans. Ind. Electron., vol. 58, no. 5, pp , [12] D. Sha, Z. Guo and X. Liao, Cross-Feedback Output- Current-Sharing Control for Input-Series-Output-Parallel Modular DC DCConverters, IEEE Trans. Power Electron., vol. 25, no.11, pp , [13] J.-W. Kim, J.-S. Yon and B.H. Cho, Modeling, control, and design ofinput-series-output-parallel-connected converter for high-speed-train power system, IEEE Trans. Ind. Electron., vol. 48, no. 3, pp , [14] T. Qian and B. Lehman, Coupled Input-Series and Output-Parallel Dual Interleaved Flyback Converter for High InputVoltage Application, IEEE Trans. Power Electron., vol. 23, no.1, pp , [15] K. Jin and X. Ruan, Hybrid Full-Bridge Three-Level LLC Resonant Converter A Novel DC-DC Converter Suitable for Fuel Cell Power System, IEEE Trans. Ind. Electron., vol. 53, no. 5, pp , Oct