Three Phase Active Power Filter Based on Current Controlled Voltage Source Inverter
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1 Volume 4, Number 4, Three Phase Active Power Filter Based on Current Controlled Voltage Source Inverter E. E. EL-KHOLY*, A. EL-SABBE*, A. EL-HEFNAWY* and Hamdy M. MHAROUS** *Electrical Engineering Department, Faculty of Engineering, Menoufiya University Shebin El-Kom, Egypt **International Steel Rolling Mill Company (ISRM), Sadat, Egypt ABSTRACT: This paper presents a shunt active power filter to compensate reactive power and reduce the unwanted harmonics. A shunt active filter is realized employing three-phase voltage source inverter (VSI) bridge with common DC bus capacitor. The shunt active filter acts as a current source, which is connected in parallel with a nonlinear load and controlled to generate the required compensation currents. Two different proposed control methods for determining the reference compensating currents of the three-phase shunt active power filters based on proportional-integral (PI) controller and Artificial Neural Network (ANN) are presented. Current controller based on modified hysteresis current controller is used to generate the firing pulses. The proposed system is implemented using a high speed Digital Signal Processor (DSP). Experimental and simulation results with different nonlinear loads are presented to confirm the validity of the proposed technique. KEYWORDS: Active Power Filter, Current Control, Artificial Neural Network INTRODUCTION The increasing use of power electronic loads in industry and by consumers results in a considerable amount of harmonic injection, and lower power factor [-2]. Conventionally, passive filters have been used to eliminate current harmonics and to increase the power factor. However, the use of passive filter has many disadvantages [3- ]. Recently, because of the rapid progress in modern power electronic technology, the presented work was oriented mostly on the active filters instead of passive filters [6-8]. The basic difference between passive and active filters is that the active filters have the capability to compensate random varying currents [9-4]. One of the most popular active filters is the shunt active power filter (SAF). The SAF s have been researched and developed, that they have gradually been recognized as a feasible solution to the problems created by nonlinear loads. They are used to eliminate the unwanted harmonics and compensate fundamental reactive power consumed by nonlinear loads through injecting the compensation currents into the AC lines. In addition to eliminating harmonic currents and improving the power factor, SAF can keep the power system balance under the condition of unbalanced and nonlinear loads [9-]. enerally, The performance of SAF is based on three design criteria [-22]: i) design of power inverter; ii) types of current controllers used; iii) methods used to obtain the reference current. Many control techniques have been used to obtain the reference current. These techniques such as instantaneous reactive power theory [], notch filters [8], flux based controller [9], power balance theory [8]-[22], and sliding mode controller [6]-[7] have been used to improve performance of the active filters. However, most of these control techniques
2 44 ACTA ELECTROTEHNICA include a number of transformations and are difficult to implement [9]. This paper presents two different approaches used to calculate the reference current of the SAF. The first technique used the DC capacitor voltage with PI controller. Although, this method is simple to implement and achieves good results, it had drawbacks at unbalance conditions of the supply voltages. The performance of the SAF is achieved by the second technique, which depends on an ANN. The output of the ANN is used to generate the reference currents of SAF. The basic objective of this control is to provide a very precise solution to get the reference current even under unbalanced conditions of supply voltages. A hysteresis controller based on PWM current control is used, which is widely used due to its fast response. Comparisons of simulation results are presented for the two control strategies. The proposed control system with ANN is implemented using a DSP. Experimental results were presented to prove the effectiveness of the design of the control strategy. SYSTEM DESCRIPTION AND OPERATION Power Circuit Description: As shown in Fig., the SAF system consists of a three-phase voltage inverter with current regulation, which is used to inject the 3-phase AC supply compensating current into the power line. The VSI contains a three-phase Isolated ate Bipolar Transistors (IBT) with antiparalleling diodes. The VSI is connected in Parallel with the three-phase supply through three inductors L f, L f2 and L f3. The DC side of the VSI is connected to a DC capacitor, C, that carries the input ripple current of the inverter and the main reactive energy storage element. The DC capacitor provides a constant DC voltage and the real power necessary to cover the losses of the system. The inductors L f, L f2 and L f3 perform the voltage boost operation in combination with the capacitor, and at the same time act as the low pass filter for the AC source current. Then the SAF must be controlled to produce the compensating currents i f, i f2 and i f3 following the reference currents i f *, i f2 * and i f3 * through the control circuit. System Modeling: The representation of a three-phase voltages and currents of the VSI in Fig. () are as follows: the voltages V f, V f2 and V f3 supplied by the inverter as a function of the capacitor voltage, V c, and the state of the switches, 3 and are : V f V f2 V f3 S S 3 S L f D D D i f L f2 if2 C i f3 D D D L f3 S 4 = V c () where, 3 and represent three logic S 6 S 2 i f Vs Control V c i L Non linear Load Fig.. The proposed shunt active power filter.
3 Volume 4, Number 4, variables of the three legs of the inverter. The inverter conduction state is represented by these logics. Then the SAF currents can be written as: di L f f = Vs V f (2) dt di L f 2 f2 = Vs2 V f2 (3) dt di L f 3 f3 = Vs3 - V f3 (4) dt where i f, i f2 and i f3 are SAF currents and, 2 and 3 are the supply voltages. The voltage in the DC capacitor can be calculated from the SAF currents and switching function as follows: Vc [ = i i i C f 3 f2 f3 ] () The set point of the storing capacitor voltage must be greater than the peak value of the line neutral mains voltage in order to be able to shape properly the mains currents. THE PROPOSED CONTROL METHODS The quality and performance of the SAF depend mainly on the method implemented to V S X i * S i L generate the compensating reference currents. This paper presented two methods to get the reference current, which is key issue in the control of the SAF. Capacitor Voltage With PI Controller: The basic operation of this proposed control method is shown in Fig. 2. The estimation of the reference currents from the measured DC bus voltage is the basic idea behind the PI controller based operation of the SAF. The capacitor voltage is compared with its reference value, V c *, in order to maintain the energy stored in the capacitor constant. The PI controller is applied to regulate the error between the capacitor voltage and its reference. The output of PI controller is multiplied by the mains voltage waveform, 2, 3 in order to obtain the supply reference currents i s *, i s2 *, i s3 *. Then the supply reference currents are proportional to the mains voltages. The three-phase compensating reference current of SAF (i f *, i f2 *, i f3 *) are estimated using reference supply currents and sensed load currents. Proposed control method with ANN: This method is based on the recovering of the fundamental active phase currents in i * f i f Hysteresis current controller 4 V c V * c PI V S2 V S3 X i L2 i * S2 i L3 i * f2 i f i f Distribution circuit 3 6 X i * S3 i * f3 2 Fig. 2. The proposed control system.
4 442 ACTA ELECTROTEHNICA i L (t - (t) AN W X i p( t i f (t) Fig. 3. The basic principle diagram of the proposed method. the load. Once these currents are obtained, they are subtracted from the total load currents to get the desired reference waveform. Let to assume that the nonlinear load current can be represented as: n n= 2 i = i cosϕ sin wt i sinϕ cos wt i l sin( nwt ϕ ) = ip ( t) iq ( t) ih ( t) (6) where i p (t) represents the active current, i q (t) represents the reactive current and i h (t) represents the sum of all harmonic currents. If the SAF can generate i f (t) which is equivalent to i q (t)i h (t), then the mains only needs to supply the active current i p (t), which is a sine wave and in phase with the supply voltage (t). Equation (6) becomes; i f (t) = i L (t) i p (t) = i L (t)- I cos(φ) sin(ωt) where sin(ωt) = /V m (7) i f (t) = i L (t)-w (t) where W=I cos(φ)/v m (8) According to calculation of W, the reference current of the SAF is easy to calculate which changes according to the nonlinear load current and the supply voltage. The calculation of W is based on neural n network. The basic principle diagram of this control algorithm is shown in Fig. 3. The ANN in Fig. 3 consists of a neuron that is an adaptive linear element, the weigh of the neuron is equivalent W and the bias is equivalent to zero. The input of the system is the supply voltage (t), the output of the ANN is fundamental active current i p (t) and the output of the system is the SAF reference current i f *. According to The Widrow-Hoff learning rule, the update weight equation as follows: W(n) =W(n-) η i fr (n-) (9) where the term η i fr (n-) is represents the change of weight. i fr (n) = i L (n)-w(n) (n) () From equations (9) and (), the proposed control method for phase-a is designed, as shown in Fig. 4. The neural network learning and training by Widrow-Hoff learning rule is used to obtain the weight (W). Then the SAF reference current can be calculated from the equation (). After calculating the reference current for the SAF, the hysteresis current controller has been presented to provide the i L - i p * f i X W Z - X η Z - Fig. 4. The proposed control method for phase-a.
5 Volume 4, Number 4, signals pulses of the VSI that controlled SAF currents. Hysteresis current controller: Hysteresis current controller with fixed band derives the switching signals of the inverter from the comparison of the current error to keep the current within the hysteresis band. Then the SAF reference currents i f *, i f2 *, i f3 * compared with SAF feedback currents i f, i f2, i f3 and the error signals are operated by the hysteresis current controller to generate the firing pulses which activate the inverter power switches in a manner that reduces the current error. Let (h b ) is the width of the hysteresis band around respective phase SAF currents, then the equation for the fixed band as follows: i f * = i max sin(ωt) () i up = i f * h b (2) i low = i f * - h h (3) where i up is the upper band, i low is the lower band. If the SAF current feedback i f > (i f * h b ) then = -, which means that the inverter output voltage switches to negative, in order to keep the current within the hysteresis band. If the SAF current feedback i f < (i f * - h b ) then =, which means that the inverter output voltage switches to positive, in order to increase the actual current. Similarly, the switching logic of other two-phase 2, 3 is derived. SIMULATION RESULTS A number of simulation results with different operating conditions were developed. In Fig., SAF is connected in parallel with nonlinear load. Then, the threephase controlled rectifier with resistive load has to be compensated by the SAF. In order to limit maximum slope of the rectifier current, a smoothing inductor, L r, has been inserted before the rectifier. Also, it is prevented the inverter saturation even in correspondence of rectifier commutations. Since the compensation of the filter depends upon the firing angle (α) of the rectifier, two operative conditions have been considered. The uncontrolled rectifier is the first case, while the second one is the controlled rectifier with firing angle α=4. The parameters of system are reported in Appendix. Simulation with uncontrolled rectifier (α = ): Steady state operations: The simulation results in steady state operation are presented. Figure 6 shows the performance of the SAF system using PI controller. Waveform of the source current without SAF is shown in Fig. 6-a. Figure 6-b shows the source current with SAF superimposed to the supply voltage. Figure 6-c shows the compensation current. The α 3-phase AC supply i s i t L r i L R L i f L f Controlled rectifier C Fig.. SAF connected in parallel with nonlinear loads.
6 444 ACTA ELECTROTEHNICA in Amp. ( Amp/div- 2 volt/div) a. Source current without filter b. Source current and source voltage with filter. is in Amp c. Compensating current. in volt d. Capacitor voltage with its reference. Fig. 6. Simulation results of proposed SAF using PI controller. capacitor voltage superimposed to its reference is shown in Fig. 6-d. It is noticed that the supply current in phase with the supply voltage, and the capacitor voltage follow its reference. Figure 7 shows the performance of the SAF using ANN algorithm. The source current with the supply voltage is shown in Fig. 7-a. Figure 7-b shows the compensation current. This figure indicates that the supply current in phase with supply voltage. In order evaluate the good performance of the control, total harmonic distortion (THD) is calculated for the source current before and after compensating. The calculation shows that THD improved from 24.4 % without the SAF to 4.66 % for using PI controller and 2.96 % for using ANN control with the SAF. The distortion in supply current with ANN is less than in case of PI controller method.
7 Volume 4, Number 4, ( Amp/div- 4 volt/div) a. Source current with SAF. in Amp b. Compensation current. i s Fig. 7. Simulation results of proposed SAF using ANN. in Amp. ( Amp/div- 2 volt/div) in volt a. Source current without filter b. Source current and source voltage with filter. i s c. Capacitor voltage with its reference. Fig. 8. Simulation results for a step change in load using PI controller.
8 446 ACTA ELECTROTEHNICA (2 Amp/div- 4 volt/div) 2 i s a. Source current and source voltage. in Amp. ( Amp/div- 2 volt/div) in Amp b. Weight magnitude. - Fig. 9. Simulation results for a step change in load using ANN a. Source current without SAF b. Source current and source voltage with filter c. Compensating current. Fig.. Simulation results of proposed SAF using PI controller. i s
9 Volume 4, Number 4, Step change in load To observe the regulating process in two control methods, the simulation has been studied under the step change of the nonlinear load current. Figure 8 shows the source current without SAF, the source current with SAF superimposed to the supply voltage, and the DC capacitor voltage in case PI controller. Once the time interval of transient is finished the DC capacitor can be recovered to be reference value, and vice versa. Figure 9 gives the simulation results in case of ANN algorithm. The supply current and voltage are in phase during the load step change, as shown in Fig. 9-a. Figure 9-b shows the weight magnitude will return to its value as the step change period finished. Simulation with firing angle α = 4 : The simulation results in case of steady state operation with static load when α=4 are presented. Figure shows the simulation waveforms of the source current without SAF, the source current with SAF superimposed to the supply voltage, the SAF current, and the capacitor voltage using PI controller. Figure shows the same waveforms for ANN control besides the weight function W. The final values of THD of supply current before and after compensating are listed in Table A. The t/div) supply current has improved significantly when using the ANN algorithm. Table A. Without SAF With SAF (PI With SAF controller) (ANN) α = THD% = THD% = THD% = α = 4 THD% = THD% =.8 THD% = Simulation results with unbalance in voltage: PI-Controller Figure 2 shows the simulation results under the unbalance three-phase supply voltages. The supply voltage and current without SAF shown in Fig. 2-a. Figure 2-b shows the SAF current superimposed to its reference current. The capacitor voltage and its reference are shown in Fig. 2-c. In order evaluate the good performance of the control, total harmonic distortion (THD) is calculated for the source current before and after compensating. The calculation shows that THD improved from % without SAF to.439 % for using SAF with PI controller. i s ( Amp/div- 2 vol in Amp. in Amp a. Source current and source voltage b. Compensating current. Fig.. Simulation results of proposed SAF using ANN.
10 448 ACTA ELECTROTEHNICA ( Amp/div- 2 volt/div) is a. The supply current and voltage with SAF in Amp - I f * I f b. The SAF current composed to its reference current. in volt c. The capacitor voltage composed to its reference. Fig. 2. Simulation results under unbalance three-phase supply voltage. ANN Algorithm Figures (3), shows the simulation results when the supply voltages are unbalance. The results have been obtained under the same pervious condition of the PI controller. Through the figures and calculation the THD of source current with SAF, the THD is reduced from.439% (value obtained by means of PI controller) to 6.98% (value obtained by this control algorithm). EXPERIMENTAL RESULTS Active power filter prototype for experimental purpose has been set up and tested. The prototype system is shown in ( Amp/div- 2 volt/div) Vs i s Fig. 3. The supply current and voltage with SAF..4
11 Volume 4, Number 4, phase AC supply 3 V c Base drive i s i s2 i s3 PWM Hysteresis current controller i * s i * s2 i * s3 PC computer Data-bus D/A converter A/D converter V c 2 3 Reference generator Control algorithm DSP board (ds2) Non-Linear Load Fig. 4. Digital Signal Processing (DSP) system hardware. Fig. 4. The power circuit is constructed IBT s. The voltage isolators sense the unit vector of supply voltages and DC capacitor voltage. Also, the current isolators using LEM hall-effect current sensors sense the compensating current. The control scheme is implemented in real time using a 32-bit digital signal processor (DSP-TMS32C3). The control algorithm generates three-phase reference supply currents, which are the output of DSP. The three-phase reference supply currents are fed to hysteresis PWM current controller. The resulting pulses are gating the IBT s of the SAF. Figure shows experimental waveforms for the load condition of uncontrolled rectifier. The power factor is improved to unity using ANN control technique. Figure -a shows the supply voltage and current without ASF. Figure -b shows the supply phase voltage, supply current and its reference. It is clear from Fig. -b that the supply current is almost sin waveform and follows the supply voltage in its waveform shape with almost a unity displacement power factor. Figure -c shows the compensating current. From these figures, it is clear that the effectiveness of the proposed controller for active power filter. CONCLUSION A control strategy for high performance three-phase shunt active filter with unity input power factor has been proposed. Two control strategies are considered, the first is PI-controller and the second is based on ANN. The control scheme using three independent hysteresis current controllers has been implemented. The operation and modeling of the SAF have been described.
12 4 ACTA ELECTROTEHNICA ( A/div- 2 V/div) is -a. Source and voltage without filter. Time (ms) (3 A/div- 4 V/div) -b. Source current, its reference and source voltage. Time (ms) (2. A/div) -c. Compensation current. Time (ms) Fig.. Experimental results of proposed SAF. An experimental SAF has been carried out on DSP to explore the advantages and practical implementation with the proposed control strategy. The experimental and simulation results showed that the proposed system has a good performance in the harmonic compensation and improving the input power factor. REFERENCES. Dixon,J.W., Venegas,., and Moran,L.A., A series active power filter based on a sinusoidal currentcontrolled voltage-source inverter. IEEE Transactions on Industrial Electronics 997; 44 (): Jeong,S., Woo,M., DSP-based active power filter with predictive current control. IEEE Transactions on Industrial Electronics 997; 44 (3): Buso,S., Malesoni,L., Mattavelli,P., Comparison of current control techniques for active filter applications. IEEE Transactions on Industrial Electronics 998; 4 (): Cheng,P., Bhattacharya,S., Divan,D., Line harmonics reduction in high-power systems using square-wave inverters-based dominant harmonic active filter. IEEE Transactions on Power Electronics 999; 4 (2).. Huang,S., Wu,J., A control algorithm for threephase three-wired active power filters under nonideal mains voltages. IEEE Transactions on Power Electronics 999; 4 (4): Verdelho,P., Marques,., Four-wire currentregulated PWM voltage converter. IEEE Transactions on Industrial Electronics 998; 4 ():
13 Volume 4, Number 4, Kamran,F., Habetler,T., Combined deadbeat control of a series-parallel converter combination used as a universal power filter. IEEE Transactions on Power Electronics 998; 3 (): Dastfan,A., osbell,v., Platt,D., Control of a new active power filter using 3-d vector control. IEEE Transactions on Power Electronics 999; 4 (4): Chandra,A., Singh,B., Al-Haddad,K., An improved control algorithm of shunt active filter for voltage regulation, harmonic elimination, power-factor correction, and balancing of nonlinear loads. IEEE Transaction on Power Electronics 2; (6): Buso,S., Malesani,L., Mattavelli,P., Veronese,R., Design and fully digital control of parallel active filters for thyristor rectifiers to comply with IEC standards, IEEE Transactions on Industry Applications 998; 34 (3): Fujita,H., Akagi,H., The unified power quality conditioner: the integration of series-and shuntactive filters. IEEE Transactions on Power Electronics 998; 3 (2): Salo,M., Tussa,H., A vector controlled currentsource PWM rectifier with a novel current damping method. IEEE Transactions on Power Electronics 2; (6): Barbosa,P., Santisteban,J., Watanabe,E., Shuntseries active power filter for rectifier's ac and dc sides. IEE Proc.-Electrical Power Application 998; 4 (6): Peng,F., Harmonic source and filtering approaches. IEEE Transactions on industry applications 2; 7(4): Akagi,H., Kanazawa,Y., Nabae,A., Instantaneous reactive power compensators comprising switching devices without energy storage components. IEEE Transactions on Industry Applications 984; IA-2: Saetieo,S., Devaraj,R., Tomey,D., The design and implementation of a three-phase active power filter based on sliding mode control. IEEE Transactions on Industry Applications 99; 3: Singh,B., Al-Haddad,K., Chandra,A., Active power filter with sliding mode control. In: proc. Inst. Elect. Eng., eneration Transm. Distrib. 997; 44: Rastogi,M., Mohan,N., Edris,A., Hybrid-active power filtering of harmonic currents in power systems. IEEE Transactions on Power Delivery 99; (3): Bhattacharya,S., Veliman,A., Divan,A., Lorenz,R., Flux based active power filter controller. In: Proc. IEEE-IAS Annual Meeting Record, 99; Jou,H., Performance compression of the threephase active power filter algorithms. In: proc. Inst. Elect. Eng., eneration Transm. Distrib. 99; 42: Dixon,J., arcia,j., Moran,I., Control system for three-phase active power filter which simultaneously compensates power factor and unbalanced loads. IEEE Transactions on Industrial Electronics 99; 42 (6): Saetieo,S., Devaraj,R., Tomey,D., A new control approach to three-phase active power filter for harmonics and reactive power compensation. IEEE Transactions on Power System 998; 3 (): Appendix Parameters of the shunt active filter: Input voltages = 38 V Rectifier load resistance R dc = Ω Rectifier input inductor L r = mh Rectifier load current i L = Amp. SAF inductor L f = 3 mh SAF DC link capacitor C = 2 µf SAF DC link voltage V c = 7 V
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