Beamspace MIMO for Millimeter-Wave Communications: System Architecture, Modeling, Analysis, and Measurements

Size: px
Start display at page:

Download "Beamspace MIMO for Millimeter-Wave Communications: System Architecture, Modeling, Analysis, and Measurements"

Transcription

1 1 Beamsace MIMO for Millimeter-Wave Communications: System Architecture, Modeling, Analysis, and Measurements John Brady, Student Member, IEEE, Nader Behdad, Member, IEEE, and Akbar Sayeed, Fellow, IEEE Abstract Millimeter-wave wireless systems are emerging as a romising technology for meeting the exloding caacity requirements of wireless communication networks. Besides large bandwidths, small wavelengths at mm-wave lead to a high dimensional satial signal sace, that can be exloited for significant caacity gains through high dimensional multile-inut multile-outut (MIMO) techniques. In conventional MIMO aroaches, otimal erformance requires rohibitively high transceiver comlexity. By combining the concet of beamsace MIMO communication with a hybrid analog-digital transceiver, Continuous Aerture Phased (CAP) MIMO achieves near-otimal erformance with dramatically lower comlexity. This aer resents a framework for hysically-accurate comutational modeling and analysis of CAP-MIMO, and reorts measurement results on a DLAbased rototye for multi-mode line-of-sight communication. The model, based on a critically samled system reresentation, is used to demonstrate the erformance gains of CAP-MIMO over state-of-the-art designs at mm-wave. For examle, a CAP-MIMO system can achieve a sectral efficiency of 10-0 bits/s/hz with a 17-31dB ower advantage over state-of-the-art, corresonding to a data rate of Gbs with 1-10GHz system bandwidth. The model is refined to analyze critical sources of ower loss in an actual multi-mode system. The rototye-based measurement results closely follow the theoretical redictions, validating CAP- MIMO theory, and illustrating the utility of the model. Index Terms Analog beamforming, Discrete lens array, Gigabit wireless, High dimensional MIMO, Lens Antennas, Millimeter-wave communication, MIMO systems, Transceiver comlexity I. INTRODUCTION The raid roliferation of consumer wireless devices is creating a sectrum crisis at the current wireless frequencies. A variety of communication and signal rocessing techniques are currently being ursued for interference management and efficient use of the available sectrum, including cognitive radio and multi-antenna technology. Desite these efforts, there is a growing consensus that meeting the dramatically increasing data demands of wireless devices and alications will require transformative new technologies and methodologies. While they have been under research for several years [1], driven by advances in enabling technology, millimeter-wave (mm-wave) wireless systems, oerating from GHz, are emerging as a romising technology for meeting the exloding bandwidth requirements by enabling multi-gbs seeds []. The authors are with the Deartment of Electrical and Comuter Engineering, University of Wisconsin-Madison, Madison, WI USA This work is suorted in art by the National Science Foundation under Grant ECCS-10568, and the Wisconsin Alumni Research Foundation (WARF). Mm-wave systems offer unique oortunities for enabling high data rate wireless communication. First, moving to the mm-wave regime oens u large ortions of unused sectrum that can suort orders of magnitude larger bandwidths (10s of GHz) comared to existing systems. Second, exloiting the satial dimension is articularly romising: for a given antenna size A the small wavelength,, leads to a dramatic increase in the dimension of the satial signal sace, n = 4A. In addition to creating narrow, high-gain beams, the highdimensional satial signal sace can be exloited by multileinut multile-outut (MIMO) transceivers for significant imrovement in sectral efficiency through satial multilexing of simultaneous data streams. Due to the highly directional nature of roagation, line-of-sight (LoS) roagation lays an imortant role at mm-wave. While the satial multilexing advantages of MIMO have traditionally rested on multiath roagation [3] [5], mm-wave systems can exloit MIMO oeration even in LoS roagation for both oint-to-oint (PP) and oint-to-multioint (PMP) network links. While the dimension of the satial signal sace, n, can be quite high ( ), due to the highly directional roagation, the actual number of satial communication modes,, is much smaller: n. However, current state-of-the-art mm-wave systems fail to take full advantage of the satial dimension. Dish systems with continuous aerture antennas, such as the commercial systems offered by Siklu (htt:// and LightPointe (htt:// for mm-wave wireless backhaul, exloit narrow, high gain beams, but only suort a single data stream ( = 1). Conventional MIMO systems with widely saced discrete antennas (due to comlexity considerations) [6] [8] suort satial multilexing ( >1), but suffer from reduced gain and comromised security due to grating lobes. In conventional MIMO designs, full exloitation of the satial dimension requires critically (half-wavelength) saced antenna arrays, but this aroach suffers from a rohibitively high transceiver comlexity on the order of n. The recently roosed Continuous Aerture Phased MIMO (CAP-MIMO) transceiver architecture [9], [10] is based on the concet of beamsace MIMO communication [5] to enable efficient access to the communication modes of an n- dimensional mm-wave link. CAP-MIMO uses a hybrid analogdigital front-end in which a high-resolution discrete lens array (DLA) is used for analog satial beamforming. The DLAbased front-end enables CAP-MIMO to achieve near-otimal erformance with dramatically lower transceiver comlexity,

2 on the order of, comared to the order n comlexity of conventional MIMO. Furthermore, initial theoretical results show that CAP-MIMO can otentially deliver very comelling erformance gains over the state-of-the-art in terms of ower efficiency, caacity, and oerational caability. This aer builds on the theoretical foundations of CAP- MIMO in [9], [10], overviewed in Sec. II, to investigate the modeling, design, and analysis of a hysically realizable CAP- MIMO system. There are three main contributions reorted in this aer. First, we outline a general framework for accurate comutational modeling of the CAP-MIMO system in Sec. III, including modeling of the DLA, channel, and overall system. Second, the comutational framework is used to analyze the erformance of a CAP-MIMO system in Sec. IV. In Sec. V, we resent romising measurement results based on a DLAbased rototye system to validate and refine the basic CAP- MIMO theory and the modeling framework. Sec. VI resents a discussion of results and concluding remarks. Notation: Lowercase boldfaced letters (e.g., h) denote comlex-valued column vectors, and uercase boldfaced letters denote matrices (e.g, H). Elements of vectors or matrices are not boldfaced; e.g, h(`) and H(`, m). tr(h) denotes the trace, det(h) the determinant, H T the transose, and H H =(H T ) the comlex conjugate transose of H. The notation x CN(m, ) denotes a comlex Gaussian vector x with mean m = E[x] and covariance matrix = E[(x m)(x m) H ], where E[ ] denotes the statistical exectation oerator. We will use the terms modes and beams interchangeably. II. CAP MIMO OVERVIEW CAP-MIMO theory is based on a finite-dimensional system reresentation induced by critical samling of the antenna aertures. This rovides a comlex baseband system model that serves as the foundation for theoretical, comutational, and exerimental design and analysis. Consider a link consisting of a 1D linear transmit antenna of length L T and a 1D linear receive antenna of length L R searated by a link length of R L T,L R oerating at carrier frequency f c, with wavelength. Critical samling results in signal sace dimensions n T and n R, where n 1D = L, (1) which is roortional to the antenna gain [5], [9] [11]. Critical samling of the transmit and receive aertures allows the system to be modeled in the aerture domain as r = Hx + w () where x = [x(1),...,x(n T )] T is the transmitted aerture domain signal vector, r =[r(1),...,r(n R )] T is the received aerture domain signal vector, H is the n R n T aerture domain channel matrix reresenting the roagation channel couling the transmitter and receiver antennas, and w is a n R 1 vector reresenting noise and interference. The model () serves as a direct reresentation for conventional MIMO systems with critically saced discrete antenna arrays, and as a virtual model for continuous aerture systems, such as CAP- MIMO, with no loss of information [5], [1]. A. Otimal Beamsace Communication Modulation of data onto orthogonal basis waveforms is a fundamental concet in communication theory, and orthogonal satial beams form an otimal basis for the satial dimension [5], [9], [10]. In articular, the satial signal sace of an antenna of dimension n can be associated with n orthogonal beams. The aerture domain system model () can be equivalently reresented in beamsace as r b = H b x b + w b, H b = U T b,rhu b,t (3) where x b = [x b (1),...,x b (n T )] T is the transmitted beamsace signal vector, r b =[r b (1),...,r b (n R )] T is the received beamsace signal vector, H b is the n R n T beamsace channel matrix that reresents the couling between the satial beams at the transmitter and receiver, U b,t is the n T n T transmit beamforming matrix, U b,r is n R n R the receive beamforming matrix, and w b = U T b,r w is the n R 1 beamsace noise vector. The transmitted (aerture) signal vector is related to the beamsace signal vector as x = U b x b = P n i=1 u( i)x b (i). Each of the n beamsace signals, x b (i), is maed onto a corresonding orthogonal beam reresented by a column u( i ) of U b. These n beams cover the entire (one-sided) satial horizon, / ale ale /, where is the satial angle relative to broadside. Each u( ) is an array steering/resonse vector, or array factor, that reresents an (all-hase) comlex satial sinusoid whose frequency, to the hysical angle via 1/ ale ale 1/, is related = d sin( )= 1 sin( ) (4) where d = / denotes the aerture domain samle sacing. The n elements of u( ) are given by u i ( ) =e j i,ii(n) ={i (n 1)/ :i =0,,n 1} (5) where I(n) is a set of n indices symmetrically arranged around the origin. For critical samling, d = /, there is a one-to-one maing between [ 1/, 1/] and [ /, /]. If the satial frequencies/directions for the n beams, i, are uniformly saced with sacing o = 1 n = L () o L (6) then the resulting u( i ) are orthogonal to each other. We note that o $ o is a measure of the satial resolution or beamwidth of an critically saced n-element array of length L [9], [13]. U b is exlicitly constructed with orthogonal u( i ) column vectors as U b = U df t = 1 n [u( i )] ii(n), i = i o = i n which is the n n unitary discrete Fourier transform (DFT) matrix, U H df t U df t = U df t U H df t = I, and the hysical angles i corresonding to the i are the orthogonal satial angles that cover the entire satial horizon [ /, /] [9], [10]. (7)

3 3 B. Beamsace Channel Modeling The array steering vector defined in (5) can also be used to define the channel and gain insight into the nature of the low-dimensional communications subsace. For a PP LoS link, the received aerture domain signal can be related to the signal at i th transmitter samle oint by the n R 1 steering vector u R ( ch,i ), where ch,i is the satial frequency corresonding to the angle subtended by the i th transmit samle oint. The channel is constructed with the u R ( ch,i ) column vectors as [9], [10] H los =[u R ( ch,i )] ii(nt ), ch,i = i 4R. (8) The sacing between the channel frequencies, 4R, is much smaller than the orthogonal sacing, o = 1 n = L, so the channel vectors will be correlated and the channel will be low rank. The channel rank can be aroximated by the number of beams that coule strongly between the transmitter and receiver. This is calculated by considering the satial bandwidth of the receive aerture, max, where max =0.5sin( max ) L R 4R [9], [10]. Thus los,1d = max L R L T L R o R o R orthogonal beams, with beamwidth o, will coule strongly from transmit to receive, where los is a fundamental quantity known as the Fresnel number in otics [14]. In general, los n T,n R due the relatively large link length, R L T,L R, which limits the angular extent of the receive aerture. We note that los,1d is a conservative estimate. In general [ los,1d, los,1d + 1] orthogonal beams will coule strongly. This will be revisited in Sec. IV. For -D lanar aertures with A = L x L y the channel matrix is the Kronecker roduct of 1-D channels consisting of a transmit antenna of length L T,x or L T,y, a receive antenna of length L R,x or L R,y, and link length R [9]. This results in n D = 4A (9) = n x n y, los,d = A T A R R = los,x los,y. (10) In the case of multiath, the channel can be modeled as [5]: H m = N X i=1 iu R ( R,i )u H T ( T,i ) (11) where N is the number of roagation aths, T,i is the angle of dearture, R,i is the angle of arrival, and i is the comlex gain of the i th ath. Here the low dimension of the communication subsace is less readily aarent. However, since we exect the roagation aths to be sarse at mmwave [], [15], due to the highly directional nature of roagation and clustered scattering, the roagation aths will only lie within m n T,n R orthogonal beams (see the concet of virtual ath artitioning in [5]). In general, H will be a combination of LoS and multiath comonents with = los + m. However, since LoS roagation lays an imortant role in mm-wave roagation, for the remainder of the aer we will focus on LoS links where H = H los, = los, and los > 1. A beamsace channel matrix (H b ) corresonding to a DLAbased LoS link with 1D antennas, with n T = n R = 6 and = los =, is illustrated in Fig. 1. These link secifications are derived from the rototye discussed in Sec. III-D. There are two key observations. First, only a sub-matrix H b of the 6 6 matrix H b is non-zero, reflecting the dimensional communication subsace. Second, Hb is nearly diagonal indicating that the orthogonal Fourier satial basis vectors used for beamforming in U b (and aroximated by the DLA) serve as aroximate eigenfunctions of the LoS channel. We will return to these observations in Sec. IV. RX BEAM DIRECTION (DEG) TX BEAM DIRECTION (DEG) Fig. 1: Normalized Contour lot (30 levels) of H b for a 1D DLA-based link with n T = n R = 6 and = C. Performance Gains The advantages of CAP-MIMO lie in its ability to otimally exloit the satial dimension at mm-wave, through high antenna gain roortional to n and satial multilexing of data streams over the dimensional communication subsace. Basic CAP-MIMO theory develoed in [9], [10] rovides accurate and insightful closed-form caacity aroximations to comare CAP-MIMO with the current state-of-the-art in LoS mm-wave communication. Consider a PP LoS link with identical transmit and receive antennas of dimension n that can suort n communication modes. For a given oerating transmit SNR (ratio of the total transmit signal ower to the received noise variance), denoted by, the caacity of CAP- MIMO can be aroximated as C CAP MIMO log (1 + rx ) bits/s/hz (1) where the leading term reflects the satial multilexing gain, and rx = n reflects the receive SNR gain, G = n over the receive SNR of an isotroic antenna,. An intuitive interretation of the receive SNR gain is G = 1 n n reflecting the equal division of transmit SNR among the beams, the transmit array gain n, and the receive array gain n (aroximately orthogonal beams are acked into the receive aerture). The caacity of the current state-of-the-art mm-wave systems, with the same antenna size and frequency, can be considered as a secial case of (1). Continuous aerture Dish systems ossess high antenna gain, but no satial multilexing

4 4 gain, so the leading term is droed and n ale rx ale n. The lower bound corresonds to los > 1 and the uer bound to los ale 1. Conventional widely saced MIMO uses discrete antennas, each with gain G MIMO [6] [9]. This results in full satial multilexing gain but reduced SNR gain rx = G MIMO, 1 ale G MIMO < n. The uer bound on G MIMO reflects the fact that the discrete antennas must be smaller than 1/ th of the aerture. Fig. comares the caacity of the three systems (G MIMO = 5dB) for a LoS link consisting of lanar antennas with 40cm 40cm aertures, oerating at f c = 80GHz, searated by a link length R = 4.67m (e.g., a mm-wave backhaul link), resulting in =( los,1d + 1) =4and n = As evident, CAP-MIMO can deliver significant caacity/snr gains relative to the state-of-the-art. In articular, CAP-MIMO can achieve a caacity of 10-0 bits/s/hz with a ower advantage of 17-31dB. This corresonds to a data rate of Gbs with 1-10GHz system bandwidth. Additionally the caacity of a Dish system using the same aerture size and link length oerating at 3GHz (n = 64 =1) is included for reference. While the caacity of the 3GHz link is exceeded at 80GHz by both Dish (through higher antenna gain) and conventional widely saced MIMO (through satial multilexing but with reduced antenna gain), CAP-MIMO is able to leverage both satial multilexing and the full antenna gain to obtain the most imrovement. U dla U df t in CAP-MIMO). We note that in conventional MIMO, blocks ii) and iv) are art of the overall DSP and in CAP-MIMO they reresent mm-wave analog devices. For a link with a dimensional communication subsace, both transceivers ma digital data streams x d (1)...x d () onto the orthogonal beams that san the communication subsace. A key observation is that conventional MIMO has an individual T/R chain associated with each of its n discrete antenna elements regardless of. On the other hand, CAP-MIMO has a T/R chain associated with each of the DLA feed antennas corresonding to the orthogonal beams that san the low dimensional communication subsace. This critical, yet subtle difference leads to a dramatic decrease in the system comlexity from the order of n ( ) in conventional MIMO to the order of ( 100) in CAP-MIMO. Fig. 3: Digital beamforming in a conventional MIMO transceiver Fig. 4: DLA-based analog beamforming in a CAP-MIMO transceiver Fig. : Aroximate caacity lots for CAP-MIMO, Dish, and widely saced conventional MIMO D. Transceiver Comlexity: Analog vs Digital Beamforming Fig. 3 shows a conventional MIMO transceiver with a critically saced discrete antenna array and digital beamforming via U df t where each orthogonal beam is associated with a distinct inut to U df t. Alternatively, Fig. 4 shows a CAP- MIMO transceiver using a continuous DLA antenna for analog beamforming where each orthogonal beam is associated with a distinct feed antenna on the DLA focal surface. Both systems have the same signal sace dimension n and consist of four main functional blocks: i) DSP (digital signal rocessor), ii) beam selector, iii) transceiver hardware consisting of analog-to-digital/digital-to-analog (A-D/D-A) converters and transceiver (T/R) modules, and iv) beamformer (reresented by the matrix U df t in conventional MIMO and Fig. 4 also illustrates the key elements of the DLA. The DLA is modeled as U dla = PU fa, where P reresents the aerture hase rofile, and U fa reresents the roagation from the focal antennas to the (critically-samled) DLA aerture elements. The DLA acts like a convex lens and mas the signals in different directions to different locations on the focal surface. Conventional DLA designs use arrays of receiving and transmitting antennas connected with variable-length transmission line, e.g [16] [18] or extensions of this concet [19] [], to create the aerture hase rofile. The high-resolution, low-loss DLA used in CAP-MIMO is comosed of subwavelength, non-resonant hase shifting elements, or ixels, that can be distributed on a lanar surface, and act as bandass filters [3]. The resonse of each ixel is tuned to achieve a desired aerture hase rofile. The DLA was chosen over alternative analog beamformers, e.g. Butler matrices [4] [6], because of its low loss and relative ease of construction for antennas with a high dimensional satial signal sace.

5 5 In the following sections, we discuss the comutational modeling of the DLA-based CAP-MIMO architecture, erformance analysis, and measurements erformed on a rototye system to validate and refine the romising results given by the basic CAP-MIMO theory. III. COMPUTATIONAL MODELING In this section, we build on the idea of critical samling to outline a comutational modeling framework for a LoS CAP-MIMO system. Sec. III-A develos a model for the aerture domain channel based on the hysical arameters of the link. In Sec. III-B, we develo a critically samled model for the DLA. In Sec. III-C we combine the channel and DLA models to develo a beamsace model for the DLAbased CAP-MIMO system. Finally, in Sec. III-D we resent some illustrative alications of the comutational modeling framework to the rototye system. A. Aerture Domain Channel Modeling Consider a LoS link consisting of a rectangular transmitter antenna with area A T = L T,x L T,y and dimension n T = n T,x n T,y, and a rectangular receiver antenna with area A R = L R,x L R,y and dimension n R = n R,x n R,y, searated by a link length of R. The transmit antenna is oriented in the x-y lane with its center at the origin, and the receive antenna is arallel to the transmitter with its center located at a distance R, as shown in Fig. 5. Let T =(x T,y T,z T = 0) and R =(x R,y R,z R = R) denote the coordinates of oints on the transmitter and receiver aertures, resectively. Modeling where m =(m x,m y ) I(n T,x ) I(n T,y ), ` =(`x,`y) I(n R,x ) I(n R,y ), and the index set I(n) is defined in (5). Using the above transmit and receive aerture samle oints in (13) constructs the aerture domain all-hase channel matrix H that comletely characterizes the LoS link. As in (8), each column of H reresents the hase relationshi between the signal at a transmit samle oint to all the receive samle oints. However, (13) accounts for the curvature of the hase front at the receiver unlike the lane wave aroximation in (8). Strictly seaking, (13) does not corresond to the Kronecker roduct of two 1D channels. However, as we will discuss in Sec. IV, it does exhibit a nearly Kronecker structure. B. DLA Modeling We now construct a hysically accurate model for the DLA matrix, U dla. We consider a DLA in the transmit mode: U dla reresents the maing from the focal surface antennas to the critically samled oints on its aerture. In the receive mode, the maing from the aerture to the focal antennas is given by U T dla. As in Sec. II, the DLA is modeled as U dla = PU fa, where P models the aerture hase rofile, and U fa models the roagation from the focal surface antennas to the aerture. Based on the CAP-MIMO theory, an ideal DLA affects a satial Fourier transform: U dla = U df t. In reality, this relationshi is aroximate. Both U fa and P rovide design degrees of freedom for imroving this aroximation. Fig. 5: The LoS link geometry the aerture oints as idealized isotroic antennas and ignoring ath loss, the signal at the receiver aerture location, R, is related to the signal at the transmitter aerture location, T, via a ure hase shift, e j D c( R, T ), where D c ( R, T )= (xt x R ) +(y T y R ) + R is the distance between the two oints. The elements of the n R n T aerture domain channel matrix H consist of all such airwise channel hase shifts corresonding to the /-saced x-y samle oints: H(`, m) =e j D c( R (`), T (m)). (13) The n T and n R critically samled transmitter and receiver coordinates are given by mx T (m) = T (m x,m y )=, m y, 0, (14) `x R(`) = R (`x,`y) =, `y,r, (15) Fig. 6: DLA geometry with feed angles x and y Starting with U fa, consider a rectangular DLA with aerture size A = L x L y, dimension n = n x n y, and focal length F, as shown in Fig. 6. The DLA aerture is oriented in the x-y lane with its center at the origin and with the feed antennas located on a focal surface at a radial distance F from the origin. Let a =(x a,y a,z a = 0) and f =(x f,y f,z f ) denote the coordinates of arbitrary oints on the aerture and the focal surface. Using Fig. 6, for a given F, we arameterize the focal surface coordinates in terms of the angles x and y : z f = F tan( x ) + tan( y ) +1 x f = z f tan( x ), y f = z f tan( y ). (16) Modeling the aerture and focal surface oints as idealized isotroic antennas, and ignoring ath loss, the signal at the aerture location, a, is related to the signal at the focal surface location, f, via a ure hase shift: e j D fa ( a, f )

6 6 q where D fa ( a, f )= (x f x a ) +(y f y a ) + zf is the distance between the aerture and focal surface oints. The elements of the n n matrix U fa consist of all such airwise hase shifts corresonding to the critically saced aerture and focal surface oints: U fa (`, m) = 1 n e j D fa ( a(`), f (m)) (17) where the 1/ n term is used for ower normalization. The n critically samled aerture coordinates are given by `x a(`) = a (`x,`y) =, `y, 0, (18) where ` =(`x,`y) I(n x ) I(n y ). The critically samled focal surface coordinates, f (m) = f (m x,m y ), are determined by choosing the angles x(m x ) and y (m y ) to be the 1D orthogonal satial angles defined in (6) and (4): x(m x )=sin 1 ( x (m x )), x (m x )= m x n x y(m y )=sin 1 ( y (m y )), y (m y )= m y n y, (19) where m x I(n x ) and m y I(n y ). We note that the focal surface coordinates in (16), arameterized by the angles in (19), determine the location of the DLA feed antennas. Combining these coordinates with the aerture coordinates given by (18) in (17) yields the matrix U fa. Each column of U fa reresents the hase relation between a articular feed signal and all the aerture samle oints, and the ower normalization ensures ower conservation between the feed antenna and the DLA aerture. For a given U fa, the aerture hase rofile matrix P is designed so that U dla best aroximates U df t. This is, in general, a comlex roblem and relates to the significant work on microwave lenses over several decades [16] []. For our initial rototye, we work with the simlest broadside DLA design in which a lane wave coming from the broadside direction is erfectly focused on the broadside feed location or vice versa. This requires that the hase shift from the broadside feed location to any oint on the DLA aerture is constant. At an arbitrary aerture location, a =(x a,y a,z a = 0), this hase shift, ( a ), is given by ( a )= max ( a ), ( a )= x a + y a + F max = r L x 4 + L y 4 + F. (0) Assuming that the different samle oints or DLA ixels do not interact, which is the assumtion in our DLA design [3], P is a diagonal matrix. Assuming that the DLA aerture is lossless, the diagonal elements of P are given by the hase shift at the critically saced aerture samle oints: P (`, `) =e j ( a(`)),`=(`x,`y) I(n x ) I(n y ), (1) where a (`) is defined in (18). We have outlined the constructions for U fa and P in terms of critically saced aerture and focal surface samles, which is the minimum samling resolution required for accurate system modeling, consistent with the dimension of the satial signal sace. While finer (higher resolution) samling may be emloyed for visualization uroses, critically samling is sufficient for the system analysis in Sec. IV. C. Beamsace System Modeling We are now in a osition to model the comlete DLA-based CAP-MIMO system in beamsace. Consider a LoS link of length R connected by lanar antennas of dimensions n T and n R. Using the aerture domain channel matrix H in (13) the aerture domain system is described by (). The beamsace channel model is given by (3), characterized by the n R n T beamsace channel matrix H b, where U b,t and U b,r denote the transmit and receive beamforming matrices. For an ideal CAP-MIMO system, the beamforming matrices are given by the DFT matrices, defined in (7). For an actual DLA-based CAP-MIMO system, the beamforming matrices are given by U dla = PU fa using the constructions in (17) and (1). As discussed in Sec. II, for a LoS link = los n T,n R beams, defined in (10), strongly coule from the transmitter to the receiver and reresent the communication modes of the link. In a CAP-MIMO system, these modes are accessed via beams that are in turn accessed via a subset of the focal surface feed antennas in a DLA-based system. Thus, in subsequent sections, we exlicitly consider such lower dimensional system reresentations corresonding to a subset of n b DLA feed antennas or beams. Let Ũ b,t and Ũb,R reresent the corresonding n T n b and n R n b beamforming sub-matrices obtained by retaining the columns corresonding to the selected beams. The corresonding n b n b beamsace channel matrix is given by H b = ŨT b,r HŨb,T. In articular, for the DLA-based system H b = ŨT dla,rhũdla,t = ŨT fa,rp T HPŨfa,T () where Ũfa,T (n T n b ) and Ũfa,R (n R n b ) denote the transmit and receive DLA roagation sub-matrices corresonding to the selected n b feed antennas. D. The Prototye System In this section, we aly the basic CAP-MIMO theory and the modeling framework to an actual rototye system that we have built as an initial test latform. The rototye consists of two 40cm 40cm square DLAs, designed for broadside focusing with a focal length of F = 40cm using the rocedure in [3], searated by a link length of R =.67m (8.75 ft), and oerating at f c = 10GHz. While this falls outside of the mm-wave regime fc [30, 300]GHz, the rototye is intended to show satial multilexing in a LoS link, as redicted by the CAP-MIMO theory. The results scale to mm-wave where the smaller wavelengths will result in larger link lengths for a given A and (see Sec. II-C). These secifications were chosen based on the available measurement equiment, and result in system arameters of n = 676 and los =4(or 1D arameters of n x = n y = 6 and los,1d =). To assess how well the broadside DLA design aroximates the ideal DFT oeration, we analyze the matrix U H df t U dla = U H df t PU fa. The (`, m) th entry of U H df t U dla contains the inner roduct between the `th and m th columns of U df t and U dla, reresenting the corresonding beams. Physically, the

7 7 Y(cm) Z(cm) (a) 0 X(cm) 10 Y (cm) X (cm) (b) Fig. 8: Tracking beam roagation through the system: (a) shows a subsection of the TX focal surface with the activated feed indicated by the filled red square, (b) shows the resulting aerture domain receive signal radiation intensity attern, and (c) shows the RX focal surface radiation intensity attern with the active feed shown as a filled blue square. In (a) and (c) broadside is indicated by a black diamond. db (c) Fig. 7: Contour lot of U H df t U dla for assessing how well U dla aroximates the ideal U df t (`, m) th entry reresents the couling from the (isotroic) m th DLA feed antenna to an isotroic sensor laced at the `th orthogonal beam direction in the far-field. For a erfectly designed DLA, we exect the matrix to be diagonal - the identity matrix; that is, ideally, the m th feed antenna should only coule with the corresonding m th orthogonal beam. Fig. 7 shows a contour lot of U H df t U dla for the 1D case (n 1D = 6) consisting of linear antennas. As evident, the broadside DLA closely aroximates the DFT for hysical angles between ±30 degrees (which contains 14 orthogonal beams) and exhibits some off-diagonal entries beyond that range. Thus, the broadside DLA design seems quite adequate for broadside links with los u to 10. We will revisit this observation in Sec. V. Next we use the modeling framework to track the roagation of one of the los beams that san the communication subsace through the system to gain a qualitative understanding of CAP-MIMO s oeration. As shown in Fig. 8, the roagation roceeds as follows: i) Fig. 8a shows a sub-section of the transmit DLA focal surface with the los =4 selected feed locations (corresonding to the orthogonal beams that san the communication subsace) shown as red squares with the activated feed filled in, ii) Fig. 8b shows the receive aerture radiation intensity, given by Hu dla where u dla is the column of U dla corresonding to the activated feed, iii) Fig. 8c shows the receive focal surface radiation intensity with the selected feeds shown as blue squares and the active feed filled in. As evident the received aerture signal is concentrated in one quadrant of the aerture and the received beamsace signal is concentrated on the intended feed with most of its ower concentrated on the selected los feeds. The signals excited by the other selected feeds will be rotations of this signal, demonstrating how CAP-MIMO acks los =4indeendent beams into the receiver aerture (see also Fig. 1). While the above discussion alies to idealized isotroic antennas, the modeling framework can be extended to account for the actual feeds used for measurements. Essentially, we need to modify the columns of U fa which reresent the relationshi between the signal at a articular feed antenna and the electric field at the critical aerture samle oints. Since the focal length F is sufficiently large comared to the size of the feed antennas, whose dimensions are on the order of, we can use well-known aroximations for the far-field atterns of the chosen feed antennas [13]. The columns of U fa can be modified by rojecting the electric field of the actual feed antennas onto the DLA aerture coordinates, accounting for different feed locations relative to the DLA aerture. The modified columns of U fa should be normalized to unit energy. This will be further discussed in the Aendix. IV. ANALYTICAL RESULTS In this section, we use the comutational modeling framework to the analyze the erformance of CAP-MIMO in a PP LoS link. In Sec. IV-A, we discuss the LoS link caacity and how it is achieved by CAP-MIMO via beamsace communication. Then in Sec. IV-B, we analyze the caacity of the rototye DLA-based CAP-MIMO system. A. Link Caacity and Otimal Beamsace Signaling The LoS link caacity is governed by the eigenvectors and eigenvalues of the n R n T samled aerture channel matrix H. Let n o =min(n R,n T ) and c =tr(h H H)=n T n R denote the channel ower. It is well-known [3] that caacity-

8 8 achieving signaling is governed by the singular value decomosition (SVD) of H, H = U c 1/ c Vc H, where U c (n R n o ) is the matrix of eigenvectors of the receive covariance matrix HH H = U c c U H c, V c (n T n o ) is the matrix of eigenvectors of the transmit covariance matrix H H H = V c c Vc H, and c = diag( 1,, n o ) is the matrix of non-negative eigenvalues. In the resence of comlex additive white Gaussian noise (AWGN), caacity-achieving signaling consists of indeendent Gaussian signals over the transmit eigenvectors. That is, the otimal n T dimensional transmit signal vector x is comlex Gaussian, x CN(0, V c s Vc H ), with s = diag( 1,, no ) denoting the matrix of eigenvalues of the signal covariance matrix, and =tr( s )= P n o i=1 i denoting the total transmit SNR (signal to noise ratio), since the noise variance is taken to be unity without loss of generality. The link caacity is given by the well-known waterfilling owerallocation formula [3], [7] C( ) = max P log i: no [det(i + c s )] bits/s/hz (3) i=1 iale = max i: P no Xn o i=1 iale i=1 los max i: P los i=1 iale i=1 los log 1+ n T n R los log (1 + i i ) (4) X log (1 + i i ) (5) (6) where the aroximation in (5) is based on the fact that H has los dominant non-zero eigenvalues, and the second aroximation, used in (6), is based on the assumtion that non-zero eigenvalues are equal ( i n T n R / los ) with equal ower ( i = / los ) allocated to them. In general, los is a conservative aroximation of the number of nonzero eigenvalues, and the eigenvalues are not exactly uniform. At low SNRs, all the ower is allocated to the dominant mode corresonding to the largest eigenvalue and additional dominant modes are activated as the SNR increases, with equal ower allocated to all the dominant non-zero modes at sufficiently high SNR. The CAP-MIMO transceiver architecture is based on a key observation that the Fourier basis functions serve as aroximate channel eigenfunctions [9], [10] which are in turn aroximated by the DLA; that is, U c U b,r U dla,r, V c U b,t U dla,t (7) and the beamsace channel matrix H b is an aroximation to the diagonal matrix of channel singular values ( 1/ c ), as illustrated in Fig. 1. Thus, beamsace communication, affected through DLA-based analog beamforming in CAP- MIMO, rovides near-otimal access to the los -dimensional communication subsace of the high-dimensional n R n T channel H. In articular, a los los sub-matrix of H b, Hb, characterizes this low-dimensional communication subsace that is accessed with transceiver comlexity on the order of as shown in Fig. 4. This corresonds to the DSP block modulating each of the data streams onto a singular vector of H b. In subsequent sections, we will evaluate the caacities of various systems by using the aroriate set of eigenvalues in (4). We note that caacity achieving signaling requires knowledge of the channel at both the transmitter and receiver. Since we exect the deterministic LoS channel to vary slowly, there will be sufficient time to obtain estimates of the channel at both the transmitter and receiver. B. Prototye Caacity Analysis In this section, we theoretically assess the eigenvalues and the caacity of the rototye LoS link described in Sec. III-D (n = 676 and los =4), and the ability of CAP-MIMO to aroach caacity with the designed DLA. Fig. 9 shows the 10 largest eigenvalues of H H H and H H b H b, and 4 eigenvalues of H H H b b, normalized with resect to the maximum eigenvalue of H H H, to comare the number of modes suorted by the aerture channel, full-dimensional beamsace channel, and the low-dimensional beamsace channel. All channels exhibit los 4 dominant modes as theoretically exected. Furthermore, the distribution of eigenvalues is very similar to what would be exected from a Kronecker matrix. This confirms that while the channel model (13) is not strictly Kronecker, it behaves very similarly to the insightful array steering vector model (8) used in develoing the basic CAP- MIMO theory. We note that the eigenvalues of H b are a little higher than the eigenvalues of H as a consequence of the broadside DLA design. 1 Fig. 9: Theoretical eigenvalues of H H H, H H b H b, and H H H b b normalized to the maximum eigenvalue of H H H Using the theoretical eigenvalues, Fig. 10 comares the caacity of the los los CAP-MIMO system with the two state-of-the-art designs a single-mode Dish system and the conventional los los MIMO system with widely saced antennas tailored to the rototye secifications. The conventional MIMO antennas have a gain of G MIMO = 10dB. The CAP-MIMO gain between broadside feed locations is used as a roxy for the Dish gain. The caacity (bits/s/hz) is lotted as a function of the transmit SNR - ratio of total transmit ower to the unit noise variance at the receiver for all systems. (The absolute SNR values are not critical here - we calibrate them to their hysical values in the next section.) In the conventional systems, Dish dominates at low SNRs whereas widely saced MIMO dominates at high SNRs. CAP-MIMO 1 This relative increase in beamsace eigenvalues in the broadside LoS link is offset by a reduction in beamsace channel ower in off-broadside directions to ensure total ower conservation; see Sec. V-B.

9 9 outerforms the state-of-the-art over the entire SNR range, reflecting the multilexing gain over Dish and ower gain over conventional MIMO. The results show that at a sectral efficiency of 10 bits/s/hz, or a data rate of 10 Gbs with 1GHz system bandwidth, CAP-MIMO has a 17dB SNR advantage over the state-of-the-art. Fig. 10 also shows that CAP-MIMO closely aroximates the caacity of the 4-beam ideal DFT channel. The caacity of a CAP-MIMO system with = 1 feeds is also shown which more closely aroximates the caacity of the full-dimensional (676x676) aerture domain channel H. This underscores two imortant observations. First, los, as calculated in (10), is an aroximate indicator larger number of modes can be exloited with higher transceiver comlexity. Second, by increasing the number of feeds to 1, CAP-MIMO can aroach the full caacity of the LoS link with a dramatically lower comlexity (on the order of 1) comared to the order n = 676 comlexity of a criticallysaced (676x676) conventional MIMO system. of CAP-MIMO relevant to oint-to-multioint (PMP) links. Finally, Sec. V-C resents system bandwidth measurements. Fig. 11: CAP-MIMO rototye measurement setu Fig. 10: Caacity comarison between CAP-MIMO and several alternative systems for the rototye LoS link V. M EASUREMENT R ESULTS In this section we discuss the measurement results obtained using the rototye LoS CAP-MIMO system (n = 676, = los = 4) secified in Sec. III-D. The rototye DLAs were constructed using the rocedure outlined in [3]. Each DLA consists of 8 dielectric layers and 9 (4 inductive and 5 caacitive) metallic layers with hase shifting ixels. The measurement setu consists of two structures that suort the DLAs as shown in Fig. 11. The feed antennas are held by an arm shown in Fig. 1 and moved to the aroriate ositions to measure the elements of the los los b using a vector network anabeamsace channel submatrix H lyzer (VNA). Two tyes of DLA feed antennas were used for measurements: vertically oriented / diole (DP) antennas, and oen-ended WR-90 waveguide (WG) antennas suorting a vertically oriented T E10 mode. Sec. V-A discusses the broadside PP measurements in which the DLAs are facing each other. We comare the measurement results with the theoretical results obtained in Sec. IV to assess the accuracy of the modeling framework. Sec. V-B discusses reliminary measurement results to assess the off-broadside erformance Fig. 1: CAP-MIMO measurement structure showing the feed arm used to erform the measurements. A. Point-to-Point Measurements In this section, we discuss PP measurements taken using ˆ both the DP and WG feeds at 10GHz. Let H bm denote the bm denote measured beamsace channel submatrix and H its theoretical rediction. We want to relate the eigenvalues b to the the corresonding eigenvalues of H bm. To address of H this we consider two key issues: i) calibration of the channel ower/eigenvalues of the samled model in an idealized lossfree system to their actual hysical values, and ii) analysis of the dominant sources of ower loss in the rototye. Define the following (transmit) covariance matrices with their corresonding eigenvalue decomositions and owers = HH H = U UH, c = tr( ) b = H HH b = U b bu H, = tr( b) b b b H H bm = H H bm bm = Ubm bm Ubm, bm = tr( bm ). While the losses can vary across modes (orthogonal beams that serve as aroximate channel eigenfunctions), we exect the variation to be small for the LoS link. Thus we study the relationshi between the average channel eigenvalues: Pn P b c i=1 i i=1 b,i = = = = ave b,ave P bm,ave = bm = i=1 bm,i. To address the first issue, the lossless channel ower is calibrated to c = 3.78 using the analysis in the Aendix. This value reflects the hysically meaningful calibration (rather than

10 10 n in the system model) that accounts for free-sace ath loss. To address the second issue, we use the following relationshis: bm,ave = L T b,ave = L P L D,T L D,R b,ave = L P L D,T L D,R L S ave (8) where L T = L P L D,T L D,R denotes the total ower loss in the system, b,ave = L S ave, and the losses have the following interretation (er mode): 1) L P reresents the fraction of ower radiated by the transmitter that coules to the receiver. ) L D reresents the fraction of ower radiated by a DLA feed that coules to the DLA aerture, and vice versa. 3) L S is the ower loss due to limiting our attention to a -dimensional subsace in beamsace. Here, L D,T and L D,R refer to the loss at the transmitter and receiver resectively. In broadside PP links L D,T = L D,R = L D. From the analysis in the Aendix, we get the following values for the different quantities in (8): ave =0.94, L S =0.58, L P,WG =0.61, L P,DP =0.61 L D,WG =0., L D,DP =0.09 where the subscrit WG refers to the WR-90 waveguide feed and DP refers to the diole feed. Using the above values, we obtain the following theoretical redictions for average beamsace channel eigenvalues b,ave =.64dB, bm,ave,w G = 17.7dB, bm,ave,dp = 6.18dB (9) which are used for comarison with the measurement results. Fig. 13 lots the eigenvalues of the measured and theoretical channels (normalized with resect to each channel s maximum eigenvalue). Both the DP and WG channels exhibit =4dominant eigenvalues, as redicted by theory. The un-normalized average eigenvalues for the two channels are ˆ bm,ave,w G = 17.67dB, ˆ bm,ave,dp = 7.77dB (30) which are quite close to their theoretically redicted values in (9). We note from Fig. 13 that the DP channel better aroximates the theoretical distribution of eigenvalues, likely due to the fact that its radiation attern better aroximates the ideal uniform attern. On the other hand, from (30) we note that the WG feeds result in significantly better ower couling ( 10dB) to the DLA aerture, comared to the DP feeds. Next we assess the caacity of the measured channels relative to the theoretical redictions. The channel owers for the theoretical channels are normalized to the following values: 1) Lossless: b = L S c = =.18 ) WG: bm,w G = L D,WG L P,WG b = ) DP: bm,dp = L D,DP L P,DP b = Fig. 14 comares the caacity of the measured DP and WG channels relative to the three theoretical channels given above. Thus, the SNR values in Fig. 14 more accurately reflect the oerational transmit SNR. The results show an excellent agreement between the theoretical and measurement erformance of WG and DP channels. The SNR ga relative Fig. 13: Normalized eigenvalues of Hb, ˆ H bm,dp ˆ H bm,w G, and to the theoretical lossless erformance reflects the ower loss incurred by the two kinds of feeds (about 15dB for WG feeds). With WG feeds, the rototye system can deliver a sectral efficiency of 10 bits/s/hz at an oerational SNR 3dB, corresonding to a data rate of 10 Gbs with 1GHz system bandwidth. Fig. 14: Caacity versus SNR lots for measured and theoretical values of DP and WG channels. The caacity of the lossless theoretical channel is also included for reference. B. Point-to-Multi-Point Measurements A CAP-MIMO transmitter can simultaneously communicate with multile receivers in a oint-to-multioint (PMP) network link by aroriately selecting the location and number of feed antennas. Thus, assessment of CAP-MIMO s erformance in off-broadside directions is imortant. Initial off-broadside measurements with WG feeds were made by rotating one of the DLA structures, relative to the broadside direction, as shown in Fig. 15. Channel measurements were made at off-broadside angles u to = 60 degrees at 10 GHz to assess PMP erformance over a 10 degree sector. These measurements corresond to a set of PP links with the transmitter and receiver oriented at different angles. Fig. 16 lots single-feed off-broadside ower measurements and the corresonding theoretical redictions. The measurements were taken by adjusting the feeds so that they were both located along the imaginary line connecting the centers of the DLAs shown in Fig. 15. The results show a fairly good

11 11 Fig. 15: To view of the setu used for off-broadside measurements relevant to PMP oeration agreement between the measured and theoretical values. Both lots are normalized to the value at =0. The theoretical value for each angle is 0 log( H b ( ) ) + 10 log(l D,R ( )) + 10 log(l P ( )) where H b ( ) is the scalar beamsace channel between the feeds shown in Fig. 15. L P ( ) and L D,R ( ) are the same as in (8) excet that the oblique angle of the receiver is accounted for as discussed in the Aendix. The reflection coefficient was also measured at each angle and varied between aroximately 8dB and 10dB. Polarization urity measurements were taken at = 0, 30, 60 degrees by rotating the feed elements so that they would excite/receive horizontally or vertically olarized waves. At each angle the ower of the cross-olarized comonent was at least 19.67dB below the ower of the co-olarized comonent. NORMALIZED POWER (db) THEORETICAL MEASURED ψ Fig. 16: Single WG feed off-broadside ower measurements Next, beamsace channel measurements were erformed at =0, 30, 45, 60 degrees, where the two feed locations were in the same horizontal osition as in the ower measurements but their vertical ositions were chosen to corresond to orthogonal beam directions (searated by o). Table I shows the normalized measured and theoretical eigenvalues of each channel. In addition to showing good agreement, the results show that even at an angle of 60 degrees, the channel ossesses dominant eigenvalues. Using the unnormalized eigenvalues, Fig. 17 comares the caacity of the channels at different off-broadside angles. While there is some SNR loss in off-broadside directions, a limitation of the current broadside DLA design, the results indicate robust CAP-MIMO PMP oeration over a 10 degree sector. Eigenvalue 30 degrees 45 degrees 60 degrees 1st Measured/Theoretical 1.00/ / /1.00 nd Measured/Theoretical 0.40/ / /0.11 TABLE I: Normalized measured and theoretical eigenvalues for the channel at different off-broadside angles Fig. 17: measured channel caacity lots for = 0, 30, 45, 60 C. Bandwidth Assessment To assess the oerational bandwidth of the rototye DLAbased system, bandwidth measurements were made from 8-1GHz using single WG feeds on each end at different angles: = 0, 30, 45, 60 degrees. Fig. 18 lots the measurement results with the ower normalized to the maximum of the measurements for each angle. Broadside measurements are also shown for DP feeds. The results clearly indicate that the rototye system suorts a 3dB bandwidth of at least 1 GHz around f c = 10GHz (10% fractional bandwidth). Since we do not exect the frequency resonse to change areciably in the neighborhood of these feed locations, the results are a good indication of the multi-mode system bandwidth for small values of in both PP and PMP oerational modes. Fig. 18: Prototye frequency resonse over 8-1GHz for different angles VI. CONCLUSIONS Mm-wave systems offer a multitude of new oortunities and challenges in the design and analysis of multi-gigabit wireless communication systems. The advantages of relatively small aertures caable of simultaneously suorting high antenna gain and MIMO oeration are temered by the high transceiver comlexity associated with otimal exloitation of the satial dimension in conventional MIMO aroaches. In this aer, we have reorted romising results on modeling,

Investigation on Channel Estimation techniques for MIMO- OFDM System for QAM/QPSK Modulation

Investigation on Channel Estimation techniques for MIMO- OFDM System for QAM/QPSK Modulation International Journal Of Comutational Engineering Research (ijceronline.com) Vol. 2 Issue. Investigation on Channel Estimation techniques for MIMO- OFDM System for QAM/QPSK Modulation Rajbir Kaur 1, Charanjit

More information

Performance Analysis of MIMO System using Space Division Multiplexing Algorithms

Performance Analysis of MIMO System using Space Division Multiplexing Algorithms Performance Analysis of MIMO System using Sace Division Multilexing Algorithms Dr.C.Poongodi 1, Dr D Deea, M. Renuga Devi 3 and N Sasireka 3 1, Professor, Deartment of ECE 3 Assistant Professor, Deartment

More information

Evolutionary Circuit Design: Information Theory Perspective on Signal Propagation

Evolutionary Circuit Design: Information Theory Perspective on Signal Propagation Evolutionary Circuit Design: Theory Persective on Signal Proagation Denis Poel Deartment of Comuter Science, Baker University, P.O. 65, Baldwin City, KS 66006, E-mail: oel@ieee.org Nawar Hakeem Deartment

More information

Control of Grid Integrated Voltage Source Converters under Unbalanced Conditions

Control of Grid Integrated Voltage Source Converters under Unbalanced Conditions Jon Are Suul Control of Grid Integrated Voltage Source Converters under Unbalanced Conditions Develoment of an On-line Frequency-adative Virtual Flux-based Aroach Thesis for the degree of Philosohiae Doctor

More information

IEEE P Wireless Personal Area Networks. UWB Channel Model for under 1 GHz

IEEE P Wireless Personal Area Networks. UWB Channel Model for under 1 GHz Setember, 4 IEEE P85-4/55r Project Title Date Submitted Source Re: Abstract Purose Notice Release IEEE P85 Wireless Personal Area Networks IEEE P85 Working Grou for Wireless Personal Area Networks (WPANs)

More information

Decorrelation distance characterization of long term fading of CW MIMO channels in urban multicell environment

Decorrelation distance characterization of long term fading of CW MIMO channels in urban multicell environment Decorrelation distance characterization of long term fading of CW MIMO channels in urban multicell environment Alayon Glazunov, Andres; Wang, Ying; Zetterberg, Per Published in: 8th International Conference

More information

A Printed, Broadband Luneburg Lens Antenna

A Printed, Broadband Luneburg Lens Antenna IEEE TRANSACTIONS ON ANTENNAS AND PROPAGATION, VOL. 58, NO. 9, SEPTEMBER 010 3055 A Printed, Broadband Luneburg Lens Antenna Carl Pfeiffer and Anthony Grbic Abstract The design of a D broadband, Luneburg

More information

SPACE-FREQUENCY CODED OFDM FOR UNDERWATER ACOUSTIC COMMUNICATIONS

SPACE-FREQUENCY CODED OFDM FOR UNDERWATER ACOUSTIC COMMUNICATIONS SPACE-FREQUENCY CODED OFDM FOR UNDERWATER ACOUSTIC COMMUNICATIONS E. V. Zorita and M. Stojanovic MITSG 12-35 Sea Grant College Program Massachusetts Institute of Technology Cambridge, Massachusetts 02139

More information

High resolution radar signal detection based on feature analysis

High resolution radar signal detection based on feature analysis Available online www.jocr.com Journal of Chemical and Pharmaceutical Research, 4, 6(6):73-77 Research Article ISSN : 975-7384 CODEN(USA) : JCPRC5 High resolution radar signal detection based on feature

More information

EXPERIMENT 6 CLOSED-LOOP TEMPERATURE CONTROL OF AN ELECTRICAL HEATER

EXPERIMENT 6 CLOSED-LOOP TEMPERATURE CONTROL OF AN ELECTRICAL HEATER YEDITEPE UNIVERSITY ENGINEERING & ARCHITECTURE FACULTY INDUSTRIAL ELECTRONICS LABORATORY EE 432 INDUSTRIAL ELECTRONICS EXPERIMENT 6 CLOSED-LOOP TEMPERATURE CONTROL OF AN ELECTRICAL HEATER Introduction:

More information

Transmitter Antenna Diversity and Adaptive Signaling Using Long Range Prediction for Fast Fading DS/CDMA Mobile Radio Channels 1

Transmitter Antenna Diversity and Adaptive Signaling Using Long Range Prediction for Fast Fading DS/CDMA Mobile Radio Channels 1 Transmitter Antenna Diversity and Adative Signaling Using ong Range Prediction for Fast Fading DS/CDMA Mobile Radio Channels 1 Shengquan Hu, Tugay Eyceoz, Alexandra Duel-Hallen North Carolina State University

More information

Advancing Test in Coherent Transmission Systems. Daniel van der Weide

Advancing Test in Coherent Transmission Systems. Daniel van der Weide Advancing Test in Coherent Transmission Systems Daniel van der Weide 1 Otametra History Comlex Measurements Made Simle First 10 Systems Sold January 2011 Acquired by Tektronix -- July 2011 Founded Set

More information

Optimum use of a 4-element Yagi-Uda Antenna for the Reception of Several UHF TV Channels

Optimum use of a 4-element Yagi-Uda Antenna for the Reception of Several UHF TV Channels ENGNEE - Vol, No 3, [67-7], 7 The nstitution of Engineers, Sri anka htt://doiorg/438/engineerv5i3766 Otimum use of a 4-element Yagi-Uda Antenna for e ecetion of Several UHF TV Channels CJSAH Perera Abstract:

More information

FROM ANTENNA SPACINGS TO THEORETICAL CAPACITIES - GUIDELINES FOR SIMULATING MIMO SYSTEMS

FROM ANTENNA SPACINGS TO THEORETICAL CAPACITIES - GUIDELINES FOR SIMULATING MIMO SYSTEMS FROM ANTENNA SPACINGS TO THEORETICAL CAPACITIES - GUIDELINES FOR SIMULATING MIMO SYSTEMS Laurent Schumacher, Klaus I. Pedersen, Preben E. Mogensen Center for PersonKommunikation, Niels Jernes vej, DK-9

More information

Multi-Aperture Phased Arrays Versus Multi-beam Lens Arrays for Millimeter-Wave Multiuser MIMO

Multi-Aperture Phased Arrays Versus Multi-beam Lens Arrays for Millimeter-Wave Multiuser MIMO Multi-Aperture Phased Arrays Versus Multi-beam Lens Arrays for Millimeter-Wave Multiuser MIMO Asilomar 2017 October 31, 2017 Akbar M. Sayeed Wireless Communications and Sensing Laboratory Electrical and

More information

RECOMMENDATION ITU-R SF

RECOMMENDATION ITU-R SF Rec. ITU-R SF.1649-1 1 RECOMMENDATION ITU-R SF.1649-1 Guidance for determination of interference from earth stations on board vessels to stations in the fixed service when the earth station on board vessels

More information

Antenna Selection Scheme for Wireless Channels Utilizing Differential Space-Time Modulation

Antenna Selection Scheme for Wireless Channels Utilizing Differential Space-Time Modulation Antenna Selection Scheme for Wireless Channels Utilizing Differential Sace-Time Modulation Le Chung Tran and Tadeusz A. Wysocki School of Electrical, Comuter and Telecommunications Engineering Wollongong

More information

Depth of Focus and the Alternating Phase Shift Mask

Depth of Focus and the Alternating Phase Shift Mask T h e L i t h o g r a h y E x e r t (November 4) Deth of Focus and the Alternating Phase Shift Mask Chris A. Mack, KLA-Tencor, FINLE Division, Austin, Texas One of the biggest advantages of the use of

More information

The Multi-Focus Plenoptic Camera

The Multi-Focus Plenoptic Camera The Multi-Focus Plenotic Camera Todor Georgiev a and Andrew Lumsdaine b a Adobe Systems, San Jose, CA, USA; b Indiana University, Bloomington, IN, USA Abstract Text for Online or Printed Programs: The

More information

Random Access Compressed Sensing in Underwater Sensor Networks

Random Access Compressed Sensing in Underwater Sensor Networks Random Access Comressed Sensing in Underwater Sensor Networks Fatemeh Fazel Northeastern University Boston, MA 2115 Email: ffazel@ece.neu.edu Maryam Fazel University of Washington Seattle, WA 98195 Email:

More information

LAB IX. LOW FREQUENCY CHARACTERISTICS OF JFETS

LAB IX. LOW FREQUENCY CHARACTERISTICS OF JFETS LAB X. LOW FREQUENCY CHARACTERSTCS OF JFETS 1. OBJECTVE n this lab, you will study the -V characteristics and small-signal model of Junction Field Effect Transistors (JFET).. OVERVEW n this lab, we will

More information

Matching Book-Spine Images for Library Shelf-Reading Process Automation

Matching Book-Spine Images for Library Shelf-Reading Process Automation 4th IEEE Conference on Automation Science and Engineering Key Bridge Marriott, Washington DC, USA August 23-26, 2008 Matching Book-Sine Images for Library Shelf-Reading Process Automation D. J. Lee, Senior

More information

D-BLAST Lattice Codes for MIMO Block Rayleigh Fading Channels Λ

D-BLAST Lattice Codes for MIMO Block Rayleigh Fading Channels Λ D-BLAST Lattice Codes for MIMO Block Rayleigh Fading Channels Λ Narayan Prasad and Mahesh K. Varanasi e-mail: frasadn, varanasig@ds.colorado.edu University of Colorado, Boulder, CO 80309 October 1, 2002

More information

Circular Dynamic Stereo and Its Image Processing

Circular Dynamic Stereo and Its Image Processing Circular Dynamic Stereo and Its Image Processing Kikuhito KAWASUE *1 and Yuichiro Oya *2 *1 Deartment of Mechanical Systems Engineering Miyazaki University 1-1, Gakuen Kibanadai Nishi, Miyazaki 889-2192

More information

UNDERWATER ACOUSTIC CHANNEL ESTIMATION USING STRUCTURED SPARSITY

UNDERWATER ACOUSTIC CHANNEL ESTIMATION USING STRUCTURED SPARSITY UNDERWATER ACOUSTIC CHANNEL ESTIMATION USING STRUCTURED SPARSITY Ehsan Zamanizadeh a, João Gomes b, José Bioucas-Dias c, Ilkka Karasalo d a,b Institute for Systems and Robotics, Instituto Suerior Técnico,

More information

University of Twente

University of Twente University of Twente Faculty of Electrical Engineering, Mathematics & Comuter Science Design of an audio ower amlifier with a notch in the outut imedance Remco Twelkemeijer MSc. Thesis May 008 Suervisors:

More information

An Overview of PAPR Reduction Optimization Algorithm for MC-CDMA System

An Overview of PAPR Reduction Optimization Algorithm for MC-CDMA System RESEARCH ARTICLE OPEN ACCESS An Overview of PAPR Reduction Otimization Algorithm for MC-CDMA System Kanchan Singla*, Rajbir Kaur**, Gagandee Kaur*** *(Deartment of Electronics and Communication, Punjabi

More information

Arrival-Based Equalizer for Underwater Communication Systems

Arrival-Based Equalizer for Underwater Communication Systems 1 Arrival-Based Equalizer for Underwater Communication Systems Salman Ijaz, António Silva, Sérgio M. Jesus Laboratório de Robótica e Sistemas em Engenharia e Ciência (LARsys), Camus de Gambelas, Universidade

More information

Ground Clutter Canceling with a Regression Filter

Ground Clutter Canceling with a Regression Filter 1364 JOURNAL OF ATMOSPHERIC AND OCEANIC TECHNOLOGY VOLUME 16 Ground Clutter Canceling with a Regression Filter SEBASTIÁN M. TORRES Cooerative Institute for Mesoscale Meteorological Studies, Norman, Oklahoma

More information

A Multi-View Nonlinear Active Shape Model Using Kernel PCA

A Multi-View Nonlinear Active Shape Model Using Kernel PCA A Multi-View Nonlinear Active Shae Model Using Kernel PCA Sami Romdhani y, Shaogang Gong z and Alexandra Psarrou y y Harrow School of Comuter Science, University of Westminster, Harrow HA1 3TP, UK [rodhams

More information

Semi Blind Channel Estimation: An Efficient Channel Estimation scheme for MIMO- OFDM System

Semi Blind Channel Estimation: An Efficient Channel Estimation scheme for MIMO- OFDM System Australian Journal of Basic and Alied Sciences, 7(7): 53-538, 03 ISSN 99-878 Semi Blind Channel Estimation: An Efficient Channel Estimation scheme for MIMO- OFDM System Arathi. Devasia, Dr.G. Ramachandra

More information

Initial Ranging for WiMAX (802.16e) OFDMA

Initial Ranging for WiMAX (802.16e) OFDMA Initial Ranging for WiMAX (80.16e) OFDMA Hisham A. Mahmoud, Huseyin Arslan Mehmet Kemal Ozdemir Electrical Engineering Det., Univ. of South Florida Logus Broadband Wireless Solutions 40 E. Fowler Ave.,

More information

Ultra Wideband System Performance Studies in AWGN Channel with Intentional Interference

Ultra Wideband System Performance Studies in AWGN Channel with Intentional Interference Ultra Wideband System Performance Studies in AWGN Channel with Intentional Interference Matti Hämäläinen, Raffaello Tesi, Veikko Hovinen, Niina Laine, Jari Iinatti Centre for Wireless Communications, University

More information

The pulse compression waveform that we have already considered is the LFM t is a quadratic phase function.

The pulse compression waveform that we have already considered is the LFM t is a quadratic phase function. 5.0 PULSE COMPRESSION WAVEFORMS There is a class of waveforms termed ulse comression waveforms. These tyes of waveforms, and their associated signal rocessors, are useful because the overall signal duration

More information

Application of Notch Filtering under Low Sampling Rate for Broken Rotor Bar Detection with DTFT and AR based Spectrum Methods

Application of Notch Filtering under Low Sampling Rate for Broken Rotor Bar Detection with DTFT and AR based Spectrum Methods Alication of Notch Filtering under Low Samling Rate for Broken Rotor Bar Detection with DTFT and AR based Sectrum Methods B. Ayhan H. J. Trussell M.-Y. Chow M.-H. Song IEEE Student Member IEEE Fellow IEEE

More information

Photonic simultaneous frequency identification of radio-frequency signals with multiple tones

Photonic simultaneous frequency identification of radio-frequency signals with multiple tones Photonic simultaneous frequency identification of radio-frequency signals with multile tones Hossein Emami,, * Niusha Sarkhosh, and Mohsen Ashourian Deartment of Electrical Engineering, Majlesi Branch,

More information

TO IMPROVE BIT ERROR RATE OF TURBO CODED OFDM TRANSMISSION OVER NOISY CHANNEL

TO IMPROVE BIT ERROR RATE OF TURBO CODED OFDM TRANSMISSION OVER NOISY CHANNEL TO IMPROVE BIT ERROR RATE OF TURBO CODED TRANSMISSION OVER NOISY CHANNEL 1 M. K. GUPTA, 2 VISHWAS SHARMA. 1 Deartment of Electronic Instrumentation and Control Engineering, Jagannath Guta Institute of

More information

Kaleidoscope modes in large aperture Porro prism resonators

Kaleidoscope modes in large aperture Porro prism resonators Kaleidoscoe modes in large aerture Porro rism resonators Liesl Burger,2,* and Andrew Forbes,2 CSIR National Laser Centre, PO Box 395, Pretoria 000, South Africa 2 School of Physics, University of KwaZulu

More information

Uplink Scheduling in Wireless Networks with Successive Interference Cancellation

Uplink Scheduling in Wireless Networks with Successive Interference Cancellation 1 Ulink Scheduling in Wireless Networks with Successive Interference Cancellation Majid Ghaderi, Member, IEEE, and Mohsen Mollanoori, Student Member, IEEE, Abstract In this aer, we study the roblem of

More information

Efficient Importance Sampling for Monte Carlo Simulation of Multicast Networks

Efficient Importance Sampling for Monte Carlo Simulation of Multicast Networks Efficient Imortance Samling for Monte Carlo Simulation of Multicast Networks P. Lassila, J. Karvo and J. Virtamo Laboratory of Telecommunications Technology Helsinki University of Technology P.O.Box 3000,

More information

RICIAN FADING DISTRIBUTION FOR 40GHZ CHANNELS

RICIAN FADING DISTRIBUTION FOR 40GHZ CHANNELS Jan 006 RICIAN FADING DISTRIBUTION FOR 40GHZ CHANNELS.0 Background and Theory Amlitude fading in a general multiath environment may follow different distributions deending recisely on the area covered

More information

Lab 4: The transformer

Lab 4: The transformer ab 4: The transformer EEC 305 July 8 05 Read this lab before your lab eriod and answer the questions marked as relaboratory. You must show your re-laboratory answers to the TA rior to starting the lab.

More information

PROVIDING ANCILLARY SERVICES IN DISTRIBUTION NETWORKS WITH VANADIUM REDOX FLOW BATTERIES: ALPSTORE PROJECT

PROVIDING ANCILLARY SERVICES IN DISTRIBUTION NETWORKS WITH VANADIUM REDOX FLOW BATTERIES: ALPSTORE PROJECT PROVIDING ANCILLARY SERVICES IN DISTRIBTION NETWORKS WITH VANADIM REDOX FLOW BATTERIES: ALPSTORE PROJECT Leoold HERMAN Boštjan BLAŽIČ Igor PAČ Faculty of Electrical Engineering, Faculty of Electrical Engineering,

More information

State-of-the-Art Verification of the Hard Driven GTO Inverter Development for a 100 MVA Intertie

State-of-the-Art Verification of the Hard Driven GTO Inverter Development for a 100 MVA Intertie State-of-the-Art Verification of the Hard Driven GTO Inverter Develoment for a 100 MVA Intertie P. K. Steimer, H. Grüning, J. Werninger R&D Drives and Power Electronics ABB Industrie AG CH-5300 Turgi,

More information

Prediction Efficiency in Predictive p-csma/cd

Prediction Efficiency in Predictive p-csma/cd Prediction Efficiency in Predictive -CSMA/CD Mare Miśowicz AGH University of Science and Technology, Deartment of Electronics al. Miciewicza 30, 30-059 Kraów, Poland misow@agh.edu.l Abstract. Predictive

More information

An Overview of Substrate Noise Reduction Techniques

An Overview of Substrate Noise Reduction Techniques An Overview of Substrate Noise Reduction Techniques Shahab Ardalan, and Manoj Sachdev ardalan@ieee.org, msachdev@ece.uwaterloo.ca Deartment of Electrical and Comuter Engineering University of Waterloo

More information

Capacity Gain From Two-Transmitter and Two-Receiver Cooperation

Capacity Gain From Two-Transmitter and Two-Receiver Cooperation 3822 IEEE TRANSACTIONS ON INFORMATION THEORY, VOL. 53, NO. 10, OCTOBER 2007 Caacity Gain From Two-Transmitter and Two-Receiver Cooeration Chris T. K. Ng, Student Member, IEEE, Nihar Jindal, Member, IEEE,

More information

Statistical Evaluation of the Azimuth and Elevation Angles Seen at the Output of the Receiving Antenna

Statistical Evaluation of the Azimuth and Elevation Angles Seen at the Output of the Receiving Antenna IEEE TANSACTIONS ON ANTENNAS AND POPAGATION 1 Statistical Evaluation of the Azimuth and Elevation Angles Seen at the Outut of the eceiving Antenna Cezary Ziółkowski and an M. Kelner Abstract A method to

More information

JOINT COMPENSATION OF OFDM TRANSMITTER AND RECEIVER IQ IMBALANCE IN THE PRESENCE OF CARRIER FREQUENCY OFFSET

JOINT COMPENSATION OF OFDM TRANSMITTER AND RECEIVER IQ IMBALANCE IN THE PRESENCE OF CARRIER FREQUENCY OFFSET JOINT COMPENSATION OF OFDM TRANSMITTER AND RECEIVER IQ IMBALANCE IN THE PRESENCE OF CARRIER FREQUENCY OFFSET Deeaknath Tandur, and Marc Moonen ESAT/SCD-SISTA, KULeuven Kasteelark Arenberg 10, B-3001, Leuven-Heverlee,

More information

Performance Analysis of Battery Power Management Schemes in Wireless Mobile. Devices

Performance Analysis of Battery Power Management Schemes in Wireless Mobile. Devices Performance Analysis of Battery Power Management Schemes in Wireless Mobile Devices Balakrishna J Prabhu, A Chockalingam and Vinod Sharma Det of ECE, Indian Institute of Science, Bangalore, INDIA Abstract

More information

Physics 54. Lenses and Mirrors. And now for the sequence of events, in no particular order. Dan Rather

Physics 54. Lenses and Mirrors. And now for the sequence of events, in no particular order. Dan Rather Physics 54 Lenses and Mirrors And now or the seuence o events, in no articular order. Dan Rather Overview We will now study transmission o light energy in the ray aroximation, which assumes that the energy

More information

An Adaptive Narrowband Interference Excision Filter with Low Signal Loss for GPS Receivers

An Adaptive Narrowband Interference Excision Filter with Low Signal Loss for GPS Receivers ICCAS5 An Adative Narrowband Filter with Low Signal Loss for GPS s Mi-Young Shin*, Chansik Park +, Ho-Keun Lee #, Dae-Yearl Lee #, and Sang-Jeong Lee ** * Deartment of Electronics Engineering, Chungnam

More information

Indirect Channel Sensing for Cognitive Amplify-and-Forward Relay Networks

Indirect Channel Sensing for Cognitive Amplify-and-Forward Relay Networks Indirect Channel Sensing for Cognitive Amlify-and-Forward Relay Networs Yieng Liu and Qun Wan Abstract In cognitive radio networ the rimary channel information is beneficial. But it can not be obtained

More information

A Novel Image Component Transmission Approach to Improve Image Quality and Energy Efficiency in Wireless Sensor Networks

A Novel Image Component Transmission Approach to Improve Image Quality and Energy Efficiency in Wireless Sensor Networks Journal of Comuter Science 3 (5: 353-360, 2007 ISSN 1549-3636 2007 Science Publications A Novel Image Comonent Transmission Aroach to Imrove Image Quality and nergy fficiency in Wireless Sensor Networks

More information

Measurement of Field Complex Noise Using a Novel Acoustic Detection System

Measurement of Field Complex Noise Using a Novel Acoustic Detection System Southern Illinois University Carbondale OenSIUC Conference Proceedings Deartment of Electrical and Comuter Engineering Fall 04 Measurement of Field Comlex Noise Using a Novel Acoustic Detection System

More information

Joint Tx/Rx Energy-Efficient Scheduling in Multi-Radio Networks: A Divide-and-Conquer Approach

Joint Tx/Rx Energy-Efficient Scheduling in Multi-Radio Networks: A Divide-and-Conquer Approach Joint Tx/Rx Energy-Efficient Scheduling in Multi-Radio Networs: A Divide-and-Conquer Aroach Qingqing Wu, Meixia Tao, and Wen Chen Deartment of Electronic Engineering, Shanghai Jiao Tong University, Shanghai,

More information

Performance Analysis and PAPR Calculation of OFDM System Under Different Modulation schemes

Performance Analysis and PAPR Calculation of OFDM System Under Different Modulation schemes SSRG International Journal of Electronics andoncommunication 2017) - Secial 2nd ndinternational Conference Innovations and- (2'ICEIS Solutions -(2'ICEIS - 2016)Issue - Aril 2017 2 International Conference

More information

ELECTRICAL TECHNOLOGY EET 103/4

ELECTRICAL TECHNOLOGY EET 103/4 ELECTRICAL TECHNOLOGY EET 103/4 Define and analyze the rincile of transformer, its arameters and structure. Describe and analyze Ideal transformer, equivalent circuit, and hasor diagram Calculate and justify

More information

Multi-TOA Based Position Estimation for IR-UWB

Multi-TOA Based Position Estimation for IR-UWB Multi-TOA Based Position Estimation for IR-UWB Genís Floriach, Montse Nájar and Monica Navarro Deartment of Signal Theory and Communications Universitat Politècnica de Catalunya (UPC), Barcelona, Sain

More information

Dynamic Range Enhancement Algorithms for CMOS Sensors With Non-Destructive Readout

Dynamic Range Enhancement Algorithms for CMOS Sensors With Non-Destructive Readout IEEE International Worksho on Imaging Systems and Techniques IST 2008 Chania, Greece, Setember 10 12, 2008 Dynamic Range Enhancement Algorithms for CMOS Sensors With Non-Destructive Readout Anton Kachatkou,

More information

Rajbir Kaur 1, Charanjit Kaur 2

Rajbir Kaur 1, Charanjit Kaur 2 Rajbir Kaur, Charanjit Kaur / International Journal of Engineering Research and Alications (IJERA) ISS: -9 www.ijera.com Vol., Issue 5, Setember- October 1,.139-13 based Channel Estimation Meods for MIMO-OFDM

More information

ANALYSIS OF ROBUST MILTIUSER DETECTION TECHNIQUE FOR COMMUNICATION SYSTEM

ANALYSIS OF ROBUST MILTIUSER DETECTION TECHNIQUE FOR COMMUNICATION SYSTEM ANALYSIS OF ROBUST MILTIUSER DETECTION TECHNIQUE FOR COMMUNICATION SYSTEM Kaushal Patel 1 1 M.E Student, ECE Deartment, A D Patel Institute of Technology, V. V. Nagar, Gujarat, India ABSTRACT Today, in

More information

Adaptive Switching between Spatial Diversity and Multiplexing: a Cross-layer Approach

Adaptive Switching between Spatial Diversity and Multiplexing: a Cross-layer Approach Adative Switching between Satial Diversity and ultilexing: a Cross-layer Aroach José Lóez Vicario and Carles Antón-Haro Centre Tecnològic de Telecomunicacions de Catalunya (CTTC) c/ Gran Caità -4, 08034

More information

This is a repository copy of Wideband outdoor MIMO channel model derived from directional channel measurements at 2 GHz.

This is a repository copy of Wideband outdoor MIMO channel model derived from directional channel measurements at 2 GHz. his is a reository coy of Wideband outdoor MIMO channel model derived from directional channel measurements at GHz. White Rose Research Online URL for this aer: htt://erints.whiterose.ac.uk/479/ Version:

More information

An Efficient VLSI Architecture Parallel Prefix Counting With Domino Logic Λ

An Efficient VLSI Architecture Parallel Prefix Counting With Domino Logic Λ An Efficient VLSI Architecture Parallel Prefix Counting With Domino Logic Λ Rong Lin y Koji Nakano z Stehan Olariu x Albert Y. Zomaya Abstract We roose an efficient reconfigurable arallel refix counting

More information

arxiv: v1 [eess.sp] 10 Apr 2018

arxiv: v1 [eess.sp] 10 Apr 2018 Sensing Hidden Vehicles by Exloiting Multi-Path V2V Transmission Kaifeng Han, Seung-Woo Ko, Hyukjin Chae, Byoung-Hoon Kim, and Kaibin Huang Det. of EEE, The University of Hong Kong, Hong Kong LG Electronics,

More information

Underwater acoustic channel model and variations due to changes in node and buoy positions

Underwater acoustic channel model and variations due to changes in node and buoy positions Volume 24 htt://acousticalsociety.org/ 5th Pacific Rim Underwater Acoustics Conference Vladivostok, Russia 23-26 Setember 2015 Underwater acoustic channel model and variations due to changes in node and

More information

Application Note D. Dynamic Torque Measurement

Application Note D. Dynamic Torque Measurement Page 1 of 9 Alication Note 221101D Dynamic Torque Measurement Background Rotary ower sources and absorbers have discrete oles and/or istons and/or gear meshes, etc. As a result, they develo and absorb

More information

Performance comparison of power delay profile Estimation for MIMO OFDM

Performance comparison of power delay profile Estimation for MIMO OFDM IOSR Journal of Engineering (IOSRJEN) ISSN (e): 2250-3021, ISSN (): 2278-8719 Vol. 04, Issue 06 (June. 2014), V5 PP 48-53 www.iosrjen.org Performance comarison of ower delay rofile Estimation for MIMO

More information

Data-precoded algorithm for multiple-relayassisted

Data-precoded algorithm for multiple-relayassisted RESEARCH Oen Access Data-recoded algorithm for multile-relayassisted systems Sara Teodoro *, Adão Silva, João M Gil and Atílio Gameiro Abstract A data-recoded relay-assisted (RA scheme is roosed for a

More information

CHAPTER 8 MIMO. Xijun Wang

CHAPTER 8 MIMO. Xijun Wang CHAPTER 8 MIMO Xijun Wang WEEKLY READING 1. Goldsmith, Wireless Communications, Chapters 10 2. Tse, Fundamentals of Wireless Communication, Chapter 7-10 2 MIMO 3 BENEFITS OF MIMO n Array gain The increase

More information

Origins of Stator Current Spectra in DFIGs with Winding Faults and Excitation Asymmetries

Origins of Stator Current Spectra in DFIGs with Winding Faults and Excitation Asymmetries Origins of Stator Current Sectra in DFIGs with Wing Faults and Excitation Asymmetries S. Williamson * and S. Djurović * University of Surrey, Guildford, Surrey GU2 7XH, United Kingdom School of Electrical

More information

Series PID Pitch Controller of Large Wind Turbines Generator

Series PID Pitch Controller of Large Wind Turbines Generator SERBIAN JOURNAL OF ELECRICAL ENGINEERING Vol. 1, No., June 015, 183-196 UDC: 61.311.4:681.5 DOI: 10.98/SJEE150183M Series PID Pitch Controller of Large Wind urbines Generator Aleksandar D. Micić 1, Miroslav

More information

Simulation and Characterization of UWB system coexistence with traditional communication Systems

Simulation and Characterization of UWB system coexistence with traditional communication Systems Simulation and Characterization of UWB system coexistence with traditional communication Systems Guided research by Oliver Wamanga International University Bremen Under suervision of Prof. Dr. Herald Haas

More information

THE HELMHOLTZ RESONATOR TREE

THE HELMHOLTZ RESONATOR TREE THE HELMHOLTZ RESONATOR TREE Rafael C. D. Paiva and Vesa Välimäki Deartment of Signal Processing and Acoustics Aalto University, School of Electrical Engineering Esoo, Finland rafael.dias.de.aiva@aalto.fi

More information

Secondary Transceiver Design in the Presence of Frequency Offset between OFDM-based Primary and Secondary Systems

Secondary Transceiver Design in the Presence of Frequency Offset between OFDM-based Primary and Secondary Systems Secondary Transceiver Design in the Presence of Frequency Offset between OFDM-based Primary and Secondary Systems Zhikun Xu and Chenyang Yang School of Electronics and Information Engineering, Beihang

More information

Influence of Earth Conductivity and Permittivity Frequency Dependence in Electromagnetic Transient Phenomena

Influence of Earth Conductivity and Permittivity Frequency Dependence in Electromagnetic Transient Phenomena Influence of Earth Conductivity and Permittivity Frequency Deendence in Electromagnetic Transient Phenomena C. M. Portela M. C. Tavares J. Pissolato ortelac@ism.com.br cristina@sel.eesc.sc.us.br isso@dt.fee.unicam.br

More information

Beamformings for Spectrum Sharing in Cognitive Radio Networks

Beamformings for Spectrum Sharing in Cognitive Radio Networks Raungrong Suleesathira, Satit Puranachieeree Beamformings for Sectrum Sharing in Cognitive Radio Networs Raungrong Suleesathira * and Satit Puranachieeree Deartment of Electronic and Telecommunication

More information

A Genetic Algorithm Approach for Sensorless Speed Estimation by using Rotor Slot Harmonics

A Genetic Algorithm Approach for Sensorless Speed Estimation by using Rotor Slot Harmonics A Genetic Algorithm Aroach for Sensorless Seed Estimation by using Rotor Slot Harmonics Hayri Arabaci Abstract In this aer a sensorless seed estimation method with genetic algorithm for squirrel cage induction

More information

ABSTRACT. GUNCAVDI, SECIN. Transmitter Diversity and Multiuser Precoding for Rayleigh

ABSTRACT. GUNCAVDI, SECIN. Transmitter Diversity and Multiuser Precoding for Rayleigh ABSTRACT GUNCAVDI, SECIN Transmitter Diversity and Multiuser Precoding for Rayleigh Fading Code Division Multile Access Channels (Under the direction of Alexandra- Duel-Hallen) Transmitter diversity in

More information

Servo Mechanism Technique based Anti-Reset Windup PI Controller for Pressure Process Station

Servo Mechanism Technique based Anti-Reset Windup PI Controller for Pressure Process Station Indian Journal of Science and Technology, Vol 9(11), DOI: 10.17485/ijst/2016/v9i11/89298, March 2016 ISSN (Print) : 0974-6846 ISSN (Online) : 0974-5645 Servo Mechanism Technique based Anti-Reset Windu

More information

Resolution Enhancement Technologies

Resolution Enhancement Technologies Tutor4.doc; Version 2/9/3 T h e L i t h o g r a h y E x e r t (May 23) Resolution Enhancement Technologies Chris A. Mack, KLA-Tencor, FINLE Division, Austin, Texas Classically seaking, otical lithograhy

More information

LDPC-Coded MIMO Receiver Design Over Unknown Fading Channels

LDPC-Coded MIMO Receiver Design Over Unknown Fading Channels LDPC-Coded MIMO Receiver Design Over Unknown Fading Channels Jun Zheng and Bhaskar D. Rao University of California at San Diego Email: juzheng@ucsd.edu, brao@ece.ucsd.edu Abstract We consider an LDPC-coded

More information

Performance Analysis of LTE Downlink under Symbol Timing Offset

Performance Analysis of LTE Downlink under Symbol Timing Offset Performance Analysis of LTE Downlink under Symbol Timing Offset Qi Wang, Michal Šimko and Markus Ru Institute of Telecommunications, Vienna University of Technology Gusshausstrasse 25/389, A-1040 Vienna,

More information

Reconfigurable Hybrid Beamforming Architecture for Millimeter Wave Radio: A Tradeoff between MIMO Diversity and Beamforming Directivity

Reconfigurable Hybrid Beamforming Architecture for Millimeter Wave Radio: A Tradeoff between MIMO Diversity and Beamforming Directivity Reconfigurable Hybrid Beamforming Architecture for Millimeter Wave Radio: A Tradeoff between MIMO Diversity and Beamforming Directivity Hybrid beamforming (HBF), employing precoding/beamforming technologies

More information

Chapter 7: Passive Filters

Chapter 7: Passive Filters EETOMAGNETI OMPATIBIITY HANDBOOK 1 hater 7: Passive Filters 7.1 eeat the analytical analysis given in this chater for the low-ass filter for an filter in shunt with the load. The and for this filter are

More information

Characteristics of optical bandpass filters employing series-cascaded double-ring resonators q

Characteristics of optical bandpass filters employing series-cascaded double-ring resonators q Otics Communications 8 (003) 91 98 www.elsevier.com/locate/otcom Characteristics of otical bandass filters emloying series-cascaded double-ring resonators q Jianyi Yang a,b, *, Qingjun Zhou b, Feng Zhao

More information

Analysis of Mean Access Delay in Variable-Window CSMA

Analysis of Mean Access Delay in Variable-Window CSMA Sensors 007, 7, 3535-3559 sensors ISSN 44-80 007 by MDPI www.mdi.org/sensors Full Research Paer Analysis of Mean Access Delay in Variable-Window CSMA Marek Miśkowicz AGH University of Science and Technology,

More information

Optimization of an Evaluation Function of the 4-sided Dominoes Game Using a Genetic Algorithm

Optimization of an Evaluation Function of the 4-sided Dominoes Game Using a Genetic Algorithm o Otimization of an Evaluation Function of the 4-sided Dominoes Game Using a Genetic Algorithm Nirvana S. Antonio, Cícero F. F. Costa Filho, Marly G. F. Costa, Rafael Padilla Abstract In 4-sided dominoes,

More information

A Pricing-Based Cooperative Spectrum Sharing Stackelberg Game

A Pricing-Based Cooperative Spectrum Sharing Stackelberg Game A Pricing-Based Cooerative Sectrum Sharing Stackelberg Game Ramy E. Ali, Karim G. Seddik, Mohammed Nafie, and Fadel F. Digham? Wireless Intelligent Networks Center (WINC), Nile University, Smart Village,

More information

COMPARISON OF DIFFERENT CDGPS SOLUTIONS FOR ON-THE-FLY INTEGER AMBIGUITY RESOLUTION IN LONG BASELINE LEO FORMATIONS

COMPARISON OF DIFFERENT CDGPS SOLUTIONS FOR ON-THE-FLY INTEGER AMBIGUITY RESOLUTION IN LONG BASELINE LEO FORMATIONS COMPARISON OF DIFFERENT CDGPS SOLUTIONS FOR ON-THE-FLY INTEGER AMBIGUITY RESOLUTION IN LONG BASELINE LEO FORMATIONS Urbano Tancredi (1), Alfredo Renga (2), and Michele Grassi (3) (1) Deartment for Technologies,

More information

Connection of CSO and JCMT to SMA

Connection of CSO and JCMT to SMA SMA memo 136 Connecting the CSO and JCMT to the SMA 1 Introduction Martina C. Wiedner March 1999 Currently there are two submillimeter telescoes on Mauna Kea in Hawaii, the Caltech Submillimeter Observatory

More information

A New ISPWM Switching Technique for THD Reduction in Custom Power Devices

A New ISPWM Switching Technique for THD Reduction in Custom Power Devices A New ISPWM Switching Technique for THD Reduction in Custom Power Devices S. Esmaeili Jafarabadi, G. B. Gharehetian Deartment of Electrical Engineering, Amirkabir University of Technology, 15914 Tehran,

More information

Antennas and Propagation. Chapter 6b: Path Models Rayleigh, Rician Fading, MIMO

Antennas and Propagation. Chapter 6b: Path Models Rayleigh, Rician Fading, MIMO Antennas and Propagation b: Path Models Rayleigh, Rician Fading, MIMO Introduction From last lecture How do we model H p? Discrete path model (physical, plane waves) Random matrix models (forget H p and

More information

Cone of Silence: Adaptively Nulling Interferers in Wireless Networks

Cone of Silence: Adaptively Nulling Interferers in Wireless Networks UCL DEPARTMENT OF COMPUTER SCIENCE Research Note RN/1/ Cone of Silence: Adatively Nulling Interferers in Wireless Networks 3 th January 1 Georgios Nikolaidis Astrit Zhushi Kyle Jamieson Brad Kar Abstract

More information

A novel High Bandwidth Pulse-Width Modulated Inverter

A novel High Bandwidth Pulse-Width Modulated Inverter Proceedings of the 10th WSEAS International onference on IRUITS, Vouliagmeni, Athens, Greece, July 101, 006 (8085) A novel High Bandwidth PulseWidth Modulated Inverter J. HATZAKIS, M. VOGIATZAKI, H. RIGAKIS,

More information

Product Accumulate Codes on Fading Channels

Product Accumulate Codes on Fading Channels Product Accumulate Codes on Fading Channels Krishna R. Narayanan, Jing Li and Costas Georghiades Det of Electrical Engineering Texas A&M University, College Station, TX 77843 Abstract Product accumulate

More information

Self-Driven Phase Shifted Full Bridge Converter for Telecom Applications

Self-Driven Phase Shifted Full Bridge Converter for Telecom Applications Self-Driven Phase Shifted Full Bridge Converter for Telecom Alications SEVILAY CETIN Technology Faculty Pamukkale University 7 Kinikli Denizli TURKEY scetin@au.edu.tr Abstract: - For medium ower alications,

More information

Interference Management via Sliding-Window Superposition Coding

Interference Management via Sliding-Window Superposition Coding Globecom 24 Worksho - Emerging Technologies for 5G Wireless Cellular Networks Interference Management via Sliding-Window Suerosition Coding Hosung ark, Young-Han Kim, Lele Wang University of California,

More information

Postprocessed time-delay interferometry for LISA

Postprocessed time-delay interferometry for LISA PHYSICAL REVIEW D, VOLUME 70, 081101(R) Postrocessed time-delay interferometry for LISA D. A. Shaddock,* B. Ware, R. E. Sero, and M. Vallisneri Jet Proulsion Laboratory, California Institute of Technology,

More information