1.6 bit/s/hz orthogonally polarized CSRZ - DQPSK transmission of 8 40 Gbit/s over 320 km NDSF

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1 TuF1 1.6 bit/s/hz orthogonally polarized CSRZ - DQPSK transmission of 8 40 Gbit/s over 320 km NDSF Y. Zhu, K. Cordina, N. Jolley, R. Feced, H. Kee, R. Rickard and A. Hadjifotiou Nortel Networks, Harlow Laboratories, London Road, Harlow, Essex CM17 9NA, UK Tel: +44-(0) , Fax: +44-(0) , yanjun@nortelnetworks.com Abstract: We report experimental results of 1.6 bit/s/hz, 8 40 Gbit/s OP-CSRZ-DQPSK transmission over 320 km NDSF and demonstration of an 8-fold extension in 40 Gbit/s chromatic dispersion tolerance and a doubling of 40 Gbit/s PMD tolerance without any adjustable compensator Optical Society of America OCIS codes: (060.23) Fiber optics; (060.45) Optical communications 1. Introduction Recent advances in high speed long haul transmission have been accelerated by enabling technologies including enhanced formats such as DPSK [1-6] and the development of strong FEC techniques [7]. On the other hand, ultra high spectral efficiencies of up to 1.6 bit/s/hz have been reported for 40 Gbit/s [8-11]. In [11], 8 40 Gbit/s RZ- DQPSK signals were transmitted over a record 200 km NDSF with the highest 40 Gbit/s spectral efficiency of 1.6 bit/s/hz demonstrated to date. In this paper, we investigate orthogonally polarised (OP)-CSRZ-DQPSK transmission at 40 Gbit/s/ wavelength and demonstrate transmission of 8 40 Gbit/s over 320 km NDSF at 1.6 bit/s/hz spectral efficiency. We also demonstrate the capability of OP-CSRZ-DQPSK for a significant extension of 40 Gbit/s chromatic dispersion tolerance and PMD tolerance in the absence of any adjustable dispersion compensators. 2. System configuration 1 2 : 8 PM Coupler CSRZ Pulse 80 km GHz 20 Gbit/s 40 Gbit/s NDSF Integrated DQPSK Modulator 3 db PMC EDFA DCM 80 km NDSF DCM 80 km NDSF Gbit/s BERT Balanced Receiver Balanced Receiver Balanced Receiver Balanced Receiver DQPSK Decoder DQPSK Decoder PBS 0.6 nm 0.25 nm 0.25 nm BPF BPF BPF Fig. 1 System configuration DCM 80 km NDSF DCM The experimental set-up is shown in Fig.1. The channel wavelengths varied from nm to nm with a spacing of 0.2 nm (25 GHz). They were combined through a PM coupler (PMC). The channels went through a chirp-free LiNbO 3 modulator driven at 5 GHz for CSRZ modulation. The CSRZ pulses had a duty cycle of approximately 60%. The second modulator was an integrated GaAs/AlGaAs DQPSK data modulator. As a result 20 Gbit/s CSRZ-DQPSK signals were produced. The phase difference of /2 between in-phase and quadrature channels was maintained through an integrated two-photon-absorption monitor. The 20 Gbit/s signals were then split, delayed and recombined to form orthogonally polarized OP-CSRZ-DQPSK signals at 40 Gbit/s per wavelength. The time delay between dual polarizations was set at 145 ns and de-correlation was confirmed to be sufficient. The transmission link included four 80 km spans with EDFA-only amplification. Transmission fibre loss varied from 15.6 db to 19 db and DCM loss was around 9.5 db. DCM dispersions at 1557 nm were ps/nm, ps/nm, ps/nm and ps/nm. At the receiver, the channels were wavelength and polarization demulptiplexed to individual 20 Gbit/s signals before demodulation. The demodulator was a temperature-controlled

2 TuF1 interferometer with a relative delay of around 0 ps. The in-phase or quadrature demodulated channels were received with a Gbit/s balanced receiver with a bandwidth of around 14 GHz. The pattern generator made use of pre-coded data and the error detector was programmed to receive the corresponding sequence. A data pattern length of 2 15 was used for all the measurements. This pattern length was only limited by the memory of the pattern generator. 3. Experimental results Chromatic dispersion and PMD tolerance: Q penalty (db) CSRZ-DQPSK CSRZ-IMDD Q penalty (db) CSRZ-DQPSK CSRZ-IMDD Chromatic dispersion (ps/nm) Fig.2 Measured chromatic dispersion tolerance DGD (ps) Fig.3 Measured PMD tolerance The measured chromatic dispersion and PMD tolerance of a 40 Gbit/s OP-CSRZ-DQPSK signal are presented in Fig.2 and Fig.3, respectively. They were compared with the tolerances of a 40 Gbit/s ETDM-based CSRZ-IMDD signal. A significant extension of the 40 Gbit/s chromatic dispersion tolerance and PMD tolerance was observed. The 2dB (20log Q) penalty chromatic dispersion tolerance of a 40 Gbit/s OP-CSRZ-DQPSK signal was measured to be 950 ps/nm. This represents an 8-fold increase compared to the dispersion tolerance of a 40 Gbit/s ETDM-based CSRZ-IMDD signal, which was about 1 ps/nm. A decoded eye diagram for a CSRZ-DQPSK signal in the presence of +680 ps/nm dispersion is shown as the inset in Fig.2. Similarly, comparison of the measured PMD tolerance is shown in Fig.3. The 1dBQ penalty PMD tolerance of OP-CSRZ-DQPSK and CSRZ-IMDD at 40 Gbit/s was 22 ps and around ps, respectively. A doubling of the PMD tolerance (for 1dBQ penalty) was achieved compared with 40 Gbit/s CSRZ-IMDD. Such benefits result from the use of a symbol rate of Gbaud instead of 40 Gbaud for a capacity of 40 Gbit/s/wavelength. Note that no adjustable chromatic dispersion or PMD compensators were used. 1.6 bit/s/hz OP-CSRZ-DQPSK transmission: Power (dbm) (a) 40 Gbit/s CSRZ-DQPSK GHz CSRZ pulse Power (dbm) (b) filtered 40G TE filtered 40G TM Wavelength (nm) Wavelength (nm) Fig.4 Back-to-back spectra of (a) 1.6 bit/s/hz 8 40 Gbit/s OP-CSRZ-DQPSK and the corresponding GHz CSRZ intensity modulation; (b) wavelength and polarization demultiplexed Channel 4. Fig.4(a) shows the back-to-back 1.6 bit/s/hz OP-CSRZ-DQPSK signal spectrum, together with the spectrum of the original GHz CSRZ pulses used for the intensity modulation. The inset in Fig.4(a) is a typical eye diagram of a 40 Gbit/s OP-CSRZ-DQPSK signal before demodulation. Fig.4(b) gives the performance of the combined

3 TuF1 wavelength and polarization demultiplexing, showing the spectra of each polarization of channel 4 after demultiplexing at the receiver. Despite the absence of pre-filtering at the transmitter, the crosstalk between adjacent wavelength channels was 12.2 db through post-filtering and the polarization extinction was more than 30 db. Transmission over 320 km NDSF was then performed. The launch power used was 3 dbm/channel for the transmission fibre and 6 dbm/channel for the DCMs. Fig.5 shows the received eye diagrams for the even channels. Clear eye opening was achieved for all wavelength channels. No phase adjustment between in-phase and quadrature channels for different wavelengths was necessary at the integrated data transmitter. Fig.6 gives the measured Q- factors for all the channels. The Q-factor varied from 14.8 db to 16.3 db, with an average Q-factor of 15.4 db in the absence of FEC Q-factor (db) Fig.5 Received eye diagrams for even channels at 320 km Wavelength (nm) Fig.6 Measured Q-factor vs. wavelength at 320 km k The significant performance margin at 320 km suggests a possibility of a longer reach at 1.6 bit/s/hz with the use of FEC techniques. With a view to future long reach applications of such high spectral efficiency 40 Gbit/s DQPSK systems, it is expected that a polarization tracking receiver will play a critical role [12]. A polarization tracking receiver was not employed in this experiment but in a real system a polarisation tracking subsystem will be necessary. 4. Conclusion In conclusion, we have successfully transmitted 8 40 Gbit/s OP-CSRZ-DQPSK signals over a distance of 320 km at a spectral efficiency of 1.6 bit/s/hz. This was enabled by using OP-CSRZ-DQPSK modulation format based on an integrated DQPSK data transmitter. The high spectral efficiency of 1.6 bit/s/hz for OP-CSRZ-DQPSK was achieved in the absence of any pre-filtering or forward error correction. We also experimentally investigated the chromatic dispersion and PMD tolerance of a 40 Gbit/s OP-CSRZ-DQPSK signal. A chromatic dispersion tolerance of 950 ps/nm and PMD tolerance of 25 ps at 40 Gbit/s was successfully demonstrated. This represents an 8-fold extension of the chromatic dispersion tolerance and a doubling of PMD tolerance of a typical 40 Gbit/s ETDM system but in the absence of any adjustable chromatic dispersion or PMD compensators. The significant increase in system tolerance benefited from the use of a Gbaud symbol rate for a capacity of 40 Gbit/s/wavelength. Polarization tracking is expected to be required for future long haul applications of high spectral efficiency DQPSK systems. References [1] A. H. Gnauck, et al, OFC 2002, paper PD-FC2. [2] C. Rasmussen, et al, OFC 2003, paper PD18. [3] B. Zhu, et al, OFC 2003, paper PD19. [4] T. Tsuritani, et al, OFC 2003, paper PD23. [5] J.-X. Cai, et al, OFC 2003, paper PD22. [6] J. Marcerou et al, OFC 2003, paper PD20. [7] T. Mizuochi, et al, OFC 2003, paper PD21. [8] S. Bigo, et al, OFC 2001, paper PD25. [9] S. Sotobayashi, et al, IEEE Photon. Technol. Lett., vol.14, pp.555, (2002). [] I. Morita, et al, ECOC 2002, paper PD4.7. [11] C. Wree, et al, IEEE Photon. Technol. Lett., (2003). [12] C. Davidson, et al, OFC 2003, paper TuF3.

4 TuF2 Study on Optimum Pre-Filtering Condition for 42.7 Gbit/s CS-RZ DPSK Signal Keiji Tanaka, Itsuro Morita, and Noboru Edagawa KDDI R&D Laboratories Inc., Ohara Kamifukuoka Saitama, , Japan TEL: , FAX: , Abstract: The impact of asymmetric pre-filtering for 42.7Gbit/s CS-RZ DPSK signals were investigated experimentally and numerically, including 9000km WDM transmission experiments. We have found that asymmetric pre-filtering is effective for quasi-linear system applications Optical Society of America OCIS codes: (060.45) Optical Communication 1. Introduction Spectral efficiency is a key issue to enhance aggregate capacity and cost-effectiveness of optical transmission systems. In order to achieve high spectral-efficiency, the signal bandlimitation by using optical filtering before transmission (i.e. pre-filtering) is an attractive solution, and many successful demonstrations using this scheme have been reported to date [1]-[9]. Regarding on-off keying (OOK) signals, detailed studies on the benefit of asymmetric pre-filtering have been already reported for carrier-suppressed return-to-zero (CS-RZ) signals [7][]. However, regarding differential phase-shift keying (DPSK) signals, which is a promising format for high bit rate ULH system applications, similar study has not been reported so far, whereas remarkable demonstrations of 40Gbit/s-based DWDM transmission have been already reported by using symmetrically pre-filtered DPSK signals [6][8][9]. In this paper, we have experimentally and numerically investigated the impact of asymmetric pre-filtering for 42.7 Gbit/s CS-RZ DPSK signal by comparing the dispersion and nonlinear tolerance between unfiltered, asymmetrically and symmetrically filtered CS-RZ DPSK signals. In addition, by conducting a DWDM transmission experiment up to 9000km, we have evaluated the long-haul transmission performance of asymmetrically pre-filtered CS-RZ DPSK signals. 2. Back-to-back performance of pre-filtered CS-RZ DPSK signals Fig. 1 shows the schematic experimental setup employed for various evaluations. In the transmitter, we used a DFB laser source and two LiNbO 3 modulators; one is for 42.7 Gbit/s data-coding of the optical phase of the signal in NRZ format with a true pseudo-random binary sequence, and the other is for bit-synchronous CS-RZ formatting. Then, the 42.7 Gbit/s CS-RZ DPSK signal was optically bandlimited by using a 50/0 GHz interleaving device with a 3 db/20db bandwidth of 45 GHz/68GHz. Here, we assumed an ultra-dense WDM system with a spectral efficiency of 80%. In the receiver, the DPSK signal was demodulated using a Mach-Zehnder delay interferometer (MZDI) and detected by a balanced receiver. The received 42.7Gbit/s electrical signal was demultiplexed to.7gbit/s. To evaluate the signal performance, we measured a bit error rate (BER) averaged over the four tributary Gbit/s data streams using a random re-synchronization technique [5] and calculated the corresponding Q-factor. OS DFB- LD DPSK/ CS-RZ MOD 50/0 GHz-IL Optical Filter (1) Filter Detuning (3) Fiber Input Power Tolerance EDFA 50km SMF direct connection (2) Dispersion Compensation Error Tolerance DCF DCF ASE Source ATT MZDI Balanced Receiver Random Re-synchronizer OR Error Detector Clock Recovery Fig. 1. Schematic experimental setup for the evaluation of bandlimiting filter detuning characteristics, dispersion and nonlinear tolerance. In the evaluation of the optimum pre-filtering condition, we bypassed the transmission line and the dispersion compensation fiber at the receiver and measured the back-to-back performance by detuning the center frequency of

5 TuF2 the bandlimiting filter. Fig. 2 (a) shows the Q-factor penalty due to optical bandlimitation for 42.7Gbit/s CS-RZ DPSK signal as a function of the filter detuning defined as the difference between the signal carrier frequency and the center frequency of the bandlimiting filter. The nominal Q-factor of the unfiltered signal was set to 14.8 db by adding the amplified spontaneous emission noise to simulate a long-distance transmission condition. As shown in Fig. 2 (a), the Q-factor penalty of the asymmetrically filtered signal due to bandlimitation was distinctively lower than that of the symmetrically filtered signal, and was less than 1dB. This low penalty condition was kept for around 7.5GHz+/-2.5GHz detuning range, which indicates that the asymmetrically filtered CS-RZ DPSK signal was robust against center frequency fluctuations of the bandlimiting filter. Next, in order to investigate the characteristics of bandlimited CS-RZ DPSK signals further, we conducted numerical simulations and the results are summarized in Fig. 2. In the simulation, the bandlimiting filter was assumed to 4 th -order Gaussian with a 3dB bandwidth of 45GHz, and the performance was evaluated by eye-opening penalty. As shown in Fig. 2 (a), the simulation results agree well with the measured results. In order to find out the source of superior performance of asymmetrical filtering over symmetrical filtering, we investigated the signal waveform after the bandlimiting filter, as shown in Fig. 2(b) and (c). From Fig. 2(b) and (c), we found that the difference in signal power fluctuations between symmetrically and asymmetrically filtered signals can be attributed to the additional Q-factor penalty of symmetrically filtered signal (a) Simulation Experiment Filter Detuning [GHz] 0 Fig. 2. Q-factor penalty due to bandlimitation as a function of filter detuning (a) and optical signal waveform of (b) symmetrically and (c) asymmetrically filtered (7.5GHz detuning) 42.7 Gbit/s CS-RZ DPSK signals. 3. Tolerance evaluation of pre-filtered CS-RZ DPSK signals Next, we evaluated the dispersion and nonlinear tolerance by measuring the penalty due to dispersion compensation error at the receiver and the increase in fiber input power for unfiltered, symmetrically filtered and asymmetrically filtered CS-RZ DPSK signals. The filter detuning was set to 7.5GHz for asymmetric filtering. In the evaluation of dispersion tolerance, the signal from the transmitter was directly fed into the DCFs with various total dispersion values. In the evaluation of nonlinear tolerance, we used a transmission line consisting of 50km of SMF followed by a DCF, which fully compensated for the accumulated dispersion of the SMF. Fig. 3(a) and (b) show the Q-factor penalty measured as a function of dispersion compensation error and fiber input power, respectively. The circles, triangles, and squares show the Q-factor penalty obtained from the experiment, and the solid, dot-dashed, and dashed lines show the eye opening penalty obtained from the numerical simulations for unfiltered, symmetrically and asymmetrically filtered signals, respectively. The acceptable dispersion compensation error for 1dB Q-factor penalty was found to be about 50ps/nm for the unfiltered and asymmetrically filtered signal and 140ps/nm for the symmetrically filtered signal. Note that this large dispersion error tolerance of symmetrically filtered CS-RZ-DPSK signal was not observed for the CS-RZ OOK signal [] and seems a specific feature of optically pre-filtered CS-RZ DPSK signal. Regarding the nonlinear tolerance, for all the filtering conditions, almost no performance degradation was observed up to the fiber input power of +dbm. However, as the fiber input power exceeded +dbm, distinct penalty appeared in the order of asymmetrically, symmetrically filtered and unfiltered signal. These phenomena were the same as the case for CS-RZ OOK signal [], and this indicates that there is a trade-off between the back-to-back performance and nonlinear tolerance for asymmetric optical bandlimitation, as in the case of CS-RZ OOK signal [7][].

6 TuF (a) 3 3 (b) Dispersion Comensation Error [ps/nm] Fiber Input Power [dbm] Fig. 3. Tolerance against (a) dispersion compensation error and (b) fiber input power of unfiltered (circles and solid lines), asymmetrically filtered (7.5GHz detuning, squares and dot-dashed lines) and symmetrically filtered (triangles and dashed lines) 42.7 Gbit/s CS-RZ DPSK signals. Points and lines show the experimental and numerical simulation results, respectively. Finally, in order to confirm the long-haul transmission performance of symmetrically and asymmetrically pre-filtered CS-RZ DPSK signals, we conducted a 50GHz-spaced 64 x 42.7Gbit/s transmission experiment up to 9000km by using a 360km-long re-circulating loop with 43km-long dispersion-flattened fiber spans [7] and 980nm-pumped EDFA repeaters. The odd and even channels were orthogonally polarization multiplexed. We measured the transmission performance of the channel 32, the center channel, as a function of transmission distance. Fig. 4 shows the results for -6dBm/ch and -5dBm/ch repeater output power with symmetric and 3.8GHz-detuning asymmetric filtering. For 6000km transmission at 5dBm/ch repeater out-put, the Q-factor calculated from the obtained BER was 11.1dB and 11.7dB for 7.5GHz and 3.8GHz detuning asymmetrically filtered signal, 12dB for symmetrically filtered signal. On the other hand, for 9000km transmission at 6dBm/ch repeater out-put, the Q-factor calculated from the obtained BER was 9.6dB and 9.7dB for 7.5GHz and 3.8GHz detuning asymmetrically filtered signal, 9.3dB for symmetrically filtered signal. These results show that asymmetric filtering is effective when the fiber nonlinearity per repeater span is well suppressed dBm/ch -6dBm/ch Symmetrically filtered signal Asymmetrically filtered signal Distance [km] Fig. 4. Long-haul transmission performance of 42.7 Gbit/s symmetrically and asymmetrically (3.8GHz detuning) filtered CS-RZ DPSK WDM signals. The solid and dashed lines show the results for -6dBm/ch and -5dBm/ch repeater output, and the circles and triangles show the results of symmetrically and asymmetrically filtered signals. 4. Conclusion We have experimentally and numerically investigated the impact of asymmetric pre-filtering for 42.7 Gbit/s CS-RZ DPSK signals. By detuning the bandlimiting optical filter, the back-to-back performance was distinctively improved with sufficient robust against center frequency fluctuations of the bandlimiting filter, although the nonlinear tolerance was decreased. Through this study, we have found that asymmetrically pre-filtering is effective for quasi-linear DWDM system applications. 5. References [1] T. Tsuritani et al., ECOC2000, PD1.5, (2000). [2] G. Charlet, et al., OAA2002, PDP1, (2002) [3] T. Tsuritani et. ai., ECOC2002, 9.1.4, (2002) [4] G. Charlet, et al., ECOC2002, PD4.1, (2002). [5] I. Morita, et al., ECOC2002, PD4.7, (2002). [6] H. Gnauck et al., Photon. Tech. Lett., 15, pp , (2003). [7] T. Tsuritani et al., OFC2003, FE4, (2003). [8] B. Zhu et al., OFC2003, PD19, (2003). [9] T. Tsuritani et al., OFC2003, PD23, (2003). []A. Agata et al., OFC2003, MF78, (2003).

7 TuF3 Nonlinear tolerance of differential phase shift keying modulated signals reduced by XPM B. Spinnler Siemens AG, Corporate Technology, Otto-Hahn-Ring 6, D Munich Germany N. Hecker-Denschlag, S. Calabrò, M. Herz, C.-J. Weiske, E.-D. Schmidt Siemens AG, ICN Optical Solutions, Hofmannstrasse 51, D Munich Germany D. van den Borne, G.-D. Khoe, H. de Waardt COBRA Institute, Eindhoven University of Technology, The Netherlands R. Griffin, S. Wadsworth, Bookham Technology PLC., Caswell, Towcester, Northants, NN12 8EQ England Abstract: We show that in order to maintain the high nonlinear tolerance of the DQPSK modulation format, XPM from neighboring OOK-modulated channels must be avoided. The negative impact on Gb/s DQPSK channels is higher than at 20Gb/s Optical Society of America OCIS codes: (060.45) Optical communications; ( ) Phase modulation 1. Introduction Advanced phase modulation formats have been very successful in extending the reach and capacity of ultra long haul, high capacity, optical transmission systems [1,2]. This success is not only due to the use of balanced receivers for a gain of 3dB at the receiver, but also that D(Q)PSK modulation is less sensitive to non-optimum dispersion compensation [3] and also allows for a larger input power before nonlinear effects degrade the signal quality [4,5]. While the high nonlinear tolerance of systems running solely with D(Q)PSK can be exploited for extending the reach of high capacity systems or increasing amplifier spacing, the effect of mixed usage, i.e. combining OOK and D(Q)PSK modulation formats at different wavelengths, on the individual channel tolerances still needs to be determined. For instance mixed usage can occur when installed systems are upgraded with new modulation formats or when meshed networks come into operation. In such scenarios, wavelength channels with both D(Q)PSK and OOK modulation formats can become nearest neighbors and the nonlinear interactions must be understood. The robustness of binary DPSK in presence of NRZ modulated channels with 0 GHz spacing for transmission over SSMF has been discussed in [6]. In this paper we discuss the reduction of the nonlinear tolerance of NRZ-DQPSK because of XPM from NRZ-OOK neighboring channels at a 50 GHz spacing. 50 GHz spacing is used instead of 0 GHz spacing as it represents the standard today for OOK systems and is more likely to be limited by XPM effects. 2. Experimental setup The two experimental setups shown in fig. 1a and 1b were used to compare the transmission performance of five wavelength channels on a 50 GHz grid over 0 km SSMF. The middle channel was always modulated with a 20 Gb/s DQPSK ( Gsymbol/s). In the first setup (fig. 1a) the four remaining channels were also 20 Gb/s DQPSK modulated. In the second setup (fig. 1b) these channels were Gb/s OOK modulated. We chose half the bit rate for the OOK channels in order to have similar spectral width of the two types of formats. Therefore, this bit rate seems a natural choice considering Gb/s OOK systems in operation today. The DQPSK modulation was achieved with a GaAs-based modulator [4] which is composed of two Mach-Zehnder interferometers (MZI) offset biased around the zero transmission point so that with a Gb/s modulation, a phase difference between bits of is obtained. Each of the MZI are modulated with the same pseudo-random bit sequence with a length , but a delay of 22 bits between the two allows for a decorrelation of the bits, since a precoder was unavailable. The two MZIs are combined with an added /2 phase shifter in a super MZI structure. We used a tunable dispersion compensator in order to optimize performance. Full compensation was found to be optimal for DQPSK. The total input power of the five wavelength channels was varied before the transmission fiber to study the effect of the XPM. At the end of line, we used a 0.2 nm passband filter to extract the channel under consideration. Differential demodulation was performed by an optical one-symbol delayed MZI. The two outputs of the demodulator were differentially detected with a balanced receiver and a limiting differential amplifier. In order to measure a BER, the receiver is given the corresponding bit sequence

8 TuF3 for detection of either the I or Q channel. At the output of the differential amplifier, a noise loading experiment was performed to determine the BERs of the middle channel as a function of the per channel power and OSNR. In the mixed modulation setup we additionally adjusted the polarization of the neighboring channels so that they were either parallel or orthogonal to the middle DQPSK channel under consideration in order to better observe the XPM effect. In the DQPSK-only setup all channels had an identical polarization. All channels have equal power independent of the modulation format. a) QPSK Mod Integrated device + - VOA MZ-Demod 0 km SSMF 0.2 nm DCF 0% 2.8 nm b) MZI QPSK Mod Integrated device + - VOA 3 db Coupler MZ-Demod 0 km SSMF 0.2 nm DCF 0% 2.8 nm Fig. 1. Experimental setup with a) 5 DQPSK channels, b) one DQPSK channel and 4 OOK channels. 3. Results In order to assess the effect of XPM on the middle DQPSK channel, we made measurements and simulations for both systems shown in fig 1. We evaluated the BER of the DQPSK channel for various values of the transmitted power. Fig. 2 shows the measured BER versus OSNR for total input powers ranging from 7 to 18.5 dbm. The figures in the first row show results obtained by measurements, the figures in the second row show the corresponding simulation results. The figures in the left column (a and d) present the BER for the system with five DQPSK channels, the figures in the middle column (b and e) show the BER for the system with one DQPSK channel and four OOK neighbors (all polarizations aligned parallel), and the figures in the right column (c and f) show the results for the system with one DQPSK channel and four OOK neighbors (polarizations of the OOK channels orthogonal to that of the middle DQPSK channel). Let us first discuss the measurement results. It can be seen that in all cases the higher the launch power the more the performance is degraded by XPM from the neighboring channels. However, the results differ widely if we compare them quantitatively. When we increase the launch power from 7 to 16 dbm the OSNR penalty for the DQPSK-only system is about 2.5 db at BER= 9. If the polarization of the OOK neighboring channels is parallel to the DQPSK channel the penalty is 2 db for 7 dbm launch power and the penalty for 16 dbm launch power could not be determined at 9 because the signal is significantly degraded. When the polarization of the OOK neighboring channels is orthogonal to the DQPSK channel, the performance is equal to that of the DQPSK-only system for 7 dbm launch power, and the OSNR penalty for 16 dbm launch power is about 6 db. The corresponding simulation results are shown in the second row of fig. 2. BERs above -5 were measured directly using Monte-Carlo simulation. For lower BERs we employed the tail extrapolation technique. While the results differ quantitatively from the measurements because we did not include all relevant impairments into the simulation (e.g. in the simulation we did not include laser phase noise and used ideal filters, de/modulators and clock recovery), the general trend in the measurements and the simulations is the same. We observe only a slight degradation for an increase in the launch power when we use DQPSK for the neighboring channels. If we use OOK with parallel polarization with respect to the middle DQPSK channel for the neighboring channels, the degradation is dramatic. If we use OOK with orthogonal polarization with respect to the middle DQPSK channel, the degradation is still larger than in the DQPSK-only system, but far less than in the case with parallel polarized OOK neighbors. These results are in agreement with our measurements and support our claim that the performance of DQPSK depends very much on the type of the modulation format and polarization of the DQPSK s neighboring channels. A prevalent measure to further enhance the dispersion tolerance of DQPSK is the reduction of the data rate. In order to investigate the influence of the data rate we repeated the simulations corresponding to the setup shown in fig. 1b with the DQPSK data rate reduced to Gb/s. The neighboring channels again use Gb/s OOK with the same polarization (worst case). Fig. 3 compares the simulation results for the cases 20 Gb/s (fig. 3a) and Gb/s (fig. 3b). While for low input power there is the usual 3 db gain in favour of the Gb/s system, we observe that for higher input power the Gb/s system performs even worse than the 20 Gb/s system. Hence, the XPM from OOK neighbors is more detrimental for the Gb/s channel than for the 20 Gb/s channel. This effect has to be taken into account when an OOK channel is to be replaced by a more tolerant DQPSK channel with the same data rate.

9 TuF3 a) -4-5 P = 7 dbm P = 12 dbm P = 16 dbm P = 18.5 dbm b) c) BER -6 BER -6 BER P = 7 dbm P = 12 dbm P = 14 dbm P = 15 dbm P = 16 dbm P = 7 dbm P = 12 dbm P = 14 dbm P = 16 dbm P = 18 dbm d) OSNR (dbm) P = 7 dbm P = 14 dbm P = 18 dbm OSNR 0 0 e) f) OSNR (dbm) BER OSNR [db] BER P = 7 dbm P = dbm 7 P = 12 dbm P = 14 dbm 8 P = 16 dbm P = 18 dbm OSNR [db] BER P = 7 dbm P = dbm 7 P = 12 dbm P = 14 dbm 8 P = 16 dbm P = 18 dbm OSNR [db] Fig. 2. BER of middle DQPSK channel versus OSNR. a) and d) 5 DQPSK channels, b) and e) DQPSK and 4 OOK channels (polarization of all channels parallel), c) and f) DQPSK and 4 OOK channels (polarization of OOK channels orthogonal to middle DQPSK channel). a) c) Measurements, d) f) Simulations Gb/s P = 7 dbm 1 P = dbm P = 12 dbm 2 P = 14 dbm a) b) 0 Gb/s P = 7 dbm 1 P = dbm P = 12 dbm 2 P = 14 dbm BER BER OSNR [db] OSNR [db] Fig. 3. Performance of DQPSK with 4 Gb/s OOK neighboring channels (polarization of all channels parallel). The DQPSK channel runs at 20 Gb/s (a) and Gb/s (b), respectively. 4. Conclusions Since practically all systems today use OOK modulation, the tolerance to interference generated by neighboring OOK channels will become a major criterion for the deployment of optical alternative modulation schemes. In this paper we addressed the, for practical reasons, very important case of and 20 Gb/s DQPSK channels being turned on in a 50 GHz WDM grid nearby Gb/s OOK channels. We showed by measurements and simulations that a middle 20 Gb/s DQPSK channel surrounded by four OOK channels suffers from a very large OSNR penalty even for moderate launch powers when the polarization of the neighboring channels is aligned parallel to the middle channel. Part of this degradation is recovered when the polarization of the neighboring channels is orthogonal to the middle DQPSK channel. But this cannot be guaranteed unless polarization interleaving is employed at the transmitter site. Furthermore, we showed by means of simulations that a Gb/s DQPSK channel is even more vulnerable to the effect of XPM from OOK neighboring channels for moderate to high launch power. However, elaborate dispersion maps can be chosen such that XPM interference is minimized. A verification by measurements and thorough investigation of this issue represent an open field for future work. 5. References [1] C. Rasmussen et al. DWDM 40G transmission over trans-pacific distances (,000km) using CSRZ-DPSK, enhanced FEC and all-raman amplified 0 km UltraWave fiber spans, OFC 2003, PD18. [2] B. Zhu et al. 6.4-Tb/s (160 x 42.7 Gb/s) transmission with 0.8 bit/s/hz spectral efficiency over 32 x 0 km of fiber using CSRZ-DPSK format, OFC 2003, PD19. [3] H. Bissessur et al. 1.6 Tb/s (40 x 40 Gb/s) DPSK transmission with direct detection, ECOC 2003, paper [4] R.A. Griffin et al. Gb/s optical differential quadrature phase shift key (DQPSK) transmission using GaAs/AlGaAs integration OFC 2002 FD6. [5] C. Wree et al. RZ-DQPSK format with high spectral efficiency and high robustness towards fiber nonlinearities, ECOC 2002, paper [6] M. Rohde et al., Robustness of DPSK direct detection transmission format in standard fibre WDM systems, Electron. Lett. Vol. 36, No. 17, Aug. 2000, pp

10 TuF4 Experimental comparison of non linear threshold and optimum pre dispersion of 43 Gb/s ASK and DPSK formats R. Dischler, A. Klekamp, J. Lazaro, W. Idler Alcatel Research and Innovation, Holderaeckerstrasse 35, D Stuttgart, Germany Abstract: 43Gb/s ASK and DPSK formats, each with 4 duty-cycles, are compared on single channel nonlinear impairments. We measured the nonlinear threshold, pre-dispersion optimum and tolerance and found a NLT benefit for DPSK up to 2dB Optical Society of America OCIS codes: ( ) Fiber optics communications, ( ) Modulation 1. Introduction Comprehensive experimental analysis of modulation formats regarding system performance and tolerances, linear and nonlinear transmission impairments are needed for cost effective system design. Comparisons of linear fiber impairments at on-off keying (OOK) and differential phase shift keying (DPSK) formats at 43 Gb/s have been reported [1,2] showing the advantages of DPSK formats: 3 db of OSNR gain, enhanced tolerances on chromatic dispersion and differential group delay by balanced detection. The duty cycle has been identified as beneficial parameter for NRZ format and RZ formats differentiation [2]. On one hand, simulations show that 43Gb/s DPSK formats benefit also from reduced nonlinear effect [3,4], but on the other hand, nonlinear phase noise has been identified to effect at DPSK formats adversely, potentially reducing the gain achieved by the use of a balanced receiver, at Gb/s even more severe than at 40 Gb/s [5]. However, DWDM experiments clearly show the benefits of DPSK formats at 40 Gb/s [6,7] and also at Gb/s [8,9] achieving impressive submarine transmission distances. Concerning single channel nonlinear transmission impairments key system design parameters are the nonlinear phase shift [,11] or the nonlinear threshold (NLT), in other words: the maximum transmission fibre input power, varying inversely proportionally with span number [11]. In this paper we present an experimental survey of NLT, including the optimum pre dispersion value and pre dispersion tolerance for single channel 43 Gb/s formats. RZ and NRZ modulation at ASK and DPSK formats by transmission experiments over 1 and 3 spans of SMF. We compare systematically the modulation formats across the different RZ duty cycles ranging from 33%, 50% to 66% (CSRZ) and finally 0% is applied for NRZ. 2. Experimental set up Fig. 1 shows the experimental set up for the measurement of NLT of the 8 modulations formats [2] for the transmission over 1 and 3 spans of SMF, with a span length of 80km. TX TX Pre DCF Pre DCF Att. P 1 Att. P 80km SMF 80km SMF Inline DCF Att. Att. P 2 Post DCF OSNR 80km SMF ASE- Filter ~ Inline DCF RX Att. P 3 80km SMF Att. Post DCF OSNR Fig. 1. Set up for NLT evaluation over 1 span (top) and 3 spans(bottom) of SMF ASE- Filter ~ RX RZ-33% RZ-50% RZ-66% NRZ (0%) Fig. 2: 43Gb/s DPSK eye-pattern (experimental, back-to-back, PRBS , balanced detection), all 4 duty cycles; hor: ps/div, vert.: different scaling With 6 Mach-Zehnder modulator configurations at the transmitter (Tx) we realized 8 modulation formats: 4xASK and 4xDPSK [2], each with 3 different RZ duty cycles of 33%, 50%, 66% (CSRZ) and NRZ modulation, which we will denote as 0% duty cycle. Examples of 4xDPSK format eye-pattern are shown in Fig. 2 at back-toback configuration, showing a good quality of the DPSK Tx. All experiments were carried out with one single optical channel at 1552nm.

11 TuF4 The Tx is attached to a double stage booster EDFA with interstage dispersion compensating fiber (DCF) for pre compensation. This booster EDFA has a fixed optical output power of 18.5 dbm. The input power to the SMF, can be varied with optical attenuators. The launched power is controlled via 20dB tap by an optical spectrum analyzer (OSA) at the span input. For the 3 span experiment this configuration is repeated 3 times. The inline dispersion compensation is set to 5%. At the end of the last span, a 4 th attenuator is used to vary the received optical power, i.e. to vary the OSNR at the receiver (Rx), measured with an OSA of 1 nm resolution bandwidth. The DCF of the post compensation could be varied in steps of about ps/nm to optimize the residual dispersion at the Rx-side. The input power into all DCF was maintained below 0 dbm, to ensure linear propagation through the DCFs. For the DPSK formats a Mach-Zehnder DPSK-demodulator was inserted in front of the receiver. BER measurements are performed here with single ended detection for the DPSK formats. From the measured OSNR value a corrected OSNR for 0.1nm bandwidth was calculated. OSNR penalty was measured with respect to BER of -5 using PRBS length of for all measurements. As non linear threshold we denote the fiber input power, which gives a penalty of 1dB with respect to OSNR sensitivity at linear transmission. To find the optimum pre compensation for each format, the DCF in the booster was varied from 0 to -420 ps/nm in steps of about 0ps/nm. For each pre compensation the post compensation was optimized at an SMF input power of 0dBm, i.e. in the linear regime. When the fiber input power was increased, only the decision threshold and the sample phase at the receiver was adapted. 3. Results and discussion Fig. 3 show the results of the NLT measurements for ASK and DPSK, e.g. with RZ modulation of 50% duty cycle. The measurement uncertainty for the NLT value is about 0.5dB. To determine the optimum pre dispersion, the maximum NLT and the pre dispersion tolerance with respect to NLT, we used a parabolic fit over the measured NLT data. NLT (dbm) ASK - RZ50% 14 1 span spans Precompensation (ps/nm) NLT (dbm) DPSK - RZ50% 1 span 3 spans Precompensation (ps/nm) Fig. 3. Non linear threshold (NLT) vs. variation of pre dispersion for 1 and 3 spans over SMF, ASK at left, DPSK at right side NLT (dbm) ASK span spans duty cycle (%) NLT (dbm) DPSK span spans duty cycle Fig.4:.Maximum non linear threshold vs. duty-cycle for 1 and 3 spans over SMF, ASK at left, DPSK at right side Fig. 4 summarizes the maximum NLT data from parabolic fit, we obtained for 4xASK and 4xDPSK formats by 1 and 3 span measurements. The difference for NLT data between 1 and 3 spans is about 5 db, indicating that 5% of inline compensation was a suitable value for all formats, since the expected difference between 1 and 3 span NLT data is 4.8dB according to a log(1/n)-correlation [], where N is the number of spans. DPSK formats show higher NLT than the corresponding ASK formats, except at RZ 66%: DPSK benefits by 2dB at NRZ format, and up to 1 db at RZ formats. The highest NLT of 15dBm at 1 span configuration was observed with DPSK-RZ33%.

12 TuF4 Further measurements under post dispersion optimisation at each fiber input power yield even higher NLT values than shown in fig. 4, e.g. NLT of ASK-NRZ could be increased by 2 db. pre-dispersion (ps/nm) ASK 3 spans 1 span 3 spans 1 span duty cycle (%) pre-dispersion tolerance (ps/nm) pre-dispersion (ps/nm) DPSK duty cycle (%) Fig.5. Optimum pre- dispersion (solid lines) and pre dispersion tolerance (dashed lines) vs. duty -cycle for 1 and 3 spans over SMF, ASK at left, DPSK at right side Fig. 5 shows the optimum pre dispersion and the pre dispersion tolerance of NLT for all 8 modulations formats, 4xASK at left and 4xDPSK at right side. The optimum pre dispersion of ASK formats is about -200 ps/nm with low variation over the duty-cycles. The positive shift of the optimum pre dispersion value for 3 spans is attributed to the residual inline dispersion of about 70ps/nm [12]. At DPSK formats we observed a more pronounced shift of optimum pre dispersion for 3 spans and a higher variation across the duty cycles than at ASK formats. The pre dispersion tolerance, i.e. the full width 1dB down of the parabolic fit in Fig. 1, where NLT is 1dB below maximum, is within 300 and 400 ps/nm for all ASK and DPSK formats. We received no clear trend within the ASK (but small increase with duty cycle) and the DPSK formats (but small decrease with duty cycle). Note, that the NLT measurement accuracy of 0.5dB yield an pre dispersion tolerance accuracy of 50ps/nm. 4. Summary We have experimentally evaluated single channel nonlinear threshold for transmission over SMF with ASK and DPSK formats with 4 different duty cycles from 33% to 0% (NRZ). At NRZ format, DPSK show 2dB higher NLT. At RZ formats, the NLT gain by DPSK is up to 1dB. Highest NLT of 15dBm for 1 span configuration was observed with DPSK-RZ33%. We confirmed a log(1/n) correlation of NLT from 1 and 3 spans measurement result. The optimum pre dispersion was found around -200 ps/nm across all formats, with a higher variation over duty cycles for DPSK formats than for ASK formats. The pre dispersion tolerance is comparable for all formats between 300 and 400ps/nm with no clear trend over the duty cycles. 5. References [1] F. An et al, Comparison of linear fiber impairments tolerance among 40Gb/s modulation formats, in Proc. OFC 2003, vol.2, paper FE2, pp [2] W.Idler et al, System performance and tolerances of 43Gb/s ASK and DPSK modulation formats, in Proc. ECOC 2003, paper Th2.6.3 [3] O. Vassilieva et al, Numerical comparison of NRZ, CS-RZ and IM-DPSK formats at 43Gb/s WDM transmission, in Proc LEOS 2001, pp [4] T. Hoshida et al, Optimal 40Gb/s modulation formats for spectrally efficient long-haul DWDM systems, Journal of Lightwave Technology, vol. 20, 2002, pp [5] H. Kim, A. H. Gnauck, Experimental investigation of the performance limit ation of DPSK systems due to nonlinear phase noise Phot. Techn. Lett. Vol.15-2, 2003, pp [6] A. H. Gnauck et al, 2.5 Tb/s (64x42.7) transmission over 4x0km NZDSF using RZ -DPSK formats and all Raman amplified spans, in Proc. OFC 2002, postdeadline paper FC2 [7] C. Ramussen et al., DWDM 40G transmission over trans-pacific distance (,000 km) using CSRZ-DPSK, enhanced FEC and all-raman amplified 0 km UltraWave fiber span, in Proc. OFC 2003, postdeadline paper PD18 [8] G. Vareille et al., 8370km with 22dB spans ULH transmission of 185x.709 Gb/s RZ-DPSK channels, in Proc. OFC 2003, postdeadline paper PD20 [9] J.X. Cai et al, A DWDM demonstration of 3.73 Tb/s over 11,000km using 373 RZ-DPSK channels at Gb/s, in Proc. OFC 2003, postdeadline paper PD22 [] G. P. Agrawal, Nonlinear fibre optics, second edition, Academic Press [11] J.-C. Antona et al., Nonlinear cumulated phase as a criterion to assess performance of terrestrial WDM systems, in Proc. OFC 2002, paper WX5 [12] Y. Frignac et al, Numerical optimization of pre- and in-line dispersion compensation in dispersion-managed systems at 40 Gbit/s, in Proc. OFC 2002, paper ThFF5 3 spans 1 span 3 spans 1 span pre-dispersion tolerance (ps/nm)

13 TuG1 Electronic Dispersion Compensation for Extended Reach G. S. Kanter, A. K. Samal, and A. Gandhi Santel Networks, Balentine Dr. #350, Newark, CA (phone), (fax), Abstract: We investigate the benefits of using post-detection electronic processing in high-speed optical links. In particular, electronic post-processing is studied in conjunction with other advanced methods, such as optical duobinary encoding and chirped modulation Optical Society of America OCIS codes: ( , , , ) 1. Introduction Electronic dispersion compensation, or more generally electronic post-detection processing (EPP), which is typically based on feed-forward equalizers (FFE) and/or decision feedback equalizers (DFE), are known to increase the inherent tolerance of a receiver to various types of impairments such as chromatic dispersion (CD), polarization mode dispersion (PMD), differential mode dispersion (DMD), and low-pass filtering [1-4]. EPP has no optical insertion loss and it promises to be both compact and inexpensive. Because EPP is a post-detection process, it can be used in addition to optical techniques [5,6]. For example, EPP may be useful for combating chromatic dispersion even in long-haul links, which also require periodic optical methods of dispersion compensation, in order to relax the specifications of the optical methods. EPP may also replace optical methods completely in some cases. For instance, although optical PMD compensation can be very effective [7], it is too bulky and expensive to find applications in current systems. Thus EPP will likely be the first line of defense for PMD troubled links. Other performance enhancing techniques such as chirped modulation, optical duo-binary (ODB) modulation, and forward-error correction (FEC) coding can be used in conjunction with EPP to form more robust and effective solutions. We will analyze the performance of EPP using chirped modulators, ODB modulation, and FEC encoding. Additionally, we will highlight the potential of using EPP with low-cost components, such as an integrated laser/electro-absorption modulator (EAM) designed for use in lower speed systems, in order to allow cost savings in shorter-reach applications. 2. Discussion Mach-Zehnder interferometer (MZI) modulators and electro-absorption modulators can be designed to have various chirp characteristics [8,9]. While chirping will not enhance the dispersion tolerance (amount of variation in dispersion which is tolerable), it can be chosen to partially compensate for the known dispersion of the transmission fiber. Simply replacing an un-chirped modulator with a chirped design can increase the acceptable amount of mean dispersion by more than 25%. We have simulated typical links using modulators of various chirps with and without EPP as shown in Fig 1. The simulation parameters are described in [4]. All the EPP simulations use an eight tap (two taps per symbol) FFE and one tap DFE architecture, with a least-mean square updating algorithm [4]. The benefit of adding EPP, when measured as the increase of positive dispersion allowable for a given power penalty, decreases as the chirping factor is optimized. However, there is still a clear benefit to adding EPP to chirped modulation. ODB is a more complex solution than chirping because pre-coding and filtering is required before the MZI []. Additionally, higher driving voltages are typically needed. However, a very substantial increase in CD tolerance (over a factor of 2) can be expected, as well as other potential benefits such as higher spectral efficiency. This makes ODB interesting for extending the metro-scale distance which can be covered without adding costly optical dispersion compensation, as well as for reducing issues associated with dispersion slope compensation in long-haul systems. The duo-binary data was simulated by placing a 2.5GHz ( th order) electrical Bessel filter after the modulator driver. The simulations were done using a PRBS input sequence without any pre-coder. Such a configuration is acceptable for Q-factor analysis, and even for laboratory measurements of BER, while a pre-coder is needed for transmitting actual data []. Our simulations suggest (see Fig. 1) adding EPP leads to about a 15% reach increase (at the 2dB penalty point), in addition to reducing the back-to-back penalty usually observed when using ODB modulation [11]. Moreover, the EPP also provides much more robustness to the ODB system, which is known to be very sensitive to modulator asymmetries [11].

14 TuG1 Forward error correction (FEC) coding is a technique used to reduce the required signal-to-noise ratio for a given BER. The extra margin supplied by FEC can offset penalties from dispersion. FEC requires overhead bits, thus increasing the effective bit-rate. However, the amount of dispersion typically allowable in an optical link is quadratically related to the inverse bitrate. Thus, the FEC coding gain can be significantly reduced, or even eliminated, depending on the dispersion in the system. We simulated RS(255,239) encoded data and passed it through an impaired channel. The signal was then detected at the optimal sample and threshold points with hard decision decoding. For lower bit error rates, the decoder performance was calculated by the ITU specified intrinsic performance equation [12]. Figure 2 shows simulated performance of uncompensated and EPP compensated links either at Gb/s or with FEC overhead. Using EPP with FEC encoding allows the coding gain to be largely maintained even with significant amounts of added dispersion. In fact, there is no penalty even after 2000ps/nm when EPP and FEC are used together (compared to not using either FEC or EPP at the native data-rate). Additionally, it can be seen that EPP at the native data rate will outperform FEC encoding without EPP after a certain amount of dispersion (>1550 ps/nm). We expect EPP to be less expensive than FEC encoding, as it is a simpler component which is only used on the receive side. Previous work [4] has demonstrated that reasonable performance can be expected from components typically used at 2.5Gb/s rates even at Gb/s when EPP is employed. We expand on that work here by experimentally demonstrating a 50km 9.95Gb/s link using an EAM designed for 2.5Gb/s applications in an un-amplified link. We use an integrated laser/eam (Fujitsu FLD5F14CN) which is interfaced to the high-speed data input pins though a hand-made circuit board. We believe better performance, in particular a reduction of the error floor, could be expected by using a properly designed high-speed interface. Our equalizer is a 4 tap (1 tap per symbol) FFE with a 1 tap DFE employing a LMS algorithm. As can be seen in figure 2, we can achieve a bit error rate less than 1e- after 50km of SMF-28 fiber using a PIN detector specified for 20dBm (typical) receiver sensitivity (BER=1e-) near 16.5 dbm input power. This represents an approximately 3.5dB penalty with respect to our measured performance using a Gb/s MZI modulator in the back-to-back condition. Considering that many commercial Gb/s EAMs specified for intermediate reach have 2dB dispersion penalties near this level of dispersion and that integrated laser/eams typically have poorer intrinsic performance than MZIs (due to, for instance, lower extinction ratios), the penalty for replacing a typical G laser/eam with a 2.5G version followed by EPP is on the order of only 1dB. The possibility also exists for increasing the sensitivity of a receiver by combining more sensitive, low bandwidth receivers with EPP [4]. More sensitive receivers allow for longer un-amplified links and low-bandwidth components allow for cost savings. Figure 3 shows the performance of a Gb/s MZI transmitter with an APD fabricated for 2.5Gb/s applications followed by our EPP equalizer as a function of data-rate. Also shown is a curve which gives a rough estimate of how we might expect the sensitivity to vary with data-rate if we had various receivers with the same bandwidth-to-data-rate ratio as our experimental receiver. The curve assumes that the sensitivity of such receivers is proportional to the number of received photons-per-bit for all data-rates. For instance, at 5Gb/s we expect the required optical power for a given BER to increase by 3dB over that required at 2.5Gb/s. The equalized receiver shows a significant improvement in sensitivity over the constant photons-per-bit model up to about 8Gb/s. We estimate our receiver (mounted on a homemade circuit board) had a bandwidth of about 2GHz. We expect that good performance will be observed beyond Gb/s if a 2.5GHz bandwidth receiver is used. 3. Conclusion Electronic post-detection processing can be used in conjunction with other performance enhancing techniques, such as chirped modulation, ODB modulation, and FEC encoding to improve system performance. Through simulations we show that EPP further improves performance in each of these cases. EPP is particularly important when FEC is employed due to the effectively higher data rate imposed by the FEC overhead. We anticipate that EPP can be used to extend the reach of short, un-amplified links by increasing the sensitivity of the receiver as well as to reduce cost by allowing the use of inexpensive, low-bandwidth transmitters. Our experimental data suggests that appropriately chosen components devised for 2.5Gb/s systems can successfully be used at Gb/s if combined with EPP. REFERENCES [1] F. Buchali et al., Reduction of the chromatic dispersion penalty at Gbit/s by integerated electronic equalizers, OFC 2000 vol 3, (2000). [2] H. Bulow et al., PMD mitigation at Gbit/s using linear and nonlinear integrated electronic equaliser circuits, Electron. Lett. 36, (2000).

15 TuG1 [3] X. Zhao and F.S.Choa, Demonstration of a Gb/s transmissions over a 1.5 km long multimode fiber using equalization techniques, IEEE Photon. Tech. Lett (2002). [4] G. S. Kanter et al., "Electronic equalization for enabling communications at OC-192 rates using OC-48 components," Opt. Express 11, (2003). [5] J. C. Cartledge and R. G. McKay, Performance of Gb/s lightwave systems using an adjustable chirp optical modulator and linear equalization, IEEE Photon Tech. Lett. 4, (1992). [6] H. Bulow, F. Buchali, and G. Thielecke, Electrinically enhanced optical PMD compensation, ECOC 2000, (2000). [7] R. Noe et al., Polarization mode dispersion compensation at, 20, and 40 Gb/s with various optical equalizers, IEEE J. of Lightwave Tech. 7, (1999). [8] A. H. Gnauch et al., Dispersion penalty reduction using an optical modulator with adjustable chirp, IEEE J. Photon. Tech. Lett., (1991). [9] J. Cartledge and B. Christensen, Optimum operating points for electroabsorption modulators in Gb/s transmission systems using nondispersion shifted fiber, IEEE Photon. Tech. Lett. 16, (1998). [] W. Kaiser and W. Rosencranz, Simple precoder for high-speed optical duobinary transmission, J. of Opt. Commun. 22, 741 (2001). [11] A. Royset and D. R. Hjelme, Symmetry requirements for -Gb/s optical duobinary transmitters, IEEE J. of Photon. Tech. Lett., (1998). [12] ITU G.975, Forward error correction for submarine systems, Section 7 (2000). Relative Received Power Penalty (db) Binary MZI, Chirp = 0 Binary MZI, Chirp = Binary EAM 40 km Duobinary BER = 1e-12 OSNR (db) Uncomp at Gb/s With FEC at.67 Gb/s With EPP at Gb/s With EPP + FEC at.67gb/s BER = 1e Distance (km) Chromatic Dispersion (ps/nm) Figure 1: (a) Left - Sensitivity results for various modulators and receivers at a BER of 1E-12. SMF fiber has a dispersion coefficient of 17ps/nm.km. Solid (dashed) lines are with (without) EPP. (b) Right- OSNR (measured in 0.1nm) required for a BER of 1E-9 as a function of dispersion for systems with and without FEC and EPP. -4 G MZI (0 km) G EAM (0 km) 2.5G EAM (50 km) Constant BER = 1e-9 Constant photons/bit BER Received Power (dbm) Received Power (dbm) Data Rate (Gb/s) Figure 2: (a) Left -BER vs. received power for a Gb/s MZI transmitter (diamonds) with no fiber, and an OC-48 specified EAM transmitter after 0km (triangles) and 50km of SMF-28 fiber (squares). (b) Right - Power (dbm) required to reach a 1E-9 BER as a function of bit-rate for the APD/Equalizer combination (diamonds). Also shown is a curve with the same sensitivity at 2.5 Gb/s which assumes a constant number of photons per bit are required to reach 1E-9 at each data rate.

16 TuG2 Electronic Signal Processing for Differential Phase Modulation Formats M. Cavallari, C.R.S. Fludger, P.J. Anslow Nortel Networks, London Rd, Harlow, CM17 9NA, UK Tel: +44-(0) , Fax: +44-(0) , Abstract: We present simulations of novel electronic signal processing techniques that may be used to greatly improve the dispersion tolerance of DBPSK and DQPSK modulation formats by at least a factor of two Optical Society of America OCIS codes: ( ) Fiber optic communications, ( ) Fiber optics sources and detectors. 1. Introduction In recent years, there has been growing interest in the use of electrical signal processing (ESP) techniques for the mitigation of optical penalties, such as chromatic dispersion [1] or PMD [2], as they offer the potential for significant cost savings compared to all-optical approaches. Although this has been a very active research topic, little has been published on the application of these techniques to the D(B/Q)PSK format, which is currently being considered as a competitive alternative to ASK for long-haul links [3] or uncompensated metro systems. In this article, we present simulations of novel ESP techniques that may be used to greatly improve (by a factor of 2 at least) the dispersion tolerance of DBPSK and DQPSK modulation formats. These techniques include a Maximum Likelihood Sequence Estimator (MLSE) using balanced detection, an MLSE using dual input detection (separate inputs from photo-detectors) and an MLSE using joint symbol estimation for DQPSK that makes decisions on I&Q symbols, taking into account any cross-coupling caused, for example, by chromatic dispersion. 2. Electrical signal processing techniques The technique at the heart of this work is the MLSE algorithm [4,5], which can be used in different ways, according to the modulation format. Rather than making a decision based on a single instance using a D-type, the MLSE searches through a whole sequence of bits, selecting the most likely sequence. Mathematically, this is expressed as selecting the sequence (S) which maximises the probability p(x S) of generating the received data (x). For DBPSK, the simplest approach is to use an MLSE with a single input in conjunction with a balanced detector. The A/D converters may operate at or 20 Gsamples/s, providing either one or two samples/bit (for Gbit/s data). The MLSE algorithm maximises the total log-likelihood probability which may be expressed as the sum of probabilities over successive bits (k): (1) max log p x S max log p xk S log p xk S S S k k T T 2 where the term in square brackets may be omitted when using only 1 sample/bit. Rather than using a balanced detector, the MLSE may process data from the constructive and destructive photodetectors (indicated as d1 and d2): (2) max log p x S max log p xk S log p xk S log p xk S log p xk S S S k T T T ; d1 k T ; d 2 k ; d1 k ; d where, once again, the term in square brackets may be omitted when using only 1 sample/bit. In the case of DQPSK the situation is more complex because there are two independent channels and four photodiodes. Two independent MLSEs can be applied to the two outputs of the balanced detectors (Fig.1.a). These would each process the data channel using equation 1. Alternatively, two independent MLSEs can process the outputs from the constructive and destructive photodiodes (Fig.1.b) using equation 2. The last approach examined here uses an MLSE that makes joint decisions on the noisy data from the I channel (x) and from the Q channel (y): (3) max log p x, y S max log p xk, yk S log p xk, yk S S S k k T T 2 where p(x k,y k S) represents the joint probability of receiving noisy sample x k and y k for a sequence of symbols S. This takes into account any cross-coupling or distortion between the I and Q channels.

17 TuG2 Optical filter T T Bessel filter MLSE MLSE I data Q data Bessel filter MLSE MLSE I data Q data Bessel filter Joint Symbol MLSE I data Q data DQPSK Interferometer Balanced Balanced 5 bit A/D detectors 5 bit A/D 1 or 2 sample/bit detectors 1 or 2 sample/bit a b c 5 bit A/D 1or 2 sample/bit Fig.1. Receiver architecture: DQPSK interferometer followed by (a) single input MLSE using balanced detectors, (b) dual input MLSE using separate photodiodes, and (c) joint symbol MLSE using balanced detectors. Here, we use Monte-Carlo simulations to assess the performance of these equalisers and their tolerance to chromatic dispersion. The DQPSK or DBPSK waveforms, consisting of about 0,000 bits are dispersed, noise loaded with ASE using optical amplifiers and then detected using the receivers shown in Fig.1. Non-linearity was not included in these simulations to assess the performance of the equaliser in mitigating the effect of chromatic dispersion alone. For GBaud data, the optical filter has a Gaussian transfer function with a FWHM of 0.11 nm, and the electrical low pass filter is a 5 th order Bessel filter with a 3 db bandwidth of 7.5 GHz. The performance is evaluated in terms of the required OSNR (ROSNR) to achieve an output bit error ratio of -3, which may then be corrected to -15 using strong FEC. Due to simulation limitations, the performance of the FEC is not included in this study. The A/D converters have 5 bits quantisation resolution (32 levels), and the MLSE algorithms use either 5 bit state sequences, or 3 symbol state sequences (for joint symbol MLSE). Note that the complexity of the MLSE scales exponentially with the number of bits or symbols: a 3 symbol joint MLSE has 2 2x3 =64 states, and a 5 symbol joint MLSE has 2 2x5 =24 states! In order to keep the computation complexity tractable, we consider a 3 symbol joint MLSE in the rest of this work. Probability tables are represented by a lookup table and are trained using separate data from that used for measurement. The search for the best match is done with the Viterbi algorithm. Finally, MLSEs can operate on data sampled at the Baud rate or at a multiple of the Baud rate, for higher performance. 3. Numerical results The results for DBPSK and DQPSK with the different receiver types are shown in figures 2 and 3, which report only positive values of chromatic dispersion for simplicity. ASK results are reported as well, as a reference. 20 ASK, D-type DBPSK, dual input MLSE (1S/B) DBPSK, single input MLSE (1S/B) DBPSK, D-type ASK, MLSE (2S/B) DBPSK, dual input MLSE (2S/B) DBPSK, single input MLSE (2S/B) ROSNR for BER=1E-3 (db) D (ps/nm) Fig. 2. Required OSNR for a BER of -3 vs. chromatic dispersion for GBaud DBPSK and ASK data. For DBPSK data (see figure 2), the improvements achieved by the MLSE algorithm depend both on the type of receiver and the number of samples per bit used. The single input MLSE using 1 sample/bit is not particularly

18 TuG2 effective until the ISI spreads the dispersion over a neighbouring timeslot, at about 2000 ps/nm. The 3 db tolerance of 1700 ps/nm when a D-type is used increases by a factor of 2 when a single input MLSE (2 samples/bit) is used, and by a factor of 3 with a dual input MLSE, which is superior to the achievable tolerance with ASK data and a 2 samples/bit MLSE. The phase distortion due to chromatic dispersion creates some independence in the data presented at the constructive and destructive photodiodes and results in incomplete extinction and eye closure when using a balanced detector. The independent data that is lost when using balanced detection, may be used by a dual input MLSE to provide a better chromatic dispersion tolerance. 3 Symbol MLSE (1S/B) dual input MLSE (1S/B) single input MLSE (1S/B) D-type 3 Symbol MLSE (2S/B) dual input MLSE (2S/B) single input MLSE (2S/B) ROSNR for BER=1E-3 (db) D (ps/nm) Fig 3. Required OSNR for a BER of -3 vs. chromatic dispersion for GBaud DQPSK data. It can be seen in figure 3 that for DQPSK, the MLSE using only a single input is not particularly effective in compensating for chromatic dispersion. Since the phase difference between symbols is only half that of the DBPSK, the tolerance for a 3 db penalty is approximately half that of DBPSK when using a D-type at the receiver. For the same number of samples per bit, the best results are obtained with a joint symbol estimator, followed by a dual input MLSE and then by the single input MLSE. This figure also shows that the MLSE with 2 samples/bit always performs better than with 1 sample/bit. Compared to the simplest form of receiver (a D-type flip-flop) a 3 symbol joint MLSE with 2 samples/bit can improve the 3 db chromatic dispersion tolerance by a factor of 3. Even with a simpler version of this device, operating on 1 sample/bit, an improvement of the 3 db tolerance by nearly a factor of 2 can be achieved. This improvement can be explained in the following way. Without any distortion in the transmission path, the data in the I and Q channels of a DQPSK system are independent. Chromatic dispersion spreads the data over their neighbours, and modifies their phase causing coupling between the I and Q channels. A joint symbol estimator makes decisions on both I and Q channels, taking into account any cross-coupling due to chromatic dispersion. Similar calculations performed at 5 GBaud ( Gbit/s) indicate that the 3 db chromatic dispersion tolerance is as high as ps/nm when a 3 joint symbol MLSE is used. This is equivalent to 760 km of NDSF or about 4640 km of NZDSF, in the absence of other system limitations such as non-linearities. 4. Conclusion We have shown that the MLSE algorithm can be successfully applied to D(B/Q)PSK data, for an improvement of chromatic dispersion tolerance by a factor of 3. This performance is obtained with a dual input MLSE in the DBPSK case, and with a joint symbol estimation scheme in the DQPSK case. At Gbit/s, the best results are obtained with DQPSK and a 3 symbol joint MLSE, for a 3 db tolerance of ps/nm. This opens the door to differential PSK links without in-line compensation for both medium and long haul applications. References [1] F.Buchali, H.Bülow et al., Reduction of the Chromatic Dispersion Penalty at Gbit/s by Integrated Electronic Equalisers, OFC2000, paper ThS1. [2] H.F.Haunstein et al., Design of near optimum electrical equalisers for optical transmission in the presence of PMD, OFC2001, Paper WAA4. [3] A.Gnauck, 40-Gb/s RZ-Differential Phase Shift Keyed Transmission, OFC 2003, paper ThE1, and references therein. [4] G.D.Forney, Maximum-Likelihood Sequence Estimations of Digital Sequences in the Presence of Intersymbol Interference, IEEE Transactions on Information Theory, Vol.IT-18, No.3, (May 1972). [5] G.D.Forney, The Viterbi Algorithm, Proceedings of the IEEE, Vol.61, No.3, (March 1973).

19 TuG3 Experimental verification of combined adaptive PMD and GVD compensation in a 40Gb/s transmission using integrated optical FIR-filters and spectrum monitoring Marc Bohn, Werner Rosenkranz Chair for Communications, University of Kiel, Kaiserstr. 2, Kiel, Germany mbo@tf.uni-kiel.de Peter M. Krummrich Siemens AG, Optical Networks, Advanced Technology, Munich, Germany peter.krummrich@siemens.com Folkert Horst, Bert Jan Offrein, Gian Luca Bona IBM Zürich Research Laboratory, 8803 Rüschlikon, Switzerland fho@zurich.ibm.com Abstract: We prove the capability of adaptive PMD compensation with integrated optical FIRfilters in lattice structure and an electrical spectrum monitoring based feedback signal in 40Gb/s experiments. Moreover we present the combined PMD and GVD compensation Optical Society of America OCIS codes: ( ) Planar Waveguides, (060.45) Optical Communications 1. Introduction In current and next generation high speed optical transmission systems solutions for adaptive polarization mode dispersion (PMD) and group velocity dispersion (GVD) compensation are of high interest. Increasing bit rates, path rerouting, temperature changes, power variations and stress induced birefringence lead to GVD and PMD values that exceed the system tolerances. With the increasing complexity of the optical layer the tolerances are even getting smaller. Therefore, an adaptive solution is necessary to meet the tolerances and to compensate for these time varying distortions. The adaptive equalizer should consist of an easy to handle PMD and GVD compensating device and a fast, robust, and easy to generate feedback signal. Integrated optical FIR-filters can compensate for both, GVD [1] and PMD [2]. These devices are well understood and their transfer function can be easily and fast tuned ( t<0µs) by the thermooptic effect. A number of adaptive control criterions have been proposed [3-6]. Because of its capability of detecting both - PMD and GVD - and its speed and ease of implementation, an adaptive feedback strategy of monitoring frequencies within the electrical spectrum of the received signal is chosen. Signal distortions caused by the transmission fiber can be determined very well in the electrical spectrum of the received signal. A fast, inexpensive and easy to implement solution is to monitor frequencies within the electrical spectrum by electrical bandpass filtering. The monitoring of the power of a single frequency component f 0 leads to a power response with well defined minima and maxima for different values of PMD and GVD. Combining the PMD and GVD response we get a 2 dimensional space with a global maximum at zero GVD and PMD. Monitoring two or more frequencies prevent from tracking a local maximum. The adaptive GVD equalization with integrated optical FIR filters in lattice structure and electrical spectrum monitoring as feedback has been demonstrated by our group, [6]. In this paper we report on the experimental adaptive PMD compensation with integrated optical FIR filters in lattice structure and electrical spectrum monitoring as feedback at 40Gb/s and show for the first time the promising results for simultaneous PMD and GVD compensation with this kind of device.

20 TuG3 2. System The setup in Fig.1 was used to demonstrate experimentally the adaptive PMD and GVD compensation with optical FIR-filters and a feedback strategy using the electrical spectrum of the received signal. pattern generator integrated optical 6th order FIR 1 lattice filter (double filter setup ) SSMF, DCF with variable laser EDFA length polarization VOA EDFA BP-filter 3dB Rx BER PMD emulator transmission channel PBS adaptive filter control adaptive feedback signal PD scope BP-filter Fig.1: Experimental system setup: Simultaneous adaptive PMD and GVD compensation at 40Gb/s with integrated optical FIR-filters in lattice structure and electrical spectrum monitoring as feedback signal The transmitter provides a 42.46Gb/s NRZ signal (PRBS sequence= ) at 193.1THz. The signal is distorted by different values of GVD from SSMF and DCF of variable length. A 1 st order PMD emulator sets the desired differential group delay (DGD) value. The compensating devices, the 6 th order optical FIR-filters, are based on a lattice structure, which is implemented by cascading symmetrical and asymmetrical Mach-Zehnder Interferometers (MZI) in a planar lightwave circuit (PLC). The FIR-filters are designed and fabricated by IBM Research using the high index contrast SiON technology [7]. An arbitrary transfer function is achieved by controlling the coupling ratios and the phase differences of the interfering signals. Both orthogonal polarization modes of the transmission signal are separated by a polarization beam splitter (PBS) and each mode is equalized with a separate filter. At the receiver, the electrical signal is bandpass filtered, f FWHM =300MHz, at a center frequency f 0 =20GHz and the detected power is used as feedback signal for the adaptive control and equalization. As we operate in the range between ±25ps, this single frequency is sufficient for the feedback signal. 3. Results and Discussion For adaptive PMD and GVD compensation, the compensating device should be able to compensate for both effects and the feedback signal should provide a unique error signal in the desired range of operation. To verify the adaptive feedback signal for PMD compensation at 40Gb/s, the DGD is varied from 0ps to +35ps and the power of the bandpass filtered (center frequency f 0 =20GHz) received electrical signal is measured, Fig.2a. The feedback signal has a maximum at zero PMD and a minimum at T bit =25ps. For increasing the range with a unique response of the feedback signal, the bandpass center frequency f 0 has to be decreased [6]. a.) measured 20GHz power ( dbm) T bit = 25ps differential group delay ( ps) b.) c.) differential group delay (ps) -log(ber) iteration Fig.2: a.) measured feedback signal at 40Gb/s for varying differential group delay values from 0ps to +35ps; b.) adaptive PMD compensation results: differential group delay values=channel+equalizer, eye pattern; c.) Bit Error Rate measurements

21 TuG3 From a starting point of an approximate differential group delay of 23ps and equal power splitting between the orthogonal modes (γ=0.5) (almost closed eye, no measurable BER), the adaptive control algorithm varies the overall differential group delay (transmission channel+equalizer) slightly to detect the optimization direction. The actual step is compared with the previous one. If the power of the BP-filtered electrical signal is decreasing, the optimization direction is changed and the step size is halved until the optimum value is reached (minimum BER). During the adaptive operation the overall DGD is decreased to 0ps and the eye pattern is very well opened up to an error free transmission, Fig.2b,c. The capability of compensating for both, PMD and GVD, with integrated optical FIR filters in the double filter setup is demonstrated by measurements, Fig.3. The worst case with equal power splitting between the orthogonal modes (γ=0.5) is considered, the filter transfer functions are set without the adaptive feedback signal. a.) -4 back to back GVD=0ps/nm GVD=0ps/nm equalized PMD=25ps equalized GVD=0ps/nm & DGD=25ps/nm equalized b.) GVD=0ps/nm DGD=25ps -5-6 log(ber) DGD=25ps & GVD=0ps/nm DGD=25ps & GVD=0ps/nm equalized input power (dbm) Fig. 3: a.) BER measurement results: back to back (star), GVD=0ps/nm (circle), GVD=0ps/nm equalized (square), DGD=25ps equalized (right triangle), DGD=25ps & GVD=0ps/nm equalized (upper triangle); b.) measured eye pattern: GVD=0ps/nm, DGD=25ps, DGD=25ps & GVD=0ps/nm, DGD=25ps & GVD=0ps/nm equalized At a first step the transmission channel is set to a GVD value of 0ps/nm. While compensating for GVD only, we realize a sensitivity gain of 4.5dB and a sensitivity penalty of less than 1dB in comparison to the back to back case at a BER= -9. Next, a PMD setting of the transmission channel of DGD=25ps is compensated. The initially closed eye pattern is clearly opened and the resulting sensitivity penalty is approximately 1dB. Finally the transmission channel is set to GVD and PMD values of GVD=0ps/nm and DGD=25ps. Equalizing the combination of PMD and GVD, the totally closed eye pattern is clearly opened and the sensitivity penalty is less than 1.5dB. 4. Conclusions The experimental results prove that two integrated optical FIR-filters in the polarization separated setup, where each filter compensates for the distortion of one polarization mode, are very well suitable to compensate for PMD, GVD and combinations of both. Equalizing a distortion of DGD=25ps and GVD=0ps/nm regenerates a totally closed eye pattern to a clearly open one and a sensitivity penalty of less than 1.5dB. A feedback criterion generated by electrical spectrum monitoring in terms of bandpass filtering the received electrical signal makes this filter setup adaptive to compensate for distortions due to PMD and GVD. 5. References [1] M.Bohn et al., Tunable Dispersion Compensation in a 40Gb/s System using a compact FIR lattice filter in SiON technology, European Conference on Optical Communications proceedings (ECOC 2002), Tu [2] M.Bohn et al., Adaptive polarization mode dispersion 40Gb/s with integrated optical FIR-filters, National Fiber Optics Engineers Conference (2002), We D6 [3] Q.Yu et al., Chromatic dispersion monitoring technique using sideband optical filtering and clock phase-shift detection, Optical Fiber Communication Conference (OFC 2002), We 3 [4] P.Westbrook et al., Measurement of residual chromatic dispersion of a 40-Gb/s RZ signal via spectral broadening, IEEE Photonics Technology Letters, Vol.14, No.3, pp (2002) [5] M.Petersen et al., Dispersion monitoring and compensation using a single in-band subcarrier tone, Optical Fiber Communication Conference (OFC 2000), WH4-1 [6] M.Bohn et al., Simultaneous adaptive equalization of group velocity and polarization mode dispersion at 40Gb/s with integrated optical FIR-filters and electrical spectrum monitoring as feedback, European Conference on Optical Communications proceedings (ECOC 2003), Th2 [7] R. Germann et al., "SiON high-refractive-index waveguide and planar lightwave circuits", IBM J. Res. & Dev. vol 47 No 2/3, March 2003

22 TuG4 Electrical PMD equalizer ICs for a 40-Gbit/s transmission M. Nakamura, H. Nosaka, M. Ida, K. Kurishima, and M. Tokumitsu NTT Photonics Laboratories, NTT Corporation, 3-1, Morinosato Wakamiya, Atsugi, Kanagawa, Japan Phone: , Fax.: , nakamura@aecl.ntt.co.jp Abstract: Electrical PMD equalizer ICs for 40-Gbit/s transmission were developed using InP/InGaAs HBT technology. In experiments using a PMD emulator, these ICs exhibit good compensation for DGD of over 20 psec at 40 Gbit/s Optical Society of America OCIS codes: ( , ) 1. Introduction In optical communications systems operating at high-data rates over 40 Gbit/s, signal distortion caused by polarization mode dispersion (PMD) limits transmission distance. PMD is regarded as birefringence with a differential group delay (DGD) caused by dual path propagation in the fiber, and it produces intersymbol interference (ISI), which causes an increase of the bit error rate (BER). As the DGD is statistically distributed, an adaptive equalizer is needed. To remove the system degradation caused by PMD, several optical and electrical equalization techniques have been proposed and demonstrated [1-4]. Electrical compensation has advantages in cost and size over optical compensation. In addition, it can also mitigate chromatic dispersion (CD), which can be handled as ISI by an electrical equalizer. However, the speed of electrical equalization is limited by the speed of the electrical components. Improving the operation data rate requires high-speed ICs that can handle large DGD. In this paper, we describe the design and performance of 40-Gbit/s equalizer ICs using InP/InGaAs HBT technology to reduce PMD-induced degradation. This is the first report of an electrical PMD equalizer for 40-Gbit/s transmissions. 2. FFE and DFE Equalizers A linear equalizer as a feed-forward equalizer (FFE) and a nonlinear equalizer as a decision feedback equalizer (DFE) were developed. It is known that the optimum equalizer structure combines both FFE and DFE components. A FFE can remove both pre- and post-cursor ISI, and a DFE can remove post-cursor PMD without increments of noise. These filters can therefore reduce the ISI in the received signal. The FFE circuit has a transversal filter structure with three taps using two delay stages. It consists of tap delays, mixers, adders, and buffer amplifiers, as shown in Fig. 1(a). Each component with unity gain is used for linear filter operation. Mixer cells adjust the tap weights C i in the range between 1 and 1. The tap delay is given as the difference between delay1 T d1 and delay2 T d2. These delayed and weighted signals are superimposed at the output. In Buffer Delay2 T d1 T d2 Delay1 T d1 T d2 Adder Mixer Buffer Out In CLK Buffer Buffer Delay Subtractor T d1 Mixer + - DEC T d2 DEC Buffer Out Tap1 Tap2 Tap3 Tap (a) (b) Fig. 1 Block diagram of equalizer circuits. (a) FFE circuit, (b) DFE circuit.

23 TuG4 The noise generated at each tap is added together and the accumulated noise degrades eye diagram at output. In addition, the noise is enlarged by increasing bandwidth. To achieve high-speed performance, small numbers of three tap stages are effective in reducing the influence of noise. The tap delay is designed as an optimal value for 40-Gbit/s operations. A differential configuration is used for stable operation at high speed. The DFE circuit is based on a feed-forward structure with one tap, as shown in Fig. 1(b). A conventional DFE circuit has a feedback loop and is difficult to speed up because of the parasitic delay in the loop. To achieve high-speed operation of over 40 Gbit/s, we devised a feed-forward circuit that works as well as a conventional DFE. This circuit has two signal paths. One is the main signal path, and the other is the weighted signal path with a decision. In this circuit, the tap delay is designed to be a period of the data rate. The weighted signal is subtracted from the main signal. Using this circuit, the tap delay is the difference between the two signal paths. It can be set to an arbitrary value, which makes it possible to increase operation speed. The ICs were fabricated using InP/InGaAs HBTs, which have f T of 150 GHz and f max of 240 GHz [5]. The die size of the FFE IC is 3 x 3 mm 2, and that of the DFE IC is 2 x 2 mm 2. For high-speed analog filters, InP-base devices have an advantage in linearity over Si devices due to their high-breakdown voltage, which enables enlargement of the linear operation region. In addition, the circuits are configured using lumped circuits instead of the distributed circuits [6] used in conventional equalizers for high-frequency applications. The lumped components improve the operation of addition and subtraction in the analog filter and reduce the chip size. 3. Experiment and Results The experimental setup for PMD compensation measurement is shown in Fig. 2. A 40-Gbit/s NRZ optical signal was generated by an external modulator driven with pseudo-random-bit-stream (PRBS) signal from a pulse-pattern generator (PPG) and fed to a polarization controller (PC). The polarized signal generated by the PC was fed to a PMD emulator that generates DGD. This PMD-emulated signal was provided to a PD through an erbium-doped fiber amplifier (EDFA). The output of the PD was directly connected to the equalizer ICs. The bit error rate was measured with a 40-Gbit/s BER test set. The performances of the ICs were evaluated using high-speed wafer probes. DGD 1-40 Gbit/s PRBS PPG LD LN Modulator PC PMD Emulator EDFA PD Equalizer IC BER PC: polarization controller, PPG: pulse-pattern generator, BER: bit-error rate detector Fig. 2 Experimental setup for PMD equalization using an emulator. Measurement results of PMD equalization using the FFE IC are shown in Fig. 3. Fig. 3(a) is the input waveform with DGD of 20 psec at the power splitting ratio of around 0.5. The eye is almost completely closed due to the ISI caused by the PMD. The eye diagram equalized using the developed equalizer IC is shown in Fig. 3(b). It is clearly open, when the normalized tap weight is C i = [0.5, 0.5, 1]. Fig. 3(c) shows measured BER characteristics with and without the FFE IC. The IC has immunity from DGD of over 20 psec. Power supply voltage of the FFE IC is 4.3 V, and its power dissipation is 0.82 W. Measured input and output waveforms of DFE IC are shown in Fig. 4, where (a) is the input waveform and (b) the output one. The input data contains DGD of 25 psec at of 0.3. The DFE IC can also mitigate the PMD of over 50 psec at the power splitting ratio of around 0.3. Power supply voltage is 4.5 V, and the power dissipation is 1.3 W.

24 TuG4 0 mv/div Gbit/s NRZ PRBS w/o FFE with FFE = ~ mv/div (a) 5ps/div Log BER = ~0.5 = ~0.34 = ~0.28 (b) 5ps/div DGD [ps] (c) Fig Gbit/s PMD equalization of the FFE IC. (a) Input waveform with DGD of 20 psec at of 0.5, (b) output waveform, (c) BER performance. 0 mv/div 0 mv/div (a) 5ps/div 5ps/div (a) (b) Fig. 4 Measured waveforms of the DFE IC. (a) Input waveform with DGD of 25 psec at of 0.3, (b) output at 40 Gbit/s. 4. Conclusions Electrical PMD equalizer ICs for 40-Gbit/s transmissions were developed, and they exhibit good performance for PMD equalization. To achieve high-speed operation, we designed a FFE circuit with minimized tap stages and optimized delay time and devised a feed-forward DFE circuit. These ICs were fabricated using InP/InGaAs HBTs. The ICs can compensate DGD of over 20 psec for 40-Gbit/s NRZ signal. Theses chips permit adaptive equalization. REFERENCES [1] S. Lanne, et al., Extension of polarization-mode dispersion limit using optical mitigation and phase-shaped binary transmission, ThH3-1, OFC2000. [2] H. Bulow, et al., PMD mitigation at Gbit/s using linear and nonlinear integrated electronic equaliser, IEE Electron. Lett., vol. 36, pp , Jan [3] K. Azadet, et al., Equalization and FEC Techniques for Optical Transceivers, IEEE J. Solid-State Circuits, vol. 37, pp , Mar [4] G. Kanter, et al., Electric equalization for extending the reach of electro-absorption modulator based transponder, ThG6, OFC2003. [5] M. Ida, et al., Undoped-Emitter InP/InGaAs HBTs for High-Speed and Low-Power Applications, Tech. Dig. IEDM2000, pp [6] J. Lee, et al., MMIC Adaptive Transversal Filtering Using Gilbert Cells and is Suitable for High-Speed Lightwave Systems, IEEE Photon. Technol. Lett., vol. 12, pp , Feb

25 TuG5 Mitigation of Nonlinear Impairments from Semiconductor Optical Amplifiers using an Optical Equalizer Ashish Bhardwaj, Christopher R. Doerr, S. Chandrasekhar and Lawrence W. Stulz Bell Laboratories, Lucent Technologies, 791 Holmdel-Keyport Road, Holmdel, NJ Abstract: We demonstrate and study significant improvements in the measured bit-error rates of 40-Gb/s signals distorted after passing through a single and a cascade of two semiconductor optical amplifiers by using a simple colorless optical equalizer Optical Society of America OCIS codes: ( ) Semiconductor optical amplifiers; ( ) Integrated optics devices; ( ) Filters, Interference 1. Introduction Optical equalizers have been proposed and demonstrated recently for the mitigation of inter-symbol interference (ISI) [1] arising from impairments such as chromatic dispersion and polarization mode dispersion. However, these previously studied impairments are linear in nature. In this paper, we investigate the improvement in the bit-error rates (BER) by using the equalizer to mitigate the intra-channel distortions arising from ultra-fast nonlinear gain dynamics from a semiconductor optical amplifier (SOA). It is well known that the nonlinearities in an SOA, primarily inter-band transitions, can cause a significant power penalty to amplitude-shift-keyed signals passing through the SOA, with the penalty worsening the closer the SOA output power is to saturation [2]. We investigate the interesting property that a linear device, the optical equalizer, can significantly reduce this penalty. We also study the improvement caused by the equalizer on nonlinear distortions caused by a cascade of two SOAs. 2. Optical equalizer The optical equalizer consists of two single-mode-connected Mach-Zehnder interferometers (MZIs) each with a relative optical path delay of 20 ps [1]. At each instant of time, the optical equalizer takes a controllable portion of the energy present and adds it at ± 20 ps with a controllable phase. Each MZI is on a separate silica waveguide chip, but both are mounted on a single thermoelectric cooler. The waveguides have a 0.8% index contrast and are on a silicon substrate. The fiber-to-fiber insertion loss for both MZIs in series is 4.2 db, and the polarization-dependent loss is less than 0.5 db. Each MZI has two tunable couplers and a thermo-optic phase shifter in one arm. Each tunable coupler consists of a small MZI with a thermo-optic phase shifter in one arm and a quarter-wavelength length increase in the other arm. Both couplers in each MZI are adjusted to be at the same value for minimum insertion loss, so the coupler drives within an MZI are wired together. The coupler drives control the impulse response satellite amplitudes, and the phase shifters control the satellite phases. 3. Experiment The SOA used in this study was a commercially available one with a fiber-to-fiber small-signal gain of 18 db at 1550 nm at a bias current of 200 ma. Non-return-to-zero (NRZ) data at 40-Gb/s was modulated on the light from an external cavity laser at THz ( nm) with a chirp-free LiNbO 3 modulator and launched into the SOA. The input power to the SOA was varied from 12 dbm to 3 dbm to study the degradation of the BER resulting from the nonlinearity of the SOA. The BER is measured right after the SOA without any equalization, as shown in Figure 1 (a). The BER is also measured after equalization, as shown in Figure 1 (b).

26 TuG5 Pattern Generator 40 Gb/s NRZ PRBS Tunable Laser MOD SOA Pre-amp PD Error Detector Pattern Generator 40 Gb/s NRZ PRBS (a) Tunable Laser MOD SOA Pre-amp Optical Equalizer PD Error Detector (b) Figure 1: Measurement set-up: (a) without and (b) with the optical equalizer. PD: photo-detector. Figure 2 (a) shows the measured BER after transmission through the SOA both with and without equalization. The input power to the SOA is 3 dbm. The SOA penalizes the sensitivity in the BER at the receiver. The equalizer substantially reduces this penalty. For a BER of -9 an improvement of nearly 5 db is observed in the receiver sensitivity for a pseudo-random bit stream (PRBS) of length Figure 2 (b) shows the receiver sensitivity for a BER of -9 both with and without equalization for different input powers to the SOA. The equalizer settings were readjusted for each power level. -Log(BER) Gb/s, NRZ, PRBS Received Power (dbm) (a) unequalized equalized back-to-back unequalized equalized Receiver Sensitivity (dbm) Gb/s, NRZ, PRBS Input Power to SOA (dbm) (b) unequalized equalized back-to-back Figure 2: (a) Measured BER after transmission through one SOA both with and without equalization after the SOA. Input power to the SOA is 3 dbm. Eye diagrams for the unequalized and equalized case are added as inserts. (b) Receiver Sensitivity for a BER of -9 versus input power to the SOA measured with and without equalization at nm. The overshoots from the SOA lead to an increase in the average power of the data-stream, resulting in a power penalty. However, this effect is small compared to the fact that the overshoots on the rising edges can cause a significant enhancement of ISI in the receiver [2]. Ref. [2] showed how an ideal integrate-anddump receiver would be relatively immune to SOA-induced distortions, whereas the usual filtered receiver exhibits significant eye closure from the overshoots.

27 TuG5 The measurements were repeated on a cascade of two SOAs using the same set-up as shown in Figure 1, except that one SOA is replaced with two SOAs in series. The power launched into each SOA was the same and was controlled by variable attenuators placed before each SOA. Figure 3 (b) shows the receiver sensitivity both with and without the equalizer for a BER of -9 for different input powers to a cascade of two SOAs. For input powers higher than 9 dbm into each SOA, the nonlinear distortion after passing through the SOAs is so high that a BER of -9 could not be measured without equalization. Figure 3 (a) shows the quality factor, Q, measured both with and without equalization after the cascade of two SOAs. The improvements caused by the equalizer after one SOA versus two SOAs can be seen by comparing Figures 2 (b) and 3 (b) Gb/s, NRZ, PRBS unequalized equalized Gb/s, NRZ, PRBS unequalized equalized back-to-back Quality Q (db) Receiver Sensitivity (dbm) Input Power to both SOAs (dbm) (a) Input Power to both SOAs (dbm) (b) Figure 3: (a) Quality factor Q versus input power into each SOA measured with and without equalization at nm. (b) Receiver sensitivity for a BER of -9 versus input power into each SOA measured with and without equalization at nm. 4. Conclusions We have shown that an optical equalizer can be used to significantly reduce the power penalty in the measured BER for NRZ data at 40-Gb/s due to the fast nonlinear gain saturation of a SOA. Significant improvements in the receiver sensitivity for a BER of -9 are observed with the use of the equalizer after the SOA. The use of such optical equalizers could enhance the performance of SOAs in a variety of applications as power boosters on the transmitter side [3], in-line amplifiers [4] and pre-amplifiers on the receiver side [5]. 5. References [1] C. R. Doerr, S. Chandrasekhar, P. J. Winzer, L. Stultz, A. R. Chraplyvy, and R. Pafchek, Simple Multi-Channel Optical Equalizer for Mitigating Intersymbol Interference, Post-deadline Paper PD 11, Optical Fiber Communication Conference, OFC 2003, Atlanta, GA. [2] A. A. M. Saleh, and I. M. I. Habbab, Effects of Semiconductor-Optical-Amplifier Nonlinearity on the Performance of High- Speed Intensity-Modulation Lightwave Systems, IEEE Trans. Commun., vol. 38, no. 6, pp , June [3] Y. Kim, H. Jang, Y. Kim, J. Lee, D. Jang, and J. Jeong, Transmission Performance of -Gb/s 1550-nm Transmitters using Semiconductor Optical Amplifiers as Booster Amplifiers, J. Lightwave Technol., vol. 21, no. 2, pp , February [4] J. Jennen, H. de Waardt, and G. Acket, Modeling and Performance Analysis of WDM Transmission Links Employing Semiconductor Optical Amplifiers, J. Lightwave Technol., vol. 19, no. 8, pp , August [5] B. Mikkelsen, C. G. Jorgensen, N. Jensen, T. Durhuus, K. E. Stubkjaer, P. Doussiere, and B. Fernier, High-performance Semiconductor Optical Preamplifier Receiver at Gb/s, IEEE Photon. Technol. Lett., vol. 5, no. 9, pp , September 1993.

28 TuG6 The impact of nonlinearity on electronic dispersion compensation of optical channels Oscar E. Agazzi and Venugopal Gopinathan Broadcom Corporation, Alton Parkway, Irvine, California Abstract: We show that the nonlinear, non-minimum phase nature of the optical channel is largely responsible for the marked performance difference between maximum likelihood sequence estimation (MLSE) and decision-feedback equalizer (DFE) approaches to dispersion compensation Optical Society of America OCIS code: (060.45) Optical communications; ( ) Fiber optics communications 1. Introduction Electrical signal processing techniques have been proposed to compensate impairments of optical channels such as chromatic and polarization mode dispersion (PMD) [1-2]. Advances in integrated circuit technology are now beginning to make the implementation of these techniques feasible at the speeds required in today s applications (Gb/s or higher) [3]. As a result, interest in electronic dispersion compensation is growing in industry, and some products have been announced. However, since this technology has not been proven in real world applications yet, it is important to clarify the potential of the different approaches proposed. Among the techniques proposed are the decision-feedback equalizer (DFE) [4], and maximum likelihood sequence estimation (MLSE)[1,5]. These techniques have already been applied in electrical channels such as voiceband modems, twisted pairs, digital radio, etc. However optical channels have unique properties, such as strong nonlinearities and non-gaussian noise, that require that these techniques be revisited. In this paper we show that MLSE is robust in the presence of nonlinearity. In Section 3, we show that an MLSE receiver based on an 8-state Viterbi decoder with a nonlinear channel estimator can compensate for the linear and nonlinear intersymbol interference (ISI) of up to 5000ps/nm of dispersion. We also show that the DFE is severely hampered not only by the nonlinearity but also by the fact that the nonlinear ISI created by a bit is observed before the bit is actually detected. 2. Model of the optical channel We are interested in intensity-modulated single-mode fiber links without optical dispersion compensation (this could apply, for example, to metro links). However, this analysis is also applicable to residual dispersion in long-haul links that have optical dispersion compensating fibers (DCFs). For sufficiently long fibers, optical amplification is used to overcome fiber loss. Nonlinearity of the optical channel stems predominantly from the photodetector. Current in photodetectors is proportional to the received optical power, which is a quadratic function of the electromagnetic field amplitude in the fiber. Another source of nonlinearity is the intensity dependence of the refractive index of the fiber. This is important in DWDM links, where the optical power is the aggregate of many channels. Although for simplicity we only consider the nonlinearity in the photodetector, our conclusions remain valid when other forms of nonlinearity are also present. It can be shown [6], that the optical channel behaves approximately linearly in spite of the square law behavior of the photodetector when there are multiple modes of propagation and/or when the coherence time of the source is small compared to the bit-interval. These conditions are often satisfied by multimode fibers, but not by single-mode fibers driven by highly coherent sources [7]. The quantity that determines the degree of nonlinearity present in a signal that has suffered chromatic dispersion is the ratio of the coherence time of the source to the bit-period. The coherence time of modern DFB lasers is in the tens of nanoseconds range, which is much larger than the symbol period at rates of Gb/s or higher. Under these conditions, the nonlinearity of the photodetector plays a major role, and should be included in the analysis of electrical equalization techniques to realistically assess their effectiveness. To model the channel we use the formulation of sections II and III-B of [7]. We assume that the transmitted power is P i t a, where k k p t kt a k 0,1 are the transmitted symbols, and p(t) is the transmitted pulse, assumed to be confined to a symbol interval T. As shown in [7], for a coherent source the received optical power can be expressed as

29 TuG6 P t a h t kt a a v t nt o k k n k n k k n (1) where h(t) = c (t) 2 is the impulse response for the received optical power, c (t) is the complex envelope of the electromagnetic field for the received pulse, and functions v m t Re t t mt c c 2 (2) are second order Volterra kernels corresponding to nonlinear interactions between symbols m bit periods apart. The Volterra kernels provide a complete description of the channel nonlinearity. It is interesting to mention that the second term of (1) vanishes if the source is incoherent [6,7]. 3. Simulation Results In the following we assume that the transmitted pulse has an unchirped Gaussian envelope exp(-t 2 /2T 0 2 ) with unit amplitude and T 0 = 36ps, and we compute c (t) solving the dispersion equation (equation (2.4.15) of [8]). Although in this example we consider chromatic dispersion only, our conclusions are still valid if PMD is also present. Figure 1 shows the impulse response and Volterra kernels v 1 (t) and v 2 (t) for the case of a single-mode fiber with a Dl product of 1700ps/nm, where D is the dispersion parameter of the fiber and l is its length. For example, it could correspond to an unchirped link of 0km of standard fiber with D=17ps/nm.km. Figure 2 shows the corresponding eye pattern at the receiver. In this figure the sampling point should be located at about 0ps with a threshold of about 0.4 (this is shown as a cross in the figure). The large sidelobes at 50ps and 150ps are caused by the v 1 (t) kernel. Fig. 1. Fig.2. Three receivers, a non-equalized clock/data recovery (CDR) detector, an 8-tap DFE, and an 8-state MLSE receiver have been simulated using this channel model. Amplified stimulated emission (ASE) noise [9], is added to the received signal. Since this noise has a non-gaussian probability density function (pdf), the thresholds for the CDR and the DFE are adjusted for minimum bit error rate. In the case of the MLSE receiver, the nonlinearities in the channel are captured in the nonlinear channel estimator, that automatically adapts itself to the channel response (1) during actual operation of the link. It also adapts automatically to the pdf of the noise. Figure 3 shows the optical signal to noise ratio (OSNR) required for a constant bit error rate of -6 for the three receivers as a function of the dispersion product Dl (to make this discussion independent of fiber properties, we use the product Dl instead of fiber length l). It can be seen that the CDR reaches its limit at about Dl=1430ps/nm at an OSNR of 20dB. The DFE tolerates an extra 360ps/nm of dispersion at the same OSNR. Finally, the MLSE receiver reaches well over 5000ps/nm under the same conditions.

30 TuG6 Tolerable dispersion at OSNR=20dB CDR DFE MLSE 1430ps/nm 1790ps/nm >5000ps/nm Fig. 3. Required OSNR vs Dispersion The performance limitations of the DFE result from the fact that the signal it creates to cancel the channel ISI is a purely linear combination of previously received bits, therefore it cannot cancel the nonlinear ISI generated by the Volterra kernels of Fig.1. It is interesting to observe that a nonlinear version of the DFE exists [], where the ISI replica can be a nonlinear function of previously received bits. Unfortunately this structure is fundamentally unable to compensate for nonlinearites excited by the bit currently being detected or by future bits (precursor type nonlinear ISI) which are present in (1). As can be seen in Fig. 1, at least part of the nonlinear ISI is of precursor type. On the other hand, the nonlinear channel estimator of an MLSE receiver can accurately model nonlinear ISI regardless of whether it is of precursor or postcursor type. It can also provide accurate modeling of ASE or any other non- Gaussian noise probability density function. 4. Conclusions As a result of its inability to compensate nonlinear ISI, particularly of the precursor type, the DFE suffers from severe performance limitations in moderate to high dispersion single-mode fiber links driven by highly coherent sources. The MLSE receiver is the appropriate electronic dispersion compensation solution for these channels. 5. References [1] J.H.Winters and R.D.Gitlin, Electrical Signal Processing Techniques in Long-Haul Fiber-Optic Systems, IEEE Trans. Commun., Vol.38, No.9, Sept.1990, pp [2] J.H.Winters, R.D.Gitlin, and S.Kasturia, Reducing the Effects of Transmission Impairments in Digital Fiber Optic Systems, IEEE Communications Magazine, June 1993, pp [3] H.Bulow, Electronic Equalization of Transmission Impairments, Proceedings of the 2002 Optical Fiber Communications Conference, March 17-22, 2002, pp [4] B.L.Kasper, Equalization of Multimode Optical Fiber Systems, Bell Syst. Tech. J., Vol.61, No.7, Sept.1982, p [5] H.F.Haunstein, K.Sticht, A.Dittrich, W.Sauer-Greff, and R.Urbansky, Design of near optimum equalizers for optical transmission in the presence of PMD, Proceedings of the Optical Fiber Communication Conference and Exhibit, Vol.3, 2001, pp [6] S.D.Personick, Baseband linearity and equalization in fiber optic digital communication systems, Bell System Technical Journal, Vol.52, No.7, Sept.1973, pp [7] B.E.A.Saleh and M.I.Irshid, Coherence and intersymbol interference in digital fiber optic communication systems, IEEE Journal of Quantum Electronics, Vol. QE-18, No.6, June 1982, pp [8] G.P. Agrawal, Fiber Optic Communication Systems, (John Wiley and Sons, 1997). [9] D.Marcuse, Calculation of Bit-Error Probability for a Lightwave System with Optical Amplifiers and Post-Detection Gaussian Noise, Jour. of Lightwave Technology, Vol.9, No.4, April 1991, pp [] S.Kasturia and J.H.Winters, Techniques for high-speed implementation of nonlinear cancellation, IEEE Journal on Selected Areas in Communications, Vol.9, No.5, June 1991, pp

31 TuH1 Efficient Physical Topologies for Regular WDM Networks Chi (Kyle) Guan and Vincent Chan MIT Laboratory for Information and Decision Systems, Cambridge, MA Abstract: In this paper, we study the design of efficient physical topology of WDM network by building an analytical framework to find optimal connectivity of the regular networks as network sizes and traffic volume scale Optical Society of America OCIS Code: ( ) Networks; ( ) Fiber optics communications. 1. Introduction With the maturity of enabling technologies such as micro-electromechanical system (MEMS), optical cross-connect (OXC) switched wavelength division multiplexing (WDM) networks become attractive transports for both the backbone and metropolitan area networks [1, 2]. This work focuses on finding the efficient topological architectures of such networks. An important question that repeatedly surfaces in the network topology design is: given the locations of the candidate nodes, enabling technologies and traffic demands (logical topologies), how to find the classes of underlying physical topologies that achieve the minimum cost. Most literatures approach this problem by formulating the design processes as cost minimization problems using integer linear or integer nonlinear programming (ILP or INLP) techniques. Since the design processes solely rely on ILP or INLP techniques, they provide limited insight into what constitutes good network architectures. We find analytical results for the various classes of regular topologies so that their behaviors can help us to better understand the network architecture trends as network parameters (e.g. number of nodes) change. We first focus on an important subset of the mesh topology - symmetrically connected regular topologies and study their optimal connectivity for simplified albeit representative all-to-all uniform traffic. Our cost model includes both the fiber cost which is proportional to the number of fibers in the network, and the OXC cost which is a convex function of the number of transit and add/drop lightpaths at the nodes. One important aspect of our work is that we quantify how the required switching resources are related to the network routing strategies, which had been treated as decoupled by most previous researches. We show that the minimum hop distance is more than a parameter for the physical size and optical reach of the lightpaths. It is also a key parameter in dimensioning the network switching resources. By assigning parametric cost structures to network fiber plants and switching resources and analyzing their tradeoffs, we are able to specify the optimal node degrees analytically for interesting classes of good regular topologies as the sizes and traffic volumes of the network scale. 2. Optimal Network Connectivity Under the Uniform Traffic We study the optimal connectivity of the topologies for the uniform traffic using minimum hop routing. We adopt the minimum hop routing because it allows each lightpath will pass through the least number of intermediate nodes and tie up the least amount of switching resources. In order to make the problem analytically tractable, we restrict our investigation on symmetrically connected regular graph topologies. Irregular graph structures will require numerical analysis. We consider two symmetrically connected regular topologies the -nearest neighbor and symmetric Hamilton graph as shown in Fig. 1. We also include Moore Graphs in our analysis, since topologies such as the Shuffle Net and de Brujin Graph have properties that their minimum hop distances are close to the Moore Bound [3]. We proceed to show that the switching resource required at each node is proportional to network size and the minimum hop distance. By assigning the first order cost structures to fiber plants as well as three most representative types of switching fabrics, namely two-dimensional (2D) single-mems, multi-stages and threedimensional (3D) single-mems optical switches, we can express the cost explicitly as functions of the network size, traffic volume units of t wavelengths, routing strategy and switching technologies. We show that with parametric cost ratio between fiber and switching resources, the cost function is convex with respect to node degree and there is an optimal node degree * for a regular network with N nodes. Table 1 shows in closed-form how the network connectivity changes as the networks size and traffic demand scale for different topologies and OXC switching fabrics. We also plot how optimal connectivity and cost scale as network sizes and traffic volumes in Fig.2 to Fig.5. 1

32 TuH1 3. Dynamic Traffic We assume that each node has exactly t wavelengths of traffic to send to every other node in the network. We use static or quasi-static traffic on the fibers and nodes to approximate the mean of the dynamic traffic and dimension the capacities of the fibers and the nodes accordingly. For simplicity of analysis, we also assume that active switching will handle every unit of the pass through and add-drop traffic at nodes. In the more realistic scenarios, fixed routing and wavelength assignment (RWA) algorithm implies a fixed setting of switches along a lightpath. If the traffic does not change, it can be supported without active optical switches. Besides restoration and protection switching, the main advantage of active optical switching lies in its capability to adapt to the fluctuating/stochastic traffic. When the arrivals of traffic exceed the capacities dimensioned, the network will encounter blocking. Thus for a network to achieve a given requirement on blocking probability, the network designers need to dimension the capacity of the fibers (or nodes) with the mean l plus some margin k, where is the estimated variance of the traffic distribution and k 1 is the amount of extra provisioning in addition to the average. To be economical, the low cost switching equipment such as fiber-patched panels can handle the static or the quasi-static portions of the traffic with fixed routing and passive switching, while the more expensive active switching accommodate the fluctuating/stochastic portions. The blocking probability of the network will depend on the detailed traffic statistics (which may not be available a priori) and the policy of reconfiguring (or not) existing connections. In this paper, we will provide some insight into the problem, by providing an upper bound on the worst case blocking probability over the set of all possible stochastic uniform all-to-all traffic distributions. We will show that for any given k the maximum blocking probability is 1/(1+k 2 ). The distribution (with mean l and variance ) as function of k that has the maximum blocking probability is given by, 1 if l l k ; 2 1 k (1) Pr( l) 2 k if l l. 2 1 k k Similarly for any given the blocking probability p, the maximum margin is k (1 p) / p. The resulting distribution as function of p is given by, 1 p p if l l ; (2) p Pr( l) p 1- p if l l. 1 p I.V. Conclusion An important result of our research is that switching technologies have huge impacts on the final architectures. Networks with 3-D switching fabrics have the best scalability. While 2-D architectures do not scale well in the sense that the topologies having the minimum cost are fully connected as the networks and their traffic volume exceed certain sizes. The other important result is that topologies close to Moore Graphs can achieve near optimum cost. A Moore Graph of 0 Nodes with 3-D fabric has optimal normalized node degree * /(N-1) 5%. Table 1. Optimal node degree * as functions of N and t. Topologies/ Switching Fabrics Optimal node degree as the function of N and t -Nearest Symmetric Moore Graphs Neighbors Hamilton Graph (asymptotic) t 1 t ( N 1) N 1 3-D 2 2 Multi- Stages Numerical Numerical 2-D 3 2 at N bt N ct N dt N 5 3 t N ln( N ) t N 2 2 t N 4 ln( N ) 2

33 TuH1 Reference: [1] V.W.S. Chan, K. L. Hall, E. Modiano, and K. A. Rauschenbach, Architectures and Technologies for High-Speed Optical Data Networks, J. Lightwave Technol. vol.16, no.12, pp , Dec [2] A. A. M. Saleh and J. M. Simmons, Architectural Principles of Regional and Metropolitan Access Networks, J. Lightwave Technol. vol. 17, pp , Dec [3] K. Sivarajan, and R. Ramaswami, Lightwave Networks Based on de Bruijn Graphs, IEEE/ACM Trans. Networking, vol.2, no.1, Feb (a) (b) Fig.1 -nearest neighbors (a) and symmetric Hamilton Graph (b) regular topologies with =3 and N=6. Normalized Optimal Node Degree ( */(N-1)) D Moore Graphs. 3D -nearest neighbors. Multi-stage Moore Graphs. Multi-stage -nearest neighbord 2D Moore Graphs 2D -nearest neighbors Normalized Optimal Cost D Moore. 3D -nearest neighbors. Multi-stage Moore. Multi-stage -nearest neigbors. 2D Moore graphs. 2D -nearest neighbors Units of traffic t Units of traffic t Fig.2 (Left) Analytical results of normalized optimal node degree as a function of traffic unit t. The fiber to switching cost ratio / is set at 40. The network size is N=50. Fig.3 (Right) Analytical results of normalized optimal cost as a function of traffic unit t. The fiber to switching cost ratio / is set at 40. The network size is N=50. Swtich Size at Each Node Nearest Neighbors Symmetric Hamilton Graph Hypercube Shufflenet DeBrujin Graph Moore Bound 1 2 Number of Nodes ( =3) Normalized Optimal Node Degree, */(N-1) Nearest Neighbor 2D Symmetric Hamilton2D Moore Bound 2D -Nearest Neighbor Multi-stage Symmetric HamiltonMulti-stage Moore Bound Multi-stage -Nearest Neighbor 3D Symmetric Hamilton3D Moore Bound 3D 1 2 N, No. of the Nodes Fig.4 (Left) Analytical results of switch sizes as functions of networks size N, =3. Fig.5 (Right) Analytical results of normalized optimal node degrees as functions of N with traffic unit t=1. The fiber to switching cost ratio / is set at 40. 3

34 TuH2 Using tunable optical transceivers for reducing the number of ports in WDM/TDM networks Eytan Modiano Lab. for Information and Decision Systems, MIT, Cambridge, MA 02139, USA Randall Berry Dept. of ECE, Northwestern Univ., Evanston, IL 60208, USA Abstract: We consider the benefits of using tunable transceivers for reducing the required number of ports in WDM/TDM optical networks. We show that tunable transceivers can be used to efficiently groom subwavelength traffic and significantly reduce the number of ports compared to the fixed tuned case. We provide a new formulation for this tunable grooming problem and develop algorithms for designing such networks. c 2003 Optical Society of America OCIS codes: ( ) Networks; ( ) Fiber optics and optical communications 1. Introduction In a WDM/TDM optical network each fiber link supports multiple wavelength channels operating at a given bit rate, e.g., 2.5 Gbps (OC-48) and sub-wavelength traffic is time-division multiplexed onto a wavelength. A significant design consideration for such networks is reducing the number of ports required at each node in the network, where a port refers to the combination of optical transceivers and electronic terminal equipment needed to access a wavelength. There has been much interest in the reducing this requirement by efficiently grooming the low rate traffic so that only a subset of the available wavelengths must be electronically processed at any node, while the remaining wavelengths optically bypass the node. Most of the work on grooming has focused on the case where optical transceivers are fixed tuned and so a fixed subset of wavelengths are dropped at a each node; each dropped wavelength requiring an electronic port (e.g. a SONET ADM). The basic traffic grooming problem, as in [1-5], is then to assign a given traffic requirement to wavelengths so that the total number of needed ports is minimized. The general traffic grooming problem has been shown to be NP-complete [1]; however, optimal algorithms have been found for several special cases, and a variety of heuristic algorithms have also been presented. We consider a different approach to designing WDM/TDM networks based on using tunable optical transceivers, where these transceivers can be tuned from TDM time-slot to time-slot. This complements work on reconfigurable WDM networks, where tunable components are used to change the virtual topology in response to traffic variations or for protection purposes. One goal of this work is to highlight a different advantage of tunable components - they can be used to to significantly reduce the required number of ports over an architecture with fixed tuned transceivers, even with a static traffic requirement. This approach requires transceivers to be tunable from time-slot to time-slot. Hence, with time-slots on the order of µs, these devices must be able to tune in sub-µs time. Presently fast-tunable transceivers are much more costly than their fixed tuned counterparts. It is reasonable to expect that as demand for tunable components increases their cost will continue to drop. Also, as we show in this paper, the use of tunable transceivers can reduce the total required amount of hardware in the network (both optical and electronic); this savings may justify their use. 2. Network Model Consider a network with N nodes numbered 1,...,N. On each wavelength in the network, up to g low-rate circuits can be time division multiplexed; the parameter g is referred to as the traffic granularity. A static traffic requirement for the network is given by an N N matrix R =[R i,j ], where R i,j indicates the number of circuits required from node i to node j. For simplicity, we assume that all traffic requirements are symmetric, i.e., R i,j = R j,i for all i, j; this represents the case where all connections are bi-directional. Each node i generates W i = j R i,j/g (fractional) wavelengths of traffic. Hence, to support the traffic requirement, node i must have at least W i optical transceivers. Also for simplicity, we focus in this paper on the case of unidirectional rings; although, many of our results are applicable to general network topologies. Let W min denote the minimum number of (fractional) wavelengths needed to support the given traffic requirement. In a unidirectional ring with symmetric traffic, each symmetric traffic demand R i,j = R j,i uses exactly R i,j circuits around the ring, and so W min = i j R i,j/2g = N i=1 W i/2.

35 TuH2 λ 1 λ w Drop Tunable Optical ADM Add λ 1 λ w Tuning, no wavelength limit Tuning, wavelength limit Fixed tuned Tunable receiver O/E Tunable Laser E/O ports User equipment Fig. 1: An example node with tunable transceiver N Fig. 2: Number of ports vs. N for a ring with uniform demand of r =1 circuits and g =4. Each node in the network is assumed to have a set of tunable transceivers. An example of such a node is shown in Fig. 1; however, many different implementations are possible. As shown, each tunable transceiver consists of a tunable optical ADM, a tunable receiver and a tunable laser. In addition, the node must also be equipped with opticalto-electrical (OE) and electrical-to-optical (EO) converters. The specific implementation of such tunable transceivers are not of interest in this paper. However, we require the nodes to be synchronized at the slot level. Moreover, in a ring network as a slot propagates around the ring it should return to its source on a slot boundary. This can be taken care of by adding extra delay with fiber delay lines or by framing transmissions so that synchronization is maintained. In the following we use the term port to refer to all of the optical and electronic equipment required to receive and transmit on one wavelength. From the above discussion, for symmetric traffic, each node requires at least W i tunable ports. To illustrate the potential advantages of tunablity consider the following simple example of a unidirectional ring with N = 4 nodes, g = 3, and assume there is a uniform demand of one circuit between every pair of nodes, i.e. R i,j =1for all i j. In this case the minimum number of wavelengths, W min =2and there is a total of N(N 1) = 12 circuits that need to be assigned to the wavelength. With g =3, as many as 6 circuits can be assigned to each wavelength; this can be accomplished by assigning both circuits for each duplex connection to the same time-slot. The traffic demand can then be supported by finding an assignment of each duplex connection to one of the g time-slots in the TDM frame, on one of the wavelengths in the ring. Without the possibility of tunable transceivers the assignment of circuits to wavelengths corresponds to the standard traffic grooming problem; for which an optimal solution is given in Table 1 requiring a total of 7 transceivers. However, if nodes are equipped with tunable transceivers, then the number of transceivers can be further reduced. For example, notice in the traffic assignment in Table 1 node 3 only transmits and receives on one wavelength at any time (i.e. λ 2 in slots 1 and 3 and λ 1 in slot 2). Hence if node 3 were equipped with a tunable transceiver, it would only need one transceiver, rather than two. An optimal traffic assignment using tunable transceivers is shown in Table 2; this requires each node to only transmit on one wavelength during each slot and hence each node need be equipped with a single tunable transceiver. Table 1: Optimal traffic assignment for fixed tuned transceivers. λ 1 λ 2 Slot 1 (1-2) (2-3) Slot 2 (1-3) (2-4) Slot 3 (1-4) (3-4) Table 2: Optimal traffic assignment with tunable transceivers. λ 1 λ 2 Slot 1 (1-2) (3-4) Slot 2 (1-3) (2-4) Slot 3 (1-4) (2-3) This example shows that the number of transceivers can be reduced from 7 to 4 by proper slot assignment. In this case the optimal assignment can be found by inspection; however, as we show next, in larger networks this can be a non-trivial combinatorial problem. 3. Minimum tunable transceiver provisioning We consider finding a time-slot assignment that minimizes the number of tunable ports needed for a ring with a given traffic requirement R =[R i,j ] and W W min available wavelengths. We refer to this as the minimum tunable

36 TuH2 port (MTP) problem. A solution to this requires specifying the number of ports at each node and specifying which wavelength each port must be tuned to during each time-slot, as in table 2. This problem can be formulated as an ILP where the objective is to minimize the N i=1 X i, where X i is an integer variable indicating the number of transceivers at node i. The constraints are given by a set of linear relations representing the conditions that (1) the traffic demand is satisfied, (2) no node can transmit or receive on more wavelengths than it has ports, and (3) each time-slot on each wavelength is not used more than once on any link in the ring. As stated next, this problem is in general NP-complete. Theorem 1 The MTP problem with W = W min available wavelengths is NP-complete. This can be shown by transforming the MTP problem into a graph edge-coloring problem [6]. Though the general MTP problem is NP-complete, the optimal solution can be found in several important cases. First we consider the case where there are sufficient available wavelengths so that the wavelength limitation is not binding. Theorem 2 If a network has no wavelength limitations, then each node requires W i tunable ports. Moreover, a time-slot allocation that achieves this can be found in polynomial time. Notice that W i tunable ports is the minimum number of ports required to support node i s traffic. Thus, removing the wavelength limitation allows each node to use the minimum possible number of ports and lets us solve an otherwise NP-complete problem in polynomial time. The proof of this also uses a correspondence with an edge coloring problem, however here the coloring is performed on a bipartite graph representing unidirectional circuits. In a bipartite graph an optimal edge coloring can be found in polynomial time [7]. The above solution applies to arbitrary network topologies (not necessarily rings). When the wavelengths are limited, the above time-slot allocation will no longer be feasible and the circuits must be packed more efficiently onto the available wavelengths. In this case the network topology is important. First we consider a unidirectional ring with a uniform traffic requirement of r circuits between each pair of nodes and a wavelength limit of W min wavelengths. In this case if the number of nodes is even, then each node again need only be equipped with the minimum number of transceivers. Theorem 3 In a ring with uniform traffic, N even, and W min wavelengths, each node requires W i tunable ports. Moreover, an optimal time-slot allocation can be found in polynomial time. For a general traffic requirement, including the case of uniform traffic with N odd, we have developed heuristic algorithms for approximating the solution to the MTP problem. Moreover, these heuristics have bounded approximation error and exhibit good performance. An example of the performance of these algorithms is shown in Fig. 2. This figure shows the number of ports in a ring with g =4and a uniform demand of r =1circuits for different values of N. Three curves are shown. The top curve is a lower bound on the number of ports required in a ring with fixed-tuned transceivers given in [1]. The middle curve is the number of ports needed with tunable transceivers and W min wavelengths. When N is even this is given by Theorem 3; when N is odd, our heuristic algorithms are used. The bottom curve is the number of tunable ports needed without any wavelength restrictions, as in Theorem 2. In the case with tunablity, the number of ports can be reduced by over 40% as compared to the lower-bound from [1] with fixed tuned transceivers. Also note that there is little difference between the case with wavelength limitation and without. Similar performance is attained for other parameters. 4. Conclusions We show that using tunable transceivers can significantly reduce the required hardware in a WDM/TDM network. As the cost and capabilities of optical hardware improve, the ability to trade-off additional complexity in optical hardware for a significant reduction in electronic hardware may become extremely beneficial. References [1] E. Modiano and A. Chiu, Traffic Grooming Algorithms for Minimizing Electronic Multiplexing Costs in Unidirectional SONET/WDM Ring Networks, CISS 98, Princeton, NJ, Feb., Extended version appeared in IEEE J.L.T., Jan., [2] J. Simmons, A. Saleh, Quantifying the benefit of wavelength add-drop in WDM rings with distance-independent and dependent traffic, IEEE JLT, vol. 17, pp , Jan [3] O. Gerstel, P. Lin, and G. Sasaki, Combined WDM and SONET network design, Proc. Infocom, New York, Mar [4] K. Zhu and B. Mukherjee, Traffic grooming in an optical WDM mesh network, IEEE JSAC, January, [5] E. Modiano and P. Lin, Traffic grooming in WDM networks, IEEE Communications Magazine, July, [6] I. Holyar, The NP-completeness of edge-coloring, SIAM J. on Computing, vol., no. 4, , [7] H. N. Gabow, O. Kariv, Algorithms for edge coloring bipartite graphs and multigraphs, SIAM J. on Comp. 11(1982),

37 TuH3 Asymmetric Reconfigurable OADMs for next generation Metro-DWDM networks Valerio Viscardi and Gianpaolo Barozzi Photonics Technology Unit, Cisco Systems, Inc., Via Philips 12, Monza, Italy - Ori Gerstel Advanced planning and Technology, Cisco Systems, Inc., 275 East Tasman Drive, San Jose, California, U.S.A Abstract: Reconfigurable OADMs allow responding to unpredictable traffic demand. We propose ROADM structures with asymmetric flexibility at the add vs. drop section and a metric to evaluate their flexibility and their applications in Metro-DWDM optical networks Optical Society of America OCIS codes: ( ) Networks; (060.18) Couplers, switches, and multiplexers. Introduction Wavelength Division Multiplexing (WDM) technologies has proven to be a successful choice for the deployment of a high speed communication backbone. The optical layer is suitable for carrying different kinds of traffic: from traditional voice services to IP traffic, but its main purpose in current installed networks is to provide physical media connectivity. The next step in the evolution of the optical layer is the capability to provide advanced functionalities, such as automatic wavelength provisioning and optical restoration. It has been shown that transparent pass-through (e.g. by means of OADMs) provides cost benefits when compared to nodes in which the entire WDM is processed electrically [1]. But this advantage, if obtained with current fixed optical Add/Drops, is counter-balanced by limitations in overall network flexibility: a wrong traffic growth estimate or a deviation from the estimated path can put the network in a blocking state and require a complete connections re-shuffle. These limitations can be removed by inserting reconfigurable Optical Network Elements (ONEs) into the network. Network Growth and ROADM Design It is important to understand how optical networks evolve in time in order to effectively design Optical Network Elements (ROADMs or OXCs). To evaluate network flexibility for architectures and technologies that are very different we need first a metric that allows defining and quantifying ONEs flexibility irrespective of the architectural details. A good starting point is Routing Power as defined in [2]. For this analysis we only need the Routing Power R and the Segmented Routing Power on a single port R 1. R is a parameter that expresses the ONE capacity to Add/Drop any wavelength combination at node level. In a similar way R 1 is related to the combination of wavelengths that can be dropped on a given physical port. R is a parameter that is related at network level to node-to-node connectivity, i.e. the capability to reach a node on a given wavelength. In a similar way, R 1 is a measure of port-to-port connectivity, i.e. the capability to reach client equipment (e.g. a SONET ADM or an IP Router) connected to a given physical ONE port on a given wavelength. In order to understand how these two parameters are related to network traffic growth it is useful to consider two possible scenarios: 1. Network growth without churn: new traffic is simply added; 2. Network with churn: new traffic is composed by a re-routing of old connections and by new ones. Case 1 requires the network to maximize the probability to find a free wavelength to interconnect two nodes. This requires that at least one wavelength is free and that that wavelength can be dropped by both the nodes. ROADM architecture has to maximize node-to-node interconnection: therefore it has to be designed in order to maximize R. Case 2 is similar to 1, but since there is wavelength re-routing, it is important to maximize the re-use of installed equipment, without changing its physical interconnection to the WDM layer. In order to do so, the ROADM architecture has to maximize R 1, or port-to-port connectivity. Of course, an ideal ROADM that maximizes both R and R 1 would be the answer to any network scenario, but that would come with high costs. The real push for re-configurability comes from the requirement to adapt the network to unforeseen traffic demands. Since much of the network churn is typically handled in the electrical layer,

38 TuH3 the bandwidth required in the optical layer grows more moderately and predictably especially in Metro-Regional Networks. Therefore, when a new connection is provisioned, it is very unlikely that it will be removed or rerouted (not referring to optical protection). As a result, maximizing the number of wavelengths accessible from a given physical port (R 1 ) is not as important as maximizing the Add/Drop combinations at ONE level (R). Transmit vs. Receive Tunability R and R 1 as defined in [2] are based on the hypothesis that the ROADM drop section and add section have the same flexibility. But this is not always the case. It is possible to design asymmetric architectures that exploit the functionality of a given technology, such as tunable filter (see Fig. 1a) or tunable lasers (see Fig. 1b) without having to use both. These architectures could provide enough overall network flexibility for certain applications. For example, in an access network, where all traffic flows to a hub, one could dedicate a wavelength to every non-hub node and use tunable lasers in the hub to reach the various nodes, without tunability on the drop side (architecture (b) in the Fig.1). Fig.1. Block diagram of: asymmetric ROADM with Drop tunability a); asymmetric ROADM with Add tunability b); ROADM with Add/Drop tunability Fig.2. Block diagram of an asymmetric ROADM; second band filter (i.e. Band B) is optional and required if the A/D capacity is up to 8 channels. Asymmetric ROADM can still be analyzed using the metric presented in [2], but with minor changes. First, we introduce R 1 *, which is a R 1 normalization in order to have R 1 * = 1 for a ROADM able to access any wavelength to/from a given physical port (1), * R1 R1 = log ( N + 1) (1) N log (2) This normalization provides more readable values that simplify architectures comparisons. By definition [2], R 1 < R 2 < R i < R and therefore R 1 can be very small even for a fully reconfigurable OADM. As the latter can drop any

39 TuH3 of the N wavelengths of a DWDM system or none, the maximum number of Add/Drop states on a single physical port is N+1. Second, R and R 1 * are now referred either to the Add section (R ADD, R 1 * ADD ) or to the Drop section (R DROP, R 1 * DROP ): the overall parameters are obtained as an average of the two (2), * * RADD + RDROP * R1 ADD + R1DROP RTOT = R1 TOT = (2) 2 2 With these new derived parameters, it is possible to analyze asymmetric architectures. Flexibility performances of ROADM architectures of Fig. 1 and 2 are shown in Table 1. The table reports also the number of dropped wavelengths (K) over total wavelengths (N) ratio. It is worth noting that Fig. 1a and 1b provides the same overall amount of flexibility, by exploiting flexibility either in the Drop section or in the Add section. The intuition behind this equivalence is that one can dedicate a wavelength to the source (in the 1b case) or to the destination (in the 1a case) and as long as the peer can tune to that wavelength, the solution is fully flexible. Fig. 1c is a composition of the two and shows, of course, better overall performances. This is because it can redistribute how wavelengths are assigned to nodes, whereas solutions 1a and 1b have fixed assignments of wavelengths to sites. Solutions 1a and 1b cannot in fact guarantee full node interconnections if some of the wavelengths are used by intermediate nodes. Solution 1c can overcome this condition, as it provides the capability to shift both the nodes on an unused wavelength. Fig. 2 shows a scalable asymmetric architecture that can grow up to 8 channels (K/N = 0.25) Table1. R and R 1 * values for the architectures shown in Fig. 1 and 2; N=32. R R 1 * R ADD R DROP R TOT R 1 * ADD R 1 * DROP R 1 * TOT K/N Fig. 1a Fig. 1b Fig. 1c Fig Flexibility vs. Remote Reconfigurability It is worth noting that an optical network based on fixed OADM can still provide network flexibility. This flexibility involves manual installation of patch-cords, but is hitless on existing traffic. It is therefore a cost effective way to answer to unpredicted traffic demands, as long as the timescales for such changes are long enough (which is the case today). Remote configurability is on the contrary the capability to reconfigure the network from a remote management station or automatically. One obvious difference is the need to pre-deploy OEOs in the second case in order to exploit the automation. Another difference is related to the relative importance of R and R1. While R1 is important in reconfigurable nodes with pre-deployed OEO, it becomes irrelevant for flexible nodes, since client equipment is manually connected to the port on which the wavelength is add/dropped. Conclusions In this paper we have shown the importance of network traffic growth scenarios in the architecture design of Reconfigurable OADM. The major push for reconfigurable ONEs is the capacity of answer, at node level, to unpredictable traffic demands and this can be obtained maximizing R parameter. R 1 has minor importance due to the low (or zero) churn level of DWDM networks and due to the fact that manual flexibility is typically sufficient. We have also shown a method to analyze and evaluate asymmetric ROADM architectures. Using this metric we have demonstrated the equivalence between Add section flexibility and Drop section flexibility. It should be noted that Drop section flexibility has an advantage due to the relative cost of tunable filters vs. tunable lasers. These asymmetric ROADM structures provide a good trade-off between flexibility requirements and ONE overall cost. References [1] Simmons, J.M.; Saleh, A.A.M, The value of optical bypass in reducing router size in gigabit networks, in ICC '99 (1999 IEEE International Conference on Communications), Volume: 1, (6- June 1999), Page(s): [2] Feuer, M.D.; Al-Salameh, D., Routing power: a metric for reconfigurable wavelength add/drops, in OFC 2002 (Optical Fiber Communication Conference and Exhibit), (17-22 March 2002), Page(s):

40 TuH4 A New Optical Network Architecture that Exploits Joint Time and Wavelength Interleaving Iraj Saniee and Indra Widjaja Bell Labs, Lucent Technologies 600 Mountain Avenue, Murray Hill, NJ {iis, iwidjaja}@research.bell-labs.com Abstract: This paper presents an optical network that provides fractional wavelength services without optical-to-electronic conversion. The architecture emulates fast switching in the passive network core through the use of ultra-fast wavelength tunable lasers at the edge. 1. Introduction End-to-end applications almost always require only a small fraction of the capacity of a wavelength. Optical Circuit Switching (OCS) with wavelength granularity is therefore not economically viable unless accompanied by optical-to-electronic conversion and traffic grooming. While electronic traffic grooming can effectively increase network efficiency, per-unit cost of bandwidth cannot be kept low due to the conversion and the electronic processing in the network. Fractional wavelength switching performed entirely in an optical domain has gained interest due to its potential in lowering the transmission cost. One prominent example is the Optical Burst Switching (OBS) [1]. OBS improves bandwidth efficiency over OCS through statistical multiplexing in an optical domain. However, OBS requires fast switching and header processing at each node, and may also require optical buffering for better performance. Fast switching and optical buffering currently present significant technological challenges. Ultra-fast tunable lasers (UFTLs) that can switch wavelengths in nanoseconds, wavelength-selective crossconnects (WSXCs) and burst-mode receivers are becoming commercially viable. We propose an optical network architecture, Time-domain Wavelength Interleaved Networking (TWIN), that exploits UFTLs to emulate fast switching (micro-second packet length) in the network core, WSXCs to perform self-routing of optical signals (bursts) and burst-mode receivers to synchronize receiver s clock to incoming signal in tens of nanoseconds. This architecture enables cost-effective transfer of information across a fiber link, typically containing tens to hundreds of wavelength each operating at the rate of 2.5 Gbps to 40 Gbps. TWIN improves bandwidth efficiency over OCS while avoiding the technological barriers inherent in OBS. Fast switching in the network core is avoided by rapid tuning of wavelengths at the sources. Tunable lasers and burst-mode receivers at the network edge enable fractional services to applications without optical-toelectronic conversion inside the network. Furthermore, WSXCs enable passive routing of optical signals to intended destinations. Table 1 summarizes the main characteristics of the three network architectures described above. Table 1 Comparison among different optical network architectures. Optical network architecture Bandwidth efficiency Core switching speed requirement Optical buffering in the core Data processing in the core OCS Low Low No No Slow OBS High High Possibly Yes Fast TWIN High Low No No Fast Response to dynamic traffic 2. Network Architecture Figure 1 shows the architecture of TWIN. Each source is equipped with a fast tunable laser and each destination is assigned a unique (set of) wavelength(s). When a source has data to send to a destination, the source tunes its laser to the wavelength assigned to that destination for the duration of the data transmission. Each intermediate node performs self-routing of optical bursts without buffering to the intended destination based solely on the wavelength of the burst. Self-routing is effected through use of

41 TuH4 WSXCs. No label/address lookup processing is needed in forwarding bursts from one node to another, thereby making the network core transparent and simple. One can implement an intermediate node with a passive combiner and a 1xK WSXC [2]. The intermediate nodes are pre-configured so that any incoming optical signal of a given wavelength will be routed to the appropriate destination. One example is to preconfigure the routes that form an optical multipoint-to-point tree for each destination, as shown in Figure 1. Since there is no buffering in the network core, each source needs to perform media-access control for burst transmission. TWIN uses a scheduling approach to arbitrate burst transmissions at tunable lasers and ensure that conflicts (potential burst overlaps) do not occur in the network. The multipoint-to-point routing considerably simplifies scheduling, as conflicts should be observed only at the transmitters and receivers, but not in the network core. One important objective of the scheduler is to maximize the achievable throughput of the network. In the context of a single node, switch scheduling in an input-queued crossbar switch has been investigated extensively in the past (e.g., [3]). However, TWIN requires network scheduling that deals with the entire network consisting of multiple nodes. More importantly, network scheduling also has to explicitly take into account the propagation delays between various nodes. Source Intermediate Destination Figure 1 TWIN architecture consisting of destination-based optical trees. 3. Network Scheduling Network scheduling can be centralized or distributed. A centralized scheduler knows the traffic demand information for each node pair and uses this information to compute the schedules for all node pairs. A centralized scheduler can be effective when the traffic demand information is quasi-static, but may not be able to respond to fast changes in traffic demand, as it needs constant communications to all nodes. A distributed scheduler resides at each destination and only computes the schedules for sources that have burst transmissions to it. In general, a distributed scheduler can better respond to dynamic changes to traffic demand than a centralized one. Centralized scheduling has been proposed in [4]. This paper presents a distributed scheduling protocol. We assume that burst transmissions are contained in repetitive scheduling cycles. Figure 2 Blocking probability versus number of sources as the batch size (number of transmissions per source) varies. In its simplest form, distributed scheduling involves uncoordinated transmissions of bursts among sources. Each source, however, ensures that the bursts it transmits are not scheduled for transmission at the same time. The performance of this simple scheduler can be obtained by means of analysis or simulation (due to space constraint, the analysis cannot be provided in this paper). Figure 2 shows the performance of this scheduler in terms of burst blocking probability with a scheduling cycle of length B =150. Consider a particular tagged destination. Each of the N sources is assumed to transmit d bursts in a cycle to the tagged

42 TuH4 destination. Some of these bursts will be blocked when they collide at the destination. The blocking probability is defined as the ratio of the number of bursts that are blocked to the total number of bursts transmitted by all sources (Nd). When Nd is fixed, we expect that the number of bursts that are blocked increases as N increases, as shown in the figure. When Nd = 150 (or ρ =1.0), observe that the blocking probability is about 35% as N becomes large. This also corresponds to the case when a source is completely uncoordinated. Note that the blocking probability drops to about 5% when Nd = 16 (ρ = 0.11). We now describe an improvement to the distributed scheduler given above when delays in the order of one round-trip time can be tolerated. The protocol relies on request and grant message exchanges to communicate schedules between a source and a destination, learns when conflicts occur, and reassign time slots upon learning a conflict. The basic idea of the protocol is illustrated in Figure 3. In this example, the source initially requests burst transmission at the rate of three bursts per cycle. Upon receiving the request, the destination grants time slots 1, 6 and 9 for the source to use in the subsequent cycles. Upon receiving the grant, the source checks whether it can transmit the bursts at the designated departure times. Here, it is assumed that the source can only transmit bursts on time slots 1 and 9, but not on 6 because of a conflict with another transmission (to a different destination). The destination learns about the conflict when an allocated time slot (which is 6 in this case) does not contain a burst. Upon detection of a conflict, the destination reassigns time slot 8 to the source. It is assumed that time slot 8 does not create another conflict, and thus the source eventually transmits at its desired rate of three bursts per cycle. Source Destination Request 3 time slots Conflict on slot 6 Grant slots 1, 6, 9 Transmits on slots 1, 9 Change slot 6 to 8 Learn conflict on slot 6 Transmits on slots 1, 9 Transmits on slots 1, 8, 9 Time Figure 3 Basic operation of the distributed Figure 4 Requests and grants as function of protocol with learning. time. It is of interest to evaluate the performance of the distributed protocol under dynamic traffic (bandwidth requests change over time). Figure 4 shows the dynamics of the distributed protocol with learning. Notice that the grants track the dynamics of the requests quite well, demonstrating the responsiveness of protocol with dynamic traffic. Due to space limitations, the description of the distributed protocol is necessarily brief. Other performance issues of interest will be presented at the conference. 4. Conclusion We have presented an all-optical network architecture that provides fractional wavelength services by exploiting joint time and wavelength interleaving. The architecture does not rely on fast optical packet switching or buffering in the network core. These functions are emulated at the network edge through the use of ultra-fast wavelength tunable lasers. We have also described a distributed scheduler that can arbitrate bursts in the network and respond to dynamic traffic. 1. M. Yoo and C. Qiao, A novel switching paradigm for buffer-less WDM networks, OFC 1999, paper ThM6, pp , D. M. Marom et al, "Wavelength-selective 1x4 switch for 128 WDM channels at 50 GHz spacing," OFC2002, postdeadline paper FB7, Los Angeles. 3. N. McKeown, The islip scheduling algorithm for input-queue switches, IEEE Trans. on Networking, vol. 7, pp , Apr K. Ross, N. Bambos, K. Kumaran, I. Saniee and I. Widjaja, Scheduling Bursts in Time-Domain Wavelength Interleaved Networks, to appear in IEEE J. on Selected Areas in Communications.

43 TuH5 Blocking Probability of a Centralized Switch with Shared Wavelength Conversion Aradhana Narula-Tam, Mark H. Brady, Steven G. Finn MIT Lincoln Laboratory, 244 Wood Street, Lexington, MA , , arad@ll.mit.edu Philip J. Lin Draper Laboratory, 555 Technology Square, Cambridge, MA Abstract: This paper presents a new blocking probability model for a centralized switch with shared wavelength conversion. The ingress and egress link wavelengths are selectable. A small number of converters achieves the performance of full conversion. c 2003 Optical Society of America OCIS codes: ( ) Networks; (060.45) Optical Communication 1 Introduction Wavelength converters in high speed WDM networks can reduce or eliminate the wavelength continuity requirement. We investigate the number of shared wavelength converters needed to achieve the blocking probability performance of full wavelength conversion. We consider a centralized switch network topology where N access stations are connected via optical fiber links to a hub optical wavelength crossconnect equipped with a set of c shared wavelength converters. Each connection must traverse two fibers, denoted as the ingress fiber to and the egress fiber from the hub crossconnect. Multiple connection requests are generated at each access node and they can be carried on any wavelength. If a wavelength converter is employed, the wavelength used on the egress fiber may be different from that used on the ingress fiber. A new analytical approximation for blocking probability under shared wavelength conversion is presented and verified via simulations. It is shown that a limited number of wavelength converters, far fewer than the switch size, are sufficient to make blocking probability due to wavelength mismatch negligible. It has been shown that a relatively small amount of conversion is sufficient for some network architectures to perform close to the capability of networks with full wavelength conversion[1],[2]. Previous studies assume large networks, hence their techniques are not readily applicable to a single switch network. For example most studies on sparse wavelength conversion assume some optical crossconnect switches are equipped with full wavelength conversion while others have no wavelength conversion capabilities, e.g. [1]. We consider a shared bank of wavelength converters on the single crossconnect switch. Mitra et al. [2] also consider a shared bank of wavelength converters on each crossconnect switch, however, they assume that calls arrive on a random wavelength on the ingress link, whereas in our model the wavelength on both the ingress and egress links is selectable. Several approximations for blocking probability comparing full wavelength conversion to no wavelength conversion have been formulated [3], [4]. In this work we generalize the analysis in [3] to include the benefits of shared wavelength conversion. Our model assumes that both the link load and wavelength utilization statistics of each link are mutually independent. Although the independence assumption is inaccurate for small switches, the error is proportional to 1= [4], where is the number of ports; hence it is appropriate for large switches as we consider here. 2 Network Model In the centralized switch network topology, each connection requires one wavelength on each link. Each physical link carries W wavelengths. Access nodes are equipped with W tunable transmitters and receivers and are capable of generating and and receiving up to W signals. Connection requests arriving at each access node are modeled as a Poisson process. Connections are blocked if they can not be established and do not reenter the system. The wavelength assignment of each connection is fixed for its duration, which is exponentially distributed. Shared wavelength conversion can be implemented by increasing the crossconnect switch size from NW NW to (NW + c) (NW + c) optical switch and adding a bank of c wavelength converters. The shared wavelength converters are connected in feedback fashion so that an arriving signal can be converted and then fed back to the input and switched to the appropriate output. All NW connections share the bank of c converters. Without wavelength conversion, each connection must utilize the same wavelength on both links between the source and destination access nodes. If each source access node knows which wavelengths are available on each link that the connection traverses, the connection can be established without a wavelength converter whenever a common wavelength is free on both links of the path. However, disseminating up-to-date wavelength utilization information requires overhead and complexity. If each access node is only aware of the wavelength utilization on the ingress physical link to the switch, then a wavelength must be selected on the first link without knowing which wavelengths are available on the second link. We investigate the benefits of shared wavelength conversion in both scenarios, i.e., (1) Coordinated wavelength assignment: which assumes the source node is aware of network wavelength assignment on

44 all links of the path and (2)Autonomous wavelength assignment: which assumes the source node only has information about the wavelength assignment on the ingress fiber. 3 Blocking Probability Analysis and Results Call requests arrive at each access node with rate λ in and are destined for another (random) access node. For the cases of limited or no wavelength conversion, the wavelength utilized to establish the connection is randomly selected from the set of wavelengths available on both the ingress and egress fiber links. If there is no single wavelength that is available on both links, but disjoint wavelengths are available on the ingress and egress links and a wavelength converter is available, the wavelength on the ingress and egress links is randomly selected from the set of wavelengths available on each respective link and a converter is used to establish the connection. We also simulated other wavelength assignment strategies such as first fit and found the results were similar to the random wavelength assignment strategy. We consider a crossconnect equipped with a bank of c shared wavelength converters. Thus up to c of the NW connections between the N nodes can use different wavelengths on the ingress and egress fibers. A connection is blocked for one of three reasons: (1) there are no wavelengths available on the ingress link (input blocking), (2) there are no wavelengths available on the egress link (output blocking), or (3) the sets of wavelengths available on the ingress and egress links are disjoint and no wavelength converter is available (limited converter availability). Without loss of generality, let T = 1 be the normalized average duration of each connection. Define λ i as the arrival rate of connections on the i th link of the path. The average load on the i th link of the path is thus L i = λ i T = λ i. Note that the arrival rate on the ingress link is not equal to the arrival rate at each access node, i.e., λ 1 6= λ in, since blocking reduces the offered load on each link. We calculate the reduced load L i using the approximation algorithm developed in [7]. We also make the common approximation that the statistics of the link loads on each link are independent. 3.1 Coordinated Wavelength Assignment In coordinated wavelength assignment, the wavelength utilization of each link is known. The probability that a wavelength converter is needed, P wc is the probability that there is a wavelength available on both the ingress and egress links, however, the same wavelength is not available on both. Let p busyi (k) be the probability that there are k busy wavelengths on the i th link. The probability that a wavelength converter is needed is W 1 P wc = k=1 W 1 l=w k k W l W l p busy1 (k)p busy2 (l) (1) where p busyi (k) = (L i) k =k! W l=0 (L, for i i) l = 1 2. =l! We approximate the bank of c shared wavelength converters as an M=M=c=c queue with offered load weighted by P wc. This allows us to use standard queuing results to approximate the probability that all converters are in use. Specifically, the load entering the bank of converters is the crossconnect load L 1 N multiplied by the probability a connection requires a wavelength converter, i.e., L wc = L 1 NP wc. The probability that a request for a converter will find all c converters busy is given by the Erlang B formula: P conv busy (c) = TuH5 (L wc) c =c! c n=0 (L wc) n =n! : (2) Note that when c = NW, the probability that a request finds all converters busy should be zero although P conv busy (NW ) is non-zero. However, our results indicate that this approximation error is insignificant. The overall network blocking probability is the sum of the blocking probability due to channel limitation and due to lack of wavelength conversion, P B (c) =1 [(1 p busy1 (W ))(1 p busy2 (W ))] + P wc P conv busy (c): (3) We compare our analytical approximations for blocking probability to simulations for a centralized switch with N = 16 nodes and W = 16 wavelengths in Figure 1(a) below as a function of the input load, ρ = λ in T for several values of c. Analytical curves for full wavelength conversion and no wavelength conversion are computed according to [3]. The analytical computations are very close to simulations, but become slightly less accurate at higher loads because of the independence assumption. As expected, blocking probability for a network with shared wavelength conversion increases with load and lies between the blocking probability for a network with full wavelength conversion and that of a network with no wavelength conversion. Also, we see that a small number of wavelength converters can reduce blocking probability to the level achieved with full wavelength conversion. For example, at a load of 0.5, only 8 wavelength converters (rather than 256) are sufficient.

45 TuH5 Coordinated Wavelength Assignment, N=16, W=16 Autonomous Wavelength Assignment, N=16, W= Blocking Probability Anl. No WC Anl. 2 WC Anl. 4 WC Anl. 8 WC Anl. 16 WC Anl. Full WC Sim. No WC Sim. 2 WC Sim. 4 WC Sim. 8 WC Sim. 16 WC Sim. Full WC Blocking Probability Anl. No WC Anl. 16 WC Anl. 32 WC Anl. 48 WC Anl. 64 WC Anl. 80 WC Anl. 96 WC Anl. 112 WC Anl Full WC Sim. No WC Sim. 16 WC Sim. 32 WC Sim. 48 WC Sim. 64 WC Sim. 80 WC Sim. 96 WC Sim. 112 WC Sim. Full WC Input Load (ρ) Input Load (ρ) (a) Coordinated wavelength assignment (b) Autonomous wavelength assignment Fig. 1. Analytical and simulation results for the blocking probability of a centralized switch with various numbers of shared wavelength converters. Shared conversion blocking probabilities are compared to the blocking probability of a switch with no wavelength conversion and full wavelength conversion [3]. 3.2 Autonomous Wavelength Assignment With autonomous wavelength assignment, the wavelength utilization of the second link is unknown at the access node. Thus the access node randomly selects a wavelength from those available on link 1. The probability that a wavelength converter is needed, P wc is the probability that there is at least one wavelength available on links 1 and 2, and the wavelength selected on link 1 is not available on link 2. The probability that a wavelength converter is needed is P wc = W 1 k=0 W 1 l=1 l W p busy1(k)p busy2 (l): (4) Again, the overall network blocking probability is the sum of the blocking probability due to input and output blocking and due to lack of a sufficient number of converters and can be computed by substituting (4) into (3). The analytical approximations for blocking probability are compared to simulations for a centralized switch with N = 16 nodes and W = 16 wavelengths in Figure 1(b) as a function of the input load, ρ = λ in T and the number of converters available. Again the analytical results match the simulations. Note that the blocking probability with full wavelength conversion is equal under coordinated and autonomous wavelength assignment. As expected, the blocking probability without wavelength conversion is greater under autonomous wavelength assignment; hence we see a larger benefit from both shared and full wavelength conversion. However, a larger number of converters are needed to attain the performance of full wavelength conversion. At a load of 0.5, 96 wavelength converters (out of 256) are sufficient. Interestingly, under both coordinated and autonomous wavelength assignment, the number of wavelength converters needed is roughly P B (0)NW. This can be explained by noting that for an M=M=c=c queuing system, the number of servers needed is approximately equal to the load. 4 Conclusions We have developed a model that accurately predicts the blocking probability of a centralized switch network. We have shown that far fewer than NW wavelength converters are needed to attain the benefits of full wavelength conversion both when full wavelength state information is available (coordinated wavelength assignment) and when wavelength state is unavailable (autonomous wavelength assignment). In future work we expand our model to investigate shared wavelength converter requirements in mesh networks. References 1. S. Subramaniam, M. Azizoglu, and A. K. Somani, All-optical networks with sparse wavelength conversion, IEEE Trans. on Net., D. Mitra, C. Nuzman, and I. Saniee, Optical crossconnect with shared wavelength conversion under dynamic loading, in OFC, M. Kovacevic and A. Acampora, Benefits of wavelength translation in all-optical clear-channel networks, IEEE JSAC, R. Barry and P. Humblet, Models of blocking probability in all-optical networks w/ and w/o wavelength changers, IEEE JSAC, R. Ramaswami and K. N. Sivarajan, Optical Networks: A Practical Perspective, Morgan Kaufmann Publishers,Inc., San Francisco, S. Rahman and Dan Blumenthal, Scalable switching in optical networks, Electronic Products, Feb F. Kelly, Blocking probabilities in large circuit switched networks, Advances in Applied Probability, vol. 18, pp , 1986.

46 TuH6 Towards a meshed ultra high speed TDM optical network : concept, OADM architecture and proof of principle H. Rohde, G. Lehmann, and W. Schairer Siemens AG, CT IC 2 Otto Hahn Ring 6, Munich, Germany, Fax: Harald.Rohde@siemens.com Abstract: Meshed ultra-high-speed OTDM networks give rise to a couple of challenges concerning the architecture of network elements. We present an architecture of an OTDM-ADM and its experimental proof of concept Optical Society of America 1. Introduction Optical data communication networks of the future will operate with much higher transfer capacities per fiber than current ones. It is still an open issue wether the increase of capacity will be accomplished by a higher number of wavelengths per fiber or by higher bitrates per wavelength or - most probably - a combination of both. This paper reviews aspects of ultra high speed OTDM networks as part of future networks. Evolution of todays networks tends towards meshed networks. The use of meshed ultra-high-speed OTDM networks implies the need of additional elements in the network nodes to compensate for differential time shifts between different paths through the network. As these timing elements are not necessary in todays ordinary (D)WDM networks they add another layer to the architecture of ADM network nodes in the case of additional use of OTDM techniques. The current paper surveys different ADM techniques together with their qualification in real transmission systems and discusses the necessary enhancements for the use in a meshed network. 2. Realisation of an OTDM-ADM In the scope of this work, three realisations of ADMs have been experimentally revised: In a Semiconductor-Optical-Amplifier Mach-Zehnder-Interferometer (SOA-MZI) differential nonlinear phase shifts on the data signal are induced by control laser pulses. Careful adaption of these phase shifts directs the through channels to one port of the MZI and the dropped channel(s) to the other port of the MZI. A detailled description can be found in [1]. While the drop functionality was satisfying in the revised setup, the add functionality was not given due to improper depletion of the dropped channel in the through signal. Further investigations and optimization of the design of the intergated MZI structure is necessary. A Four Wave Mixing based ADM uses the former effect to extract single bits out of the data stream with a control pulse. With one nonlinear element for the through and the drop channel respectively, ADM functionality can be shown. First results are very promising [2]. The Gain-Transparent Ultrafast-Nonlinear-Interferometer (GT-UNI) also uses nonlinar phase shifts induced by control laser pulses. These phase shifts are translated into polarization rotations. The resulting different polarizations are used to separate through and drop channels. For a detailed description of the principle please see [3,4]. First very auspicious results of the GT-UNI will be discussed in the next chapter. 3. Experimental realization of an OTDM-ADM in a network 75 km SSMF DCF 75 km SSMF DCF 160 Gbit/s Transmitter TX Gbit/s Mux G 160G GT-UNI OTDM-OADM OTDM-ADM OSO 2 x TX Gbit/s RX Gbit/s SOA RX 160 Gbit/s 2 x Figure 1 Experimental setup: 160 Gbit/s transmitter, 2 75 km fiber links, OTDM-ADM and another 2 75 km fiber links.

47 TuH6 Figure 1 presents the experimental setup used for transmission of 160 Gbit/s over a fiber link, OTDM-add-drop functionality and further transmission over another fiber link. A Gbit/s data stream is multiplexed into a 160 Gbit/s signal and sent over two fiber spans consisting of 75 km SSMF and corresponding DCF. The dispersion at the end of the two spans is compensated to 0%, the residual dispersion slope is less than ± 0.8 ps/nm 2. After transmission through this 150 km fiber link the signal is fed into a GT-UNI OTDM-ADM. Here one of the Gbit/s tributaries is coupled out and the empty bit slot is filled with another Gbit/s signal. A SOA-Demultiplexer [2] is used to demultiplex the signal after the GT-UNI to measure the quality of the single channels. After the GT-UNI the signal is retransmitted over two further fiber spanss of about 75 km each, again at the end of the link with 0% dispersion compensation and slope compensation to less than ± 0.8 ps/nm 2. As showed in Figure 2, error free transmission could be achieved for all dropped channels (direclty received with a Gbit/s receiver) as well as for all through channels which are demultiplexed by an FWM based demultiplexer. The retransmitted signal after the second link is recorded by an Optical Sampling Oscilloscope (OSO) and a clear open eye pattern is observed. -4 a log(bit Error Rate) -5 B2B G Drop Through Add Received optical power (dbm) b c Figure 2 Bit error rates of the ADM: Back-to-Back Gbit/s reference, dropped Gbit/s channels, Gbit/s through channels and one Gbit/s add channel. The OSO pictures show: a) input signal into the ADM, B) dropped channel, c) through channels with added channel after transmission over another 150 km; for identification purpose a clock signal has been used as add signal.. 4. Meshed OTDM networks As shown in the previous chapter a GT-UNI is well suited as an ADM node in an OTDM network. A series of concatenated nodes can be linked into a ring structure or more flexible into a meshed network. The system under the scope of this work is a single wavelength Optical Time Domain Multiplexed (OTDM) system with an aggregated data rate of 160 Gbit/s and a tributary data rate of Gbit/s. The smallest meshed network consists of three network nodes, named A,B and C, interconnected as shown in Figure 1. TX 1 A 16x Gbit/s OADM AA Through Add Drop OADM B B Through Add Drop RX A1 TX B 2 OADM C C Through Add Drop Figure 3 Example of a meshed OTDM network with three ADM network nodes. Transmitter TX 1 sends a 160 Gbit/s data stream over a fiber optical link to network node ADM A where one or more of the Gbit/s tributaries are dropped. The undropped channels are transmitted to ADM B while the dropped channels are transmitted to ADM C. In ADM C these channels are added to another data stream collected in transmitter TX 2. Part of the traffic passing ADM C is sent over another fiber link to ADM B. The network element ADM B now has to combine the two data streams; one coming from ADM A and another coming from ADM C. For OTDM network nodes some additional effects in comparison to standard WDM nodes have to be considered: a typical fiber link may consist of 75 km SSMF. With a typical change of the optical length of -7 /K a temperature change of 1 K induces a change of the link length of 6 bits at 160 Gbit/s. Two data streams coming in from different network nodes through two different fiber links will experience different temperature fluctuations on the links and therefore a substantial shift of the two data streams with respect to each other. Even in RX 2 B

48 TuH6 an air conditioned laboratory a shift of several bits per minute was observed at the end of a link of 150 km SSMF. In (D)WDM systems time shifts of the data in one wavelength compared to the data in another wavelength do not effect system performance as each wavelength is received seperately. This situation changes in a high-bitrate OTDM system when the different channels are coming from different sources and are multiplexed into a single data stream. Each channel has to fit very precisely with a tolerance of less than a picosecond into its timeslot. Therefore, a mechanism to detect and to correct phase shifts due to fluctuating lengths of optical fiber links has to be applied. Figure 2 schematically shows the resulting structure of an OTDM-ADM for meshed optical networks. Back to Back OTDM -OADM incoming data OADM A Through Add Drop Data Clock Timeslot Control Monitor out-bound data Variable delay line Control incoming data Figure 4 schematic structure of an OTDM-ADM for a meshed network An OTDM-ADM, like the one described above, is supplemented by a variable delay line (which can be for example a motorized free space delay line or a fiber wound onto a piezoelectric cristal) for one of the incoming data channels. After the ADM a timeslot control electronics permanently monitors the correct timing of all bit slots. This information if fed into a control circuit and used to set the delay line to its optimal position. With the help of that mechanism phase drifts of the different data streams relative to each other are compensated. The timeslot monitor can be realized by any phase detector which works at bitrates of 160 Gbit/s, i.e. cascaded EAMs [5], a nonlinearly driven single EAM [6] or devices based on four wave mixing in SOAs [7] or other nonlinear materials [8]. The phase detector scans the 160 Gbit/s data signal against the GHz base data rate clock. Depending on the resolution of the phase detector a signal with distinct peaks for each single channel is generated which can be used to adjust the delay line of one of the incoming data channels for optimal ADM performance This enhancement of the architecture of an OTDM-ADM node enables the use of these elements in meshed OTDM ultra high speed networks. 5. Conclusion The functionality of an OTDM-ADM network node has been demonstrated. All necessary elements to expand the node for the use in a meshed OTDM network are avaiable and experimentally tested. Therefore, from the technological point of view meshed OTDM networks of the future are ready to be realized. Acknowledgement The work was supported by European Commission under the IST project FASHION (ultra Fast Switching in High-speed OTDM Networks). We would like to thank E. Tangiongga, J. P. Turkiewicz, H. de Waardt from TU Eindhoven for their contribution to the experiments and helpful discussions. References 1 M. Heid et al. 160 Gbit/s demultiplexing based on a monolithically integrated Mach-Zehnder interferometer, ECOC 2001 PD.B H. Rohde et al. All-Optical Add/Drop Multiplexer for High Speed Optical Networks based on Four-Wave Mixing in SOA, ECOC 2003 WE4.P127 3 J.P.Turkiewicz et al., Simultaneous high speed OTDM add-drop multiplexing using GT-UNI switch, Electron. Lett., 39, pp. 795 (2003) 4 C. Schubert et al., Error-free all-optical add-drop multiplexing at 160 Gbit/s, Electron. Lett., 39, pp. 74. (2003) 5 D.T.K. Tong et al., 160Gbit/s clock recovery using electroabsorption modulator based phase locked loop, Electron. Lett. 36, p.1951 (2000) 6 J.P. Turkiewicz et al., 160 Gbit/s clock recovery using a single unidirectional electroabsorption modulator and its applications in the transmission experiments, submitted to OFC Jansen et al, Electron. Lett., OFC 2003, ThO5 8 Y. Fukuchi et al., All-optical time division demultiplexing of 160 Gbit/s signal using cascaded second order nonlinear effect quasi phase matched LiNbO3 waveguide device, Electron. Lett. 39, pp. 768, (2003)

49 TuH7 An Overview of the Network Global Expectation Model Steven K. Korotky Lucent Technologies, 1 Crawfords Corner Road, Holmdel, NJ , USA skk@lucent.com (Invited Paper) Abstract: I describe and illustrate the application of the Network Global Expectation Model, in which expectation values evaluated over the network provide both exact and approximate analytic descriptions of the required quantities and variances of network resources and commensurate costs Optical Society of America OCIS codes: ( , ) fiber optics and communications, networks 1. Introduction Fundamental to the comparison and selection of network architectures and their technological implementations is the total cost of ownership of the network, which includes the expenses for capital equipment, network operation, and network management. While operational and management expenses represent the largest share of the total cost of ownership, capital costs are a considerable and highly visible portion of the initial investment, and so are a very important factor in the choice of architecture and technology. The Network Global Expectation Model provides a versatile means to quickly gauge the network equipment needs and costs using only modest computational resources, and therefore can be of value in achieving the desired performance at minimum capital expense. Here I provide an overview of the concept, formalism, results, and application of this model, which has recently been developed and detailed [1]. In the Network Global Expectation Model, expectation values of network variables and functions of these variables are formally evaluated by averaging over the entire network to establish either exact or approximate analytic relationships between dependent and independent variables. As such the model constitutes a multi-moment description of the required quantities of key network and network element resources and commensurate network costs. This approach naturally and accurately connects the global (network) and local (network element) views of the communication system, and results for a very wide range of network sizes and large number of variations can be computed very fast with useful accuracy. Consequently, the model can be used as a tool to gain insight, evaluate preliminary network designs, establish element feature requirements, determine costs, provide sensitivity analyses, assess scaling performance, make comparisons, define products, and identify application domains. The uncomplicated and transparent accounting of network elements, systems, and costs inherent in the model can also constitute a framework for roadmapping and the cooperative exchange of critical planning information on evolving network needs across the many sectors of the communication business. 2. Costs and Expectation Values As the cost of the network for a specified set of features is considered the metric for comparison of architectures and technologies, the model is constructed from this perspective. The fundamental accounting of network costs may be expressed as C T c i, (1) where C T is the total network cost and c i is the unit cost of the ith component. The costs contributing to this sum may be arranged without approximation into subtotals to reflect categories of costs. As a case of interest we consider link costs and node costs. The summation of costs may subsequently be expressed exactly as expectation values of cost over the members of the sets of the categories, in this case the links and nodes. i C T = L c l + N c n. (2)

50 TuH7 The global expectation values of c l and c n are themselves sums of products of expectation values of the number of network elements and subsystems required for the links and nodes and the costs of these element and subsystems. and c c l n = v c (3a) i i l i = v c. (3b) As the costs of the individual network elements and subsystems are considered known, the objective of the model is to formulate accurate representations of the expectation values of the number of network elements and subsystems. The advantage of this approach is that by giving up detailed knowledge of the placement of each network element and subsystem within the network while retaining an accurate count, an enormous reduction in the required computation resources and time is achieved by avoiding repeatedly solving the time-consuming routing and capacity-deployment optimization problem. 3. Network Definition, Variables, and Relationships To illustrate representative results of the model, following we summarize some of the key input and output network variables and present without derivation formulae that relate the expectations values of these variables. We define a communication network as the combination of a network graph, denoted G, consisting of a set of N nodes {n i } and set of L connecting two-way links, or edges, {l i }, and a network traffic. The network graph may be represented by the symmetric matrix [g] with elements g ij. The pair-wise, two-way communication traffic between terminals located at different nodes may be represented by the symmetric demand matrix [d] with elements d ij with total number of two-way demands D and total ingress/egress traffic T. The primary model input variables are taken to be G (N,L), D, and T together with the demand model. All other variables of interest may be determined from these. For specificity, here we consider individual demands to be carried on an individual channel, eg. wavelength. The traffic demand and channel bit-rate, τ, can be computed exactly as the ratio of the total ingress/egress traffic, T, and total number of two-way network demands, D, terminating at all nodes. We state j j n j τ T/D. (4) The average degree of a node δ is calculated straightforwardly by summing the number of one-way (directed) links and by dividing by the number of nodes yielding δ = 2L/N. (5) The number of hops between a pair of terminals is defined as the minimum number of links traversed by a demand between the terminating node pair. The expectation value of the number of hops is over the set of demands and for minimum hop routing of uniform demand over two-dimensional networks we find that h is accurately approximated by h = [(N - 2)/( δ - 1)] 1/2. (6) The mean number of channels, W o, appearing on a link of the network has been derived and is given by W o = d h / δ, (7) where d is the mean number of one-way demands terminating at a node. This important new result is exact and valid independent of the demand model. Note, however, the value of h is implicitly dependent upon the demand model, as specified above. For uniform unit demand d = N-1. For one estimate of the variance of W o we calculate σ 2 (W o ) W o [1-1/ h ]. (8) Extra capacity for restoration is expressed as a fractional increase, κ, in the total deployed capacity to service the traffic and provide survivability relative to the case without survivability and using minimum hop routing. For a heuristic for shared mesh restoration that limited the additional number of hops of the restoration path relative to the working path, we found that the fractional extra capacity can be estimated by κ = 2/ δ. (9)

51 TuH7 Recently we have established accurate analytic approximations for the fractional extra capacity for path-disjoint restoration without routing restrictions for general, nominally planar, mesh networks [2]. The average number of channels on a link including extra capacity for restoration can be expressed as W κ W o (1 + κ ). () The average traffic carried on a link, β, is the product of the mean number of channels on a link W and the bitrate per demand,τ, i.e. β W τ. (11) The mean number of ports, P κ, required on a cross-connect to service the working and line-side restoration channels present at a node is P κ = d + W o (1 + κ ) δ, (12) which scales as P κ N 3/2 for large N. The average traffic handled by a cross-connect χ, measured in bits/second for example, is computed straightforwardly from the average number of ports P and the traffic demand bit-rate τ: χ P τ. (13) Independent of the demand model, the ratio, ρ', of the number of terminated channels to total (termination + thru) channels present at a node is given by ρ' = 2/[1 + h ]. (14) Considering Eq.'s 6 and 14 we observe that ρ' scales as 2/ N for large N. It is also worth noting that Eq.14 may be inverted to express h as a function of ρ', viz. h = [2/ ρ' - 1]. (15) The mean length of a link may be estimated given the geographic area, A, serviced by the network using the approximation s A /( N 1). (16) 4. Network Costs and Applications Given a cost structure for the network elements, the above relationships can be used to compute costs for categories of the network elements, as well as total network costs. For illustration, examples of rudimentary cost structures, γ, for transmission line systems, electronic cross-connects and optical cross-connects are γ B-s $30/Gbps/km, γ ep $1K/Gbps, and γ op $2.5K/port, respectively. Sample network costs using these forms will be discussed. 5. Summary Here I have presented an overview of the Network Global Expectation Model, which is a comprehensive and structured statistical formalism for estimating the number of network elements, network element characteristics, and costs of communication networks using analytic formulae. The model includes the calculation of both the mean value and variance of all key network quantities and may be applied to a wide range of topologies, architectures, and demand profiles. Currently the general approach has been established, applied to single-tier mesh networks and location-independent demands, and many of the results have been shown to be valid and applicable independent of the demand model. For uniform demands it as also been shown that the number of nodes, the degrees of the network nodes, the total ingress/egress traffic, the geographic extent of the network, and the equipment cost structures are sufficient to estimate the network variables and costs of interest. Either exact or approximate semiempirical functions and closed form expressions for the network variables, which are easily incorporated into software spreadsheet calculators, have been formulated. This methodology and corresponding analytic tool can quickly provide insight and approximate results for preliminary network evaluation, design, and optimization. References 1. S.K. Korotky, "Network global expectation model: A statistical formalism for quickly quantifying network needs and costs," submitted to J. Lightwave Technol., July, ,M. Bhardwaj, L. McCaughan, S.K. Korotky, and I. Saniee, "Global expectation values of shared restoration capacity for general mesh networks," submitted to this conference, September, 2003.

52 TuI2 Low-loss and flat/wide-passband CWDM demultiplexer using silica-based AWG with multi-mode output waveguides S. Kamei, Y. Doi, Y. Hida, Y. Inoue, and S. Suzuki NTT Photonics Laboratories, NTT Corporation 3-1 Morinosato-Wakamiya, Atsugi, Kanagawa, , Japan. K. Okamoto NTT R&D Fellow, NTT Corporation 3-1 Morinosato-Wakamiya, Atsugi, Kanagawa, , Japan. Abstract: We demonstrated compact and cost-effective 8-channel filters for unidirectional and bidirectional CWDM systems with a low loss of 1.7 db, a wide 1-dB bandwidth of 14 nm, and low crosstalk by using an AWG with multi-mode waveguides Optical society of America OCIS codes: ( ) Integrated optics devices, ( ) waveguides, planar 1. Introduction The recent rapid diversification of services based on IP technologies will move wavelength division multiplexing (WDM) systems beyond backbone and metropolitan area use into access networks. For access networks, coarse WDM (CWDM) with a large channel spacing of 20 nm is now available with a view to realizing cost-effective systems, where the wavelength tolerance of the laser transmitter is relaxed and no optical amplifier is used [1]. This wide tolerance and amplifier-free configuration requires a multi/demultiplexing filter with both low loss and a wide passband. Moreover, the filter must be inexpensive and compact, because the filter to total system cost ratio is greater in CWDM systems that have a relatively small number of channels. There are several types of CWDM filter. Thin film filters (TFFs) have a good spectral response with a low loss and a rectangular shape. However, it is difficult to reduce the production cost, because TFFs require complicated fiber handling. In contrast, planar waveguide filters including the lattice-form filter [2] and arrayed-waveguide grating (AWG) [3,4] are superior as regards cost-effectiveness thanks to their mass-producibility and simplicity of fiber assembly. In particular, the AWGs, which have already been installed in practical dense WDM systems, also have advantages of compactness and the capacity for increasing the channel number. However, the optical performance of conventional AWGs is somewhat inferior to that of the other filters because of the trade-off relationship between their loss and bandwidth. To overcome this disadvantage, an AWG with multi-mode waveguides (MM-AWG) has been proposed [5]. Although its function is premised on the use of a demultiplexer with receivers through multi-mode fibers (MMFs), the MM-AWG can realize both low loss and a wide spectral response while maintaining the advantages of conventional AWGs. Therefore this filter is one of the most attractive candidates for use with CWDM demultiplexers. However, as yet only some basic studies have been undertaken on MM-AWGs tuned for CWDM. Here, we describe two types of 20-nm-spacing 8-channel MM-AWG filter for CWDM that employ a silicabased planar lightwave circuit (PLC). One is a simple demultiplexer for unidirectional transmission and the other is a multi/demultiplexer for mono-fiber bidirectional transmission. We designed the AWGs to be compact by using high index contrast waveguides, and carefully avoided any crosstalk degradation by optimizing the layout of the AWG components. Consequently, we achieved both excellent optical performance and a compact size for these AWGs, which are suitable characteristics for a practical CWDM filter. 2. Design and fabrication The schematic configuration of the MM-AWG and an enlarged representation of the output side are shown in Fig. 1(a) and (b), respectively. The MM-AWG has a configuration in which the single-mode output waveguides found in a conventional AWG are replaced with relatively wide multi-mode waveguides. In the MM-AWG, the focusing field on the imaging plane traverses the plane with a lateral displacement that depends on the input light wavelength. Then the spectral response of the AWG is determined by the coupling between the field and the eigenmodes of the output multi-mode waveguide. This coupling is lossless as long as the position of the focusing field is within the range of the multi-mode waveguide width, although the power distribution to these modes varies depending on the position. Therefore we obtain a rectangular spectral shape. Here, we optimized the width of the multi-mode waveguides so that they would couple efficiently with 50- m-diameter graded-index (GI50) MMFs and set the lateral displacement of the AWG so that the 1-dB bandwidths would be > 14 nm.

53 TuI2 (a) Arrayed waveguides Input single-mode waveguide Output multi-mode waveguides Multi-mode fibers (b) Arrayed waveguides Slab waveguides Output multi-mode waveguides Multi-mode fibers Single-mode fiber Slab waveguides Stray light Enlargement: Fig1(b) Focusing field Imaging plane Fig. 1. (a) Schematic configuration of MM-AWG and (b) enlarged representation of output side. Although the MM-AWG provides the desired rectangular spectral shape, there is a problem associated with the multi-mode waveguides, which is the crosstalk caused by the re-coupling with the output MMFs of faint stray lights that are mainly generated as a result of the coupling loss between the arrayed waveguides and the second slab waveguide. To avoid this crosstalk problem we carefully optimized the layout of the AWG and output multi-mode waveguides. Since we found that the stray lights mainly propagate along the elongated part of the second slab as also shown in Fig. 1(a), we purposely arranged the multi-mode waveguides to bend so that their connecting edges with the MMFs were not positioned on that elongated part. We designed 20-nm-spacing 8-channel MM-AWG filters for unidirectional and bidirectional CWDM systems. Figure 2(a) and (b) show schematic views of the two types of 8-channel CWDM system. In the unidirectional system, the MM-AWG is employed as a demultiplexer whose outputs are all multi-mode waveguides connected to receivers through MMFs. The bidirectional system is expected to be more economical and more suitable for access networks than the unidirectional system because the upstream and downstream channels are transmitted together through one fiber. Here, the MM-AWG is applicable to each end of the system as a multi/demultiplexer. The AWG has a 2 x 9-port configuration where the odd output ports are single-mode waveguides for multiplexing, and the even output ports are multi-mode waveguides for demultiplexing. As shown in Fig. 2(b), we can use the same MM-AWG for both ends of the system by choosing one of two input ports. (a) Multiplexer MM-AWG Tx 1 Tx 2 Tx 3 Tx 4 1 Tx SMF for transmission 5 Tx 6 Tx 7 Tx 8 Single-mode waveguide SMF Multi-mode waveguides Rx 1 Rx 2 Rx 3 Rx 4 Rx 5 Rx 6 Rx 7 Rx 8 MMF Tx: Transmitter Rx: Receiver (b) 1 Tx 2 Rx 3 Tx 4 Rx 5 Tx 6 Rx 7 Tx 8 Rx SMF MMF MM-AWG SMF for transmission 1 2 Multi-mode waveguides Single-mode waveguides Fig. 2. Schematic views of (a) unidirectional and (b) bidirectional CWDM systems We fabricated the two types of AWG using conventional silica-based PLC technology [6]. For compactness we employed waveguides with a high index difference of 1.5% that we recently developed [7]. The respective core sizes and minimum-bending radii were ~ 4.5 x 4.5 m 2 and 2 mm for the single-mode waveguide, and ~ 20.0 x 4.5 m 2 and 5 mm for the multi-mode waveguide. The AWGs were fabricated with vertically tapered waveguides [8] to reduce the coupling loss between the arrayed and slab waveguides. The chips were 8.0 x 40.0 mm 2 for both MM- AWGs and attached to single-mode optical fibers (SMFs) and GI50-MMFs. 3. Experimental results We first describe the optical performance of our unidirectional MM-AWG. Figure 3 shows the 8-channel transmission spectra. We successfully achieved a low loss, and a flat and wide passband for every channel. The insertion losses were < 1.7 db and the 1-dB bandwidths were > 14.0 nm. These excellent rectangular shapes confirm the suitability of our spectral optimization. We also obtained steep suppression slopes at the channel boundaries and a low background level, which was due to our layout optimization of the AWG and multi-mode output waveguides. The crosstalk was suppressed to < -29 db for adjacent channels and < -35 db for the background level. We also evaluated the power penalty associated with the fluctuation in the power distribution between the eigenmodes of the multi-mode waveguides and MMFs caused by the wavelength drift of a transmitter. Figure 4 shows the experimental setup and the obtained bit error rate characteristics. We used a single transmitter with a 2.5 Gb/s modulation. The signal was demultiplexed in the MM-AWG to port no. 5 and guided to a receiver thorough a 3-m-long GI50-MMF. We operated the transmitter, assuming its wavelength drift, at wavelengths of 1550 and 1545 nm, which correspond to the respective wavelengths of the center and edge of the AWG passband at port no. 5. We Rx Tx Rx Tx Rx Tx Rx Tx

54 TuI2 obtained error free operation and no power penalty can be seen in the figure. Consequently, we confirm that the application of the MM-AWG to a CWDM system poses no problems in relation to wavelength drift. 0 Transmittance [db] =1550 nm 1545 nm Wavelength [nm] Fig channel transmission spectra of fabricated unidirectional MM-AWG Tx SMF 2.5 Gb/s PRBS Modulator Back to back Optical received power [dbm] Fig. 4. Experimental setup for testing fabricated unidirectional MM-AWG and obtained bit error rate characteristics We next describe the results we obtained for the bidirectional MM-AWG. Figure 5 shows the 8-channel transmission spectra at input port no. 1. We also obtained good optical performance for this AWG. The insertion losses and 1-dB bandwidths were < 1.6 db and > 13.5 nm for demultiplexing multi-mode ports, and < 4.3 db and > 5 nm for multiplexing single-mode ports, respectively. The crosstalk between adjacent demultiplexing ports was < - 35 db and that between adjacent multiplexing ports was < -40 db. 0 MM-AWG MMF 3 m Power meter Rx Error detector Bit error rate -3 Back to back 1550 nm nm Transmittance [db] Wavelength [nm] Fig channel transmission spectra of fabricated bidirectional MM-AWG 5. Conclusion We used silica-based PLC fabrication technology to develop two types of 20-nm-spacing 8-channel MM-AWG filter for unidirectional and bidirectional CWDM systems. We demonstrated their advantages of low loss, and a flat and wide passband resulting from the response of the multi-mode waveguides, compactness achieved by using high index contrast waveguides, and low crosstalk obtained by optimizing the circuit layout. Along with the costeffectiveness of the PLC filters, this performance confirms that these MM-AWGs are suitable and attractive filters for general application to CWDM systems. We believe that these MM-AWG filters will be key devices in future optical access networks. References [1] P. P. Iannoue, et al., WDM access networks, in Proc. of ECOC2002, paper (2002). [2] Y. Inoue, et al., Low-crosstalk 4-channel coarse WDM filter using silica-based planar-lightwave circuit, in Proc. of OFC2002, paper TuK6, pp (2002). [3] K. Okamoto, et al., Recent progress of integrated optics planar lightwave circuits, Opt. Quantum Electron., 31, (1999). [4] R. Adar, et al., Broad-band array multiplexers made with silica waveguides on silicon, IEEE J. Lightwave Technol., 11, (1993). [5] M. R. Amersfoort, et al., Phased-array wavelength demultiplexer with flattened wavelength response, Electron. Lett., 30, (1994). [6] M. Kawachi, et al., Silica waveguides on silicon and their application to integrated-optic components, Opt. Quantum Electron., 22, (1990). [7] Y. Hibino, Recent advances in high-density and large-scale AWG multi/demultiplexers with higher index-contrast silica-based PLCs, IEEE J. Selected Topics Quantum Electron., 8, (2002). [8] A. Sugita, et al., Very low insertion loss arrayed-waveguide grating with vertically tapered waveguides, IEEE Photon. Technol. Lett., 12, , (2000).

55 TuI3 Spectral engineering of wavelength-division multiplexers based on planar holographic Bragg reflectors C. Greiner, D. Iazikov and T. W. Mossberg LightSmyth Technologies, Inc., 860 W. Park St. Ste 250, Eugene, OR Phone: (541) , FAX: (541) , Abstract: We report wavelength-division-multiplexing based on lithographically-fabricated slab-waveguidecontained planar holographic Bragg reflectors. Partial diffractive contour writing and contour displacement are successfully demonstrated to enable precise spectral engineering of multiplexer transfer functions and make possible hologram overlay Optical Society of America OCIS codes: ( , , , , ) Planar holographic Bragg reflectors (HBRs) [1-3] are two-dimensional lithographically-scribed volume holograms contained within a planar slab waveguide. In the slab waveguide, optical signals are free to propagate without constraints in two dimensions a geometry that enables 2D Bragg structures to provide powerful spectral and spatial holographic functions. A single HBR can simultaneously spatially image an input signal to an output port (or from one point within an integrated photonic circuit to another) while at the same time providing spectral filtering of the signal. Unlike fiber and channel-waveguide gratings, where separation of the counter-propagating input and output signals typically requires additional elements, planar HBRs provide spatially distinct and thus easily accessed outputs. HBRs constitute the building blocks of unique integrated photonic circuits that operate entirely without wire-analog channel waveguides, being based on HBR-mediated signal transport where signals freely overlap as they are imaged from active element to active element. The HBR approach marries the power of free-space optics and volume holography with a fully integrated environment. The powerful volume-holographic function provided by HBR structures provides, via computer-generated complex-shaped diffractive contours, fully-optimized spatial mapping of an arbitrary complex input field distribution to an entirely different output field mode. This holographic imaging function is generally more powerful than that provided by simple confocal elliptical DBRs [4], whose focusing power degrades when input and output optics deviate from the point source limit. Recently [3], we demonstrated that photolithographic fabrication of HBRs in silica-on-silicon provides for the highly accurate placement of constituent diffractive contours thus enabling fully coherent centimeter-scale planar holographic structures. Additionally, a robust and fabrication-friendly method to control the reflective amplitude of diffractive element contours via partial writing of the latter was presented [3]. Together, the precise photolithographic feature placement and the partial contour writing approach provide control over the phase and amplitude of diffractive elements on an individual line basis. This precision control not only offers a path to unprecedented flexibility in the design of HBR spectral transfer functions but also enables the overlay of several (a) (b) y Diffractive Element x z x z Upper Cladding, n clad y Output Input Core, n core d w Lower Cladding, n clad clad OUT IN Figure 1. Planar holographic Bragg reflector schematic. 1a, cross-sectional View; 1b, top View. 1

56 TuI3 (partially written) planar holograms on the same substrate highly enabling to the designing of (de)multiplexing devices for multi-wavelength signals. In the present paper, we demonstrate, for the first time, the application of these methods to the spectral and spatial engineering of multiplexing devices based on holographic Bragg reflectors. The operational principle of an individual planar hologram is illustrated in Figure 1. Figure 1a is a schematic cross section of the device. It consists of a silica slab waveguide with a single-mode core and bilateral 15- m-thick cladding layers. The core-cladding index differential is 0.8%. The cross section also depicts representative lithographically-scribed diffractive elements located at the upper core-cladding interface. The diffractive elements, with depth d 400 nm, are filled with cladding material. The device operates in first order with a diffractive element period, 500 nm, i.e. half the in-medium wavelength of resonant light. In the schematic cross-section of Figure 1a light enters the device from the left side and is coherently backscattered to the left by the diffractive elements. Figure 1b is a top view of the simple planar hologram. An optical signal is coupled into the device via an input port, IN, typically comprised of the endpoint of a channel waveguide that is used to connect the planar waveguide chip to access optical fibers. The input beam expands in the slab region and is subsequently spectrally filtered and spatially directed to the output port, OUT, by the planar grating. The HBR consists of diffractive contours, represented by the dashed lines in Figure 1b, that may be designed individually to match the back-diffracted input field to the output port. The detailed spacing and relative amplitude of the diffractive elements as a function of position along the input direction determine the spectral transfer function of the device. Fully optimized holographic contours, not implemented here, will improve input-output coupling through optimization of spatial wavefront transformation and are equally consistent with overlay and apodization effects. All devices discussed here with HBR and access waveguides occupy die areas of about 1.7 cm 2. Multiplexer fabrication employed standard photolithography using a DUV optical stepper. In Fig 2, we present results on a 4-channel 0-GHz channel-spacing HBR-based wavelength-division multiplexer with flat-top channel passbands. The device (Fig. 2a) comprises apodized individual-channel HBRs that are staggered along the device depth dimension but are heavily overlapping as well. Fig 2b, upper (lower) graph, depicts the measured (simulated) spectral transfer functions of the various multiplexer channels for TE-polarized input light. The passbands clearly show the designed flat passband and channel spacing. The long-wavelength shoulder on the measured passbands is believed to arise from a second-order apodization effect unaccounted for in the present design comprising an effective refractive index variation concomitant to amplitude apodization. Incorporation of this effect in design algorithms is straightforward. The multiplexer s adjacent channel isolation exceeds -22 db. This is excellent for a first iteration design. Overall, the results shown in Figure 2a clearly (a) 0 (b) In Relative Insertion Loss (db) Wavelength (um) Figure 2. 4-channel HBR-based multiplexer. 2a, device schematic; 2b, upper (lower) graph, measured (simulated) multiplexer spectral transfer function. 2

57 TuI3 demonstrate the feasibility of spectral passband engineering as described and the ability to overlay HBRs. The absolute multiplexer insertion loss (IL) through coupled fibers was ~6 db implying a ~5.0 db device intrinsic IL. As constructed, the HBRs operate at low reflectivity. Calculations indicate that achievable alterations in diffractive structure geometry and refractive-index contrast will provide strong reflectivity and thus low IL loss up to channel counts of In Figure 3, we demonstrate demultiplexing of 8 wavelength channels with 0-GHz channel-spacing using an HBR multiplexer consisting of eight stacked, apodized planar holograms. Similar to the device of Figure 2, the multiplexer comprises a common input port and eight wavelength-specific outputs. In Figure 3, upper (lower) graph, we show the measured (simulated) multiplexer spectral transfer function for TE-polarized input light. Agreement between measured and designed performance is good. The average insertion loss was measured to be ~4 db or about 3.0 db device intrinsic. The bandpass profiles are slightly broader than the weak-reflection-limit simulation, consistent with the measured 50 percent reflectivity. Long-wavelength sidelobes again are traced to uncompensated second-order apodization effects as mentioned above. In summary, we have demonstrated the viability of planar holographic Bragg reflectors as powerful building blocks for wavelength-division multiplexers. For the first time, spectral engineering of multiplexer bandpass functions via diffractive contour design was demonstrated. More generally, the powerful spectral and spatial beam control inherent to the planar volume-holographic approach offers the possibility of channel-waveguide-free integrated photonic circuits wherein signal routing and processing occurs entirely through interaction with distributed diffractive structures like the HBR. Furthermore, as planar surface-relief structures, HBRs promise consistency with low-cost, mass-production, nanoreplication techniques such as hot embossing or nanoimprint lithography. In embossed/stamped formats, HBR s present an economic route to volume production of high performance optical communications components for datacom and access networks. References [1] T. W. Mossberg, Planar holographic optical processing devices, Opt. Lett. 26, 414 (2001). [2] T. W. Mossberg, Lithographic Holography in Planar Waveguides, SPIE Holography Newsletter, Nov. 2001; see: [3] C. Greiner, D. Iazikov, and T. W. Mossberg, Lithographically-fabricated planar holographic Bragg reflectors, J. Lightwave Technol., accepted for publication. [4] C. H. Henry, R. F. Kazarinov, Y. Shani, R. C. Kistler, V. Pol, and K. J. Orlowsky, Four-channel wavelength division multiplexers and bandpass filters based on elliptical Bragg reflectors, J. Lightwave Technol. 8, 748 (1990). 0 Relative Insertion Loss (db) Wavelength (um) Figure 3. 8-channel HBR-based multiplexer. Upper (lower) graph, measured (simulated) multiplexer spectral transfer function. 3

58 TuI4 Low-loss and compact TFF-embedded silica-waveguide WDM filter for video distribution services in FTTH systems M. Yanagisawa, Y. Inoue, M. Ishii, T. Oguchi, and Y. Hida NTT Photonics Laboratories, 3-1 Morinosato Wakamiya, Atsugi, Kanagawa , Japan H. Izumita 1), N. Araki 1), and T. Sugie 2) 1) NTT Access Network Service Systems Laboratories, Hanabatake, Tsukuba, Ibaraki , Japan 2) NTT Access Network Service Systems Laboratories, 1-6 Nakase, Mihama, Chiba, Chiba , Japan Abstract: Silica-waveguide 1.50/1.55 m WDM filters with a TFF-embedded configuration have been developed for video distribution services in FTTH systems. These filters exhibit low insertion losses of less than 1 db and high reliability Optical Society of America OCIS codes: ( ) Optical devices; ( ) Waveguides, planar 1. Introduction A broadband passive optical network (B-PON) system is the most promising approach to establishing a cost-effective optical access network [1]. Three major U.S. carriers have adopted a set of common technical requirements based on this system. In this enhanced system, specified in ITU-T G.983.3, the wavelength bands for video distribution services and ATM-PON transport services are allocated at m and m, respectively. While bulk-type modules have generally been used for this type of filter, cost-effective and compact / m WDM filters are strongly required. This is because bulk-type filters pose certain problems as regards cost reduction at the mass production stage and the realization of the compact array modules required for central offices. To meet the above demand, WDM filters using a thin film filter (TFF) embedded silica-based planar lightwave circuit (PLC) [2,3] are very attractive thanks to their compactness, mass-producibility and high reliability. However, it has been considered very difficult to obtain a low insertion loss and high isolation when the TFF-embedded waveguide WDM filter is applied to a filter whose guard band is narrower than 0.1 m (i.e. 0 nm). This is because the flat and sharp TFF spectrum deteriorates due to the diffraction of the incident light in the TFF. An L/U-band WDM coupler with a guard band of 25 nm, and composed of a TFF and PLC waveguides, has a high insertion loss of 3 db. [4] Here we demonstrate a new waveguide technology for the TFF-embedded WDM filter designed to overcome the guard band limitation. We have successfully realized / m (50-nm guard band) WDM filters with an extremely low loss and a high isolation for both 1-ch and 8-ch arrays. We have shown for the first time that the optical performance of a PLC-type filter with a 50-nm guard band can be comparable to that of bulk-type filters. 2. WDM filter configuration and fabrication Figure 1 shows the schematic configuration of our proposed WDM filter. A TFF composed of a dielectric multilayer evaporated on a polyimide substrate [5] is inserted into a groove formed at a cross-waveguides intersection. The TFF is designed to have a passband at m and a stopband at m, because we require higher isolation of the 1.50 m signal from the 1.55 m signal than for the opposite wavelength allocation. A 1.55 m 1.55 m Groove Thin-film filter (TFF) 1.50 m < Port E > < Port A > < Port C > m 1.31 m Taper 1.31 m PLC (cross waveguides) Fig. 1 Configuration of TFF-embedded WDM filter and photograph of 1-ch filter module

59 TuI4 wavelength light input into the common port C is therefore reflected by the TFF and then output from the reflection port E. A lateral waveguide taper is adopted to expand the mode-field of the incident light to the TFF and thus suppress the diffraction of the incident light in the TFF, preventing any increase in the insertion loss or deterioration of the transmission spectrum. This waveguide taper is very effective for realizing a WDM filter with a narrow guard band. The silica-based cross-waveguides were fabricated on a Si substrate by a combination of flame hydrolysis deposition and reactive ion etching. The refractive index difference was set at 0.3 %, which is also effective in decreasing the diffraction. Eight sets of cross-waveguides were arranged in a single chip for the 8-ch array filter. A groove was cut at the waveguide intersection with a dicing saw. A 30 m-thick dielectric multilayered TFF was then inserted into the groove and fixed in place with adhesive. The WDM filters were connected with fibers and then housed in a 35 mm-long and 5 mm-diameter package for the 1-ch module, and a 60 mm x 8 mm x 6 mm package for the 8-ch array module. 3. Results Figure 2 shows the wavelength response of the fabricated 1-ch module. Figure 2(a) shows the insertion loss and PDL spectra. The insertion losses were less than 0.85 db at m in the transmission port (C-A) and less than 0.55 db at m in the reflection port (C-E). The PDL was less than 0.1 db for all signal wavelength ranges. The isolation characteristics are also shown in Fig. 2(b) and an extremely high isolation of more than 60 db was obtained at m in the C-A port. Transmittance (db) C-A C-E Wavelength ( m) PDL (db) Transmittance (db) 0 - C-E C-A Wavelength ( m) The loss spectra for the 8-ch array module are also shown in Fig. 3(a) and (b). We also obtained low insertion losses of less than 0.82 db in the m range and of less than 0.66 db in the m range. The insertion loss uniformities for both wavelength ranges were as low as less than 0.2 db. A flat wavelength response was realized in the m wide wavelength range. A high isolation property of more than 42 db at m in the C-A port was also achieved. These low and uniform loss characteristics confirm that our TFF-embedded PLC-type filter is very promising for realizing the array modules. Transmittance (db) (a) Insertion loss (solid line) and PDL (dashed line) (b) Isolation characteristics Fig. 2 Wavelength responses of 1-ch WDM filter module C-A C-E Wavelength ( m) Transmittance (db) C-A C-E Wavelength ( m) (a) Insertion loss (b) Isolation characteristics Fig. 3 Wavelength responses of 8-ch array filter module

60 TuI4 The optical performances of both 1-ch and 8-ch array filters is summarized in Table 1. The target values are estimated from the system performance described in ITU-T G The isolation in the A-E port shows the directivity, which is the crosstalk of the 1.31 m wavelength signal into the port E. Good performance with high clearance to the target is achieved for both the 1-ch and 8-ch modules. These results show that a PLC-type WDM filter can be fabricated with a high yield leading to cost reduction. Table 1 Optical performance of 1-ch and 8-ch array WDM filter modules Item Insertion loss PDL Isolation Port Wavelength Target Measured (db) ( m) (db) 1-ch 8-ch A-C C-A 1.31 < C-E A-C C-A 1.31 < C-E C-A > C-E > ( ) A-E 1.31 > The reliability of the 1-ch modules has been also investigated based on Telcordia GR Figure 4 shows the loss changes during damp heat storage at 75 degrees and 90 %RH for 2000 hours, which is one of the most severe tests for PLC modules. Changes in the insertion losses for the C-A and C-E ports were both less than +/- 0.2 db, and the isolation loss at 1.55 m in the C-A port was kept sufficiently high enough at over 60 db during the test. These results indicate that the proposed WDM filter can be used for practical applications. Loss change (db) C-A (@1.50 m) C-E (@1.55 m) Elapsed time (hrs) Loss (db) C-A (@1.55 m) Elapsed time (hrs) (a) Insertion loss (b) Isolation characteristics Fig. 4 Loss change in 1-ch modules during damp heat storage test 4. Conclusions We have developed silica-waveguide TFF-embedded / m WDM filters that exhibit low insertion losses of less than 1 db, high isolation losses of more than 42 db, and good reliability. We have shown for the first time that the optical characteristics are comparable to those of a bulk-type filter. These compact and high performance PLC filters will enable us to provide cost-effective video distribution services in FTTH systems. References [1] Y. Maeda, K. Okada, and D. Faulkner, FSAN OAN-WG and future issues for broadband optical access networks, IEEE Communication Magazine, Dec, (2001) [2] Y. Inoue, T. Oguchi, Y. Hibino, S. Suzuki, M. Yanagisawa, K. Moriwaki, and Y. Yamada, Filter-embedded wavelength-division multiplexer for hybrid-integrated transceiver based on silica-based PLC, Electron. Lett., 32, (1996) [3] Y. Hida, Y. Inoue, F. Hanawa, T. Fukumitu, Y. Enomoto, and N. Takato, Silica-based 1 x 32 splitter integrated with 32 WDM couplers using multilayer dielectric filters for fiber line testing at 1.65 m, IEEE Photon. Technol. Lett., 1, (1999). [4] N. Araki, H. Izumita, N. Honda, and M. Nakamura, Extended optical fiber line testing system using L/U-band crossed optical waveguide coupler for L-band WDM transmission, in Tech. Dig. OFC 02, (2002) [5] T. Oguchi, J. Noda, H. Hanafusa, and S. Nishi, Dielectric multilayered interference filters deposited on polyimide films, Electron. Lett., 25, (1991)

61 TuI5 Flexible pulse waveform generation using a silica waveguide based spectrum synthesis circuit Koichi Takiguchi, Toshimi Kominato, Hiroshi Takahashi, and Tomohiro Shibata NTT Photonics Laboratories, NTT Corporation, 3-1 Morinosato Wakamiya, Atsugi, Kanagawa , Japan taki@aecl.ntt.co.jp Katsunari Okamoto NTT R&D Fellow, NTT Corporation, 3-1 Morinosato Wakamiya, Atsugi, Kanagawa , Japan Abstract: We demonstrate pulse waveform shaping with a circuit employing an arrayed waveguide grating pair and amplitude-phase controllers. We flexibly synthesized the frequency components of a mode-locked laser to produce 40 GHz square pulses with various widths Optical Society of America OCIS codes: ( ) Integrated optics devices; (060.45) Optical communications 1. Introduction The bit rate of optical communication systems is being steadily increased to 40 Gbit/s and more. Moreover, their functions are being improved to achieve photonic routing in addition to conventional point-to-point transmissions [1]. We are developing optical functional devices based on a planar lightwave circuit (PLC) for these advanced optical networks [2]. Flexible shaping of the temporal pulse profile is also an important function that must be investigated. For example, the manipulation of signal chirping is indispensable in high-speed transmissions, and particular waveforms including square pulses are required in optical signal processing. It was difficult, however, to generate these signals directly with conventional semiconductor or fiber light sources. Spectral amplitude and phase control of the pulses is attractive because this would allow us to produce various kinds of waveforms without using high-speed modulators or optical nonlinear processes. This procedure corresponds to the manipulation of each complex frequency component. A bulk grating pair and masks were adopted as a dispersive element and frequency component controllers, respectively [3, 4]. The use of the grating pair made the apparatus large (~m 2 ), and it was difficult to generate a variety of pulses with mask patterns where the amplitudes and phases were fixed. We have reported a spectrum synthesis circuit that we fabricated using PLC technology [5]. The circuit consists of two arrayed-waveguide gratings (AWGs), variable optical attenuators (VOAs), and phase shifters. We described the preliminary frequency responses to clarify its operation, but did not show its pulse responses and variable functions. In this paper, we demonstrate, for the first time, temporal waveform shaping using this type of PLC circuit. We were able to generate square (flat-top) pulses with various pulse widths, whose repetition rate was 40 GHz. The square pulses produce less nonlinearity in the transmission than conventional pulse waveforms, and can be used for optical gating [6] or metrology [4]. 2. Operating principle and experimental results Figure 1 shows the configuration and operating principle of our spectrum synthesis circuit. Two frequency multi/demultiplexing AWGs were interleaved with an array of 32 Mach-Zehnder interferometer (MZI) type VOAs and thermo-optic (TO) phase shifters. Each VOA was composed of two multi-mode interference (MMI) couplers with a 3 db coupling ratio, and a cross-port was used to obtain a better extinction ratio. The AWGs had a Gaussian transmittance, and their channel spacing, channel number, and free spectral range were 40 GHz, 32, and 3,200 GHz, respectively. We inserted an adjustment part to equalize the waveguide lengths between the AWGs, and monolithically integrated all the components on a silica waveguide on a silicon wafer (66 x 92 mm 2 ) with a relative index difference of 0.75 %. With a view to generating desired waveforms, we designed the VOAs and phase shifters to accurately provide arbitrary amplitude and phase control, respectively, of 32 frequency components, that were demultiplexed by the first AWG. Figure 2 shows the circuit transmittance in the TE mode when all the VOAs were either open or closed. This demonstration of TE mode operation is sufficient because the circuit was installed just behind a laser. The total fiber-to-fiber loss was 13 db. The extinction ratio, namely, the dynamic range was about 30 db, and we were able to set the amplitude with an accuracy of about 0.1 db because the TO phase shift resolution is better than This VOA performance demonstrates a wide dynamic range and precise resolution meeting our requirements. Another type of integrated-optic circuit with a transversal-filter configuration has been reported for these purposes [7]. However, its parameters are difficult to adjust because they do not directly correspond to each frequency component, and the intrinsic loss varies greatly depending on the filter settings. Figure 3 shows the measurement setup. We used a mode-locked fiber ring laser to provide a seed pulse train

62 TuI5 Fig. 1. Configuration and operating principle of spectrum synthesis circuit. Fig. 2. Circuit transmittance. Fig. 3. Experimental setup for measuring pulse spectrum and waveform. whose repetition rate and waveform were 40 GHz and sech 2, respectively. We measured the waveform by utilizing an auto- or cross-correlator. The seed pulse chirping was compensated with a single-mode fiber, and the tapped pulse was used as a reference for the cross-correlation. We generated square pulses with different widths from a seed light source to clarify the circuit validity. Figure 4 shows the original and obtained spectra and auto- or cross-correlated pulse waveforms. The seed pulse had a full width at half maximum (FWHM) width t of 0.9 ps, whose auto-correlation is shown in Fig. 4 (b-1). Figure 4 (a-2, a-3) and (b-2, b-3) show two examples of synthesized spectra and corresponding cross-correlated pulse waveforms for designed pulse widths t of 11.9 and 4.4 ps, respectively. The broken and solid curves show calculated and experimental values, respectively. In the spectra, the value of 0 or indicates the required phase at each frequency component. We were able to synthesize the line intensity spectra with an average accuracy of 0.6 db. The ripples in the flat-top pulse region were caused by the finite available bandwidth, and were estimated to be 0 to 0.1 db compared with the calculated values of 0.1 to 0.2 db. The obtained flat-top regions were generally smoother than the calculated regions. The rise and fall time ( to 90 %) was 2.9 to 5.4 ps, while the target was 1.5 ps. The deterioration in the rise and fall time was mainly brought about by phase setting errors that originated in the thermal crosstalk among the phase shifters used for TO phase adjustment. We must adopt heat-insulating grooves [8] to reduce the crosstalk and thus obtain better performance. The results in Fig. 4 (b-2, b-3) show the trade-off relation between the ripples and the rise and fall time. Figure 5 is an example of measured pulse trains (cross-correlation), corresponding to Fig. 4 (b-2), to demonstrate that we could obtain square waveforms with a 40 GHz repetition rate. Figure 6 shows the relation between the obtained and designed pulse widths. The obtained results (closed ellipses) agreed well with the designed values. To conclude, we were able to achieve our target of generating square pulses with different widths. 3. Summary We have shown the first temporal pulse shaping achieved using a PLC based spectrum synthesis circuit. The circuit was composed of two AWGs and an array of 32 controllers, and enabled us to synthesize frequency components much more accurately and flexibly than with other apparatus or devices. We successfully confirmed its validity by carrying out experiments to generate square pulses with different widths and a 40 GHz repetition rate.

63 TuI5 The circuit has other applications including chirp manipulation and the generation of various modulation formats on high-speed signals, and pulse generation for metrology. These various functions are difficult to achieve with a conventional optical device, and our circuit will play an important role in the coming advanced optical networks. 4. References [1] P. Green, Progress in optical networking, IEEE Commun. Mag. 39, (2001). [2] K. Takiguchi, Photonic functional devices based on a silica planar lightwave circuit, in Proc. ECIO 03 FrB2, (2003). [3] J. P. Heritage et al., Picosecond pulse shaping by spectral phase and amplitude manipulation, OL, (1985). [4] A. M. Weiner et al., Synthesis of phase-coherent, picosecond optical square pulses, OL 11, (1986). [5] K. Okamoto et al., Fabrication of frequency spectrum synthesizer consisting of arrayed-waveguide grating pair and thermo-optic amplitude and phase controllers, EL 35, (1999). [6] T. Morioka et al., Multiple-output, 0 Gbit/s all-optical demultiplexer based on multichannel four-wave mixing pumped by a linearly-chirped square pulse, EL 30, (1994). [7] K. Kitayama et al., Optical pulse train synthesis of arbitrary waveform using weight/phase-programmable 32-tapped delay line waveguide filter, in Proc. OFC 01 WY3 (2001). [8] S. Sohma et al., Low switching power silica-based super high delta thermo-optic switch with heat insulating grooves, EL 38, (2002). (a-1) Original spectrum. (a-2) Synthesized spectrum ( t=11.9 ps). (a-3) Synthesized spectrum ( t=4.4 ps). (b-1) Original waveform. (b-2) Generated waveform ( t=11.9 ps). (b-3) Generated waveform ( t=4.4 ps). Fig. 4. Spectra (a) and corresponding waveforms (b) of pulses. Broken curve : calculated and solid curve: experimental. Fig. 5. Measured pulse train. Fig. 6. Obtained versus designed pulse width.

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