PLANNING AND DESIGN OF FRONT-END FILTERS

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1 PLANNING AND DESIGN OF FRONT-END FILTERS AND DIPLEXERS FOR RADIO LINK APPLICATIONS Kjetil Folgerø and Jan Kocba Nera Networks AS, N-52 Bergen, NORWAY. Abstract High capacity radio links with data rate up to 155 Mbps per annel operating in the requency range rom 4 GHz to 4 GHz are used in mobile inrastructure and backbone networks. The requency bands are crowded, and thereore subject to strict regulation rom national and international governing bodies. Despite eorts o standardization, there exist numerous region-speciic requirements to whi equipment manuacturers must comply. Hence, there is a need or a large number o ilters and diplexers to cover the complete range o requencies. It is important with eicient planning and short design cycles when su a large number o variants are to be designed. This paper describes an eicient way o planning and designing ront-end diplexers using automated design procedures and commercial electromagnetic solvers diplexer is oten made meanically symmetrical su that the same unit can be used both or transmitting in the upper and lower side-band. 28 MHz MHz #1 #2 #3 #4 #5 #6 #7 #8 #1 #2 #3 #4 #5 #6 #7 #8 (a) ( #2) ( # 2 ) I. INTRODUCTION A radio link system uses requency division duplexing to separate transmitted and received signals. Fig. 1a shows an example o a annel plan in the lower 6 GHz requency band. To minimize RF annel intererence, all transmit requencies on one terminal station are either in the lower or upper sideband and the receive requencies are in the other sideband. For example, i annel 2 in the lower sideband is used or transmitting at one terminal, annel 2 in the upper side-band will be used or receiving as illustrated in Fig. 1b. The adjacent station will obviously have transmit and receive requencies in the other side-band. The iltering needs or access radio systems (su as mobile inrastructure networks) are connected to regulatory and intererence issues; and the driving concern is that the iltering must suppress signals rom own transmitter to interere with the received signals (cr. Fig. 2). The receive ilter thereore needs a high rejection at the transmit requencies, and the transmit ilter needs to suppress the leakage rom the transmitter at receive requencies. In addition, the transmit iltering must ensure that the spurious and harmonics rom the transmitter are attenuated below regulatory levels, and the receive iltering must ensure that interering signals outside the requency band is suppressed beore entering the LNA. Usually, transmit and receive iltering is combined in a diplexer. It is important that one diplexer can cover as many annels as possible. The requency plan is divided into sub-bands in order to make realizable diplexer requirements as shown in Fig. 3. The (b) Fig. 1. (a) Frequency plan in the lower 6 GHz requency band, (b) Radio link hop utilizing requency division duplexing. Tx Rx Fig. 2. Filtering #1 #2 #3 #4 Diplexer 28 MHz MHz #5 #6 #7 #8 #1 #2 #3 #4 #5 #6 #7 #8 Fig. 3. The requency plan is divided into sub-bands. A diplexer covering one sub-band is shown Despite eorts o standardization, there exist a large number o annel plans. Fig. 4 shows an example or 28-4 MHz plans in the 8 GHz requency band. Eort must be made to reduce the number o variants to a minimum in order to lower the manuacturing cost and logistics. Nevertheless, a large number o diplexer variants have to be designed. There is thereore a need or reliable and automated design procedures or diplexers. This paper presents an eicient way o planning and designing ront-end diplexers using automated design procedures and commercial electromagnetic solvers.

2 (29.65 MHz) models instead o less accurate ideal models or timeconsuming ull-wave simulations o complex ilters. In the ollowing section we describe su a reliable circuit model. (3 MHz) (29.65 MHz) (29.65 MHz) Fig. 4. Example o requency plans (28-3 MHz plans in the 8 GHz requency band). Division into diplexer subbands is also shown. 1) Trained circuit model The basic circuit model [1], [2] shown sematically in Fig. 6 consists o a ain o K- inverters and transmission lines with phase constant β and length l. The K-inverters K i,i+1 or the individual couplings are calculated directly rom the ilter speciications (i.e. centre requency, bandwidth, ilter order and pass-band ripple) [1], [2]. The phase i,i+1 or an ideal K-inverter is given as i,i+1 =±. Plan diplexer variants Check perormance Design ilters Design diplexer Prototype Fig. 5. Procedure or planning and designing diplexer variants. II. DESIGN APPROACH The suggested planning and design procedure can be divided into the steps shown in Fig. 5. In the ollowing we describe these steps in more detail. A. Variant planning and perormance evaluation A irst attempt o deining the diplexer variants or optimum coverage o the annel plans is done based on experience, or by simply guessing. Filter parameters (i.e. centre requency, bandwidths, ilter order and return loss) or various sub-bands are osen (see example in Fig. 4), and the ilters responses are calculated using an accurate trained circuit model. I the requirements are ulilled, we can proceed to the next step. I not, the ilters parameters need to be redeined, and new calculations must be made. This process is repeated until the requirements are ulilled. It is obvious that a ast method o calculating the ilter response is needed i numerous iterations have to be made. Further on, it is important that the response calculated is as close as possible to the response that can be realized. It is also important to consider practical issues su as temperature eects, tolerance inluence and tuning margins already at this stage. Hence, eective planning o variants highly relies upon using reliable circuit Fig. 6. Circuit model The K-inverters o the ideal circuit model is independent o requency. In order to increase the accuracy o the model, a Taylor expansion is applied to the K-inverters and phases as a unction o requency to include the requency dependency o the couplings in the circuit model; K ( ) K( + ) K( ) + α, α = φ( ) φ ( + ) φ( ) + κ, κ =. The expansion coeicients α and κ are calculated rom CAD simulation o the couplings. Typically, the requency dependency (and the value o the expansion coeicients) will increase with increasing iris thicknesses. A similar Taylor expansion can also be applied to small anges in the dimensions, and this can be utilized to perorm tolerance analysis in short time rames. More details are ound in [3]. The trained circuit model can also be generalized to included conductor losses and temperature eects, but this is not described any urther here. Fig. 7 illustrates that the ideal Chebyev response diers rom the ull wave simulations by approximately 8 db at 16 GHz, while there is excellent agreement between the trained circuit model and ull-wave simulations. Fig. 8 shows an example o using the trained circuit model in planning o diplexer variants. The response o a diplexer including tolerances is calculated within a ew seconds, and it can automatically be determined i the requirements are satisied.

3 Step 1: Step 2: S (db) a Tune a ai 12 Calculate lr requency (GHz) EM solver Basic Circuit model Trained Circuit model Fig. 7. There is excellent agreement between trained circuit model and ull wave simulations. Fig. 8. Fast estimation o diplexer response with the trained circuit model. B. Filter design The next step in the process is to design the diplexers or ea sub-band. The shape o the diplexer is irst osen based on the meanical outline o the product. The two ilters that constitute the diplexer are then designed independently to the exact shape wanted. The realization procedure may be divided into simple steps (see Fig. 9), leading to a design-process well suited or implementation as an automatic procedure on a computer. Ea step involves the tuning o one dimension (coupling size or cavity length) in a simple waveguide structure until the simulated s-parameters it the s-parameters o the representative circuit model. The waveguide structure is kept simple by only including the couplings and resonators that interact strongly, i.e. either a single coupling, a single cavity or two coupled cavities. Thus, even a 3D EM tool su as HFSS can be used or the design [4]. Fig. 9. Step-by-step illustration o the design procedure or an in-line direct-coupled inductive iris ilter. A ast convergence or ea step is obtained because (i) the simulated substructure is very simple, (ii) only one variable is tuned, (iii) only one or a ew requency points are considered and (iv) the tuning goals are easy to aieve with ew iterations. The procedure is described in detail in [4] and [5]. Here, we repeat the main steps or an in-line direct coupled ilter: 1. Calculate the theoretical S 21 or the couplings as reerence values. Then simulate a waveguide with a single iris in a ull-wave simulator (see Fig. 9) at ω, and ange the dimension o the iris until the simulated S 21 equals the reerence values. 2. Calculate the length o the cavities as l r = π 2 ( φ + φ ) λ g 2π 1 where φ 1 and φ 2 are the phases o S 11 or the input and output coupling o the corresponding resonator, respectively, as ound in step 1, and λ g is the guide wavelength at ω. Alternatively, simulate a cavity with a ull-wave solver and tune the length o the cavity until the simulated S 21 curve has a peak at the centre requency ω. To meet the exact shape requirements o the ilters, it is oten required to use olded ilters or ilters with cross-coupling. A thorough description o the design process applicable or su ilters is given in [5]. The method described has been ully automated using Matlab and the mode-mating/inite element program Wasp-Net [6]. Fig. 1 and Fig. 11 show examples o ilters with a lexible shape that are designed automatically rom speciications with this process. 1 2

4 S (db) the relection coeicients at the common port is optimized. 2. Adjust the two couplings and the length o the two resonators closest to the junction until the relection coeicients at the common port is optimized. 3. Successively add adjacent couplings and resonators to the optimization variables, and adjust the dimensions until the return loss requirements are ulilled. Fast convergence is obtained i optimization is only done on the ilters transmission poles. This method has also been ully automated using Matlab and Wasp-Net [6]. Fig. 12 shows measurements and simulations o a metal insert diplexer designed using this approa requency (GHz) Fig. 1. Folded ilter designed with the automated procedure III. EXAMPLE Fig. 13 shows an example o a diplexer variant planned and designed using the approa described in this paper. It was ound that a seventh order ilter was necessary to cover 4 annels in the lower 6 GHz requency plan. Temperature eects are taken into account but tolerance eects are neglected because diplexer tuning is used to account or meanical inaccuracies in this case. IV. CONCLUSION Short time-to-market requires clever planning o diplexer variants and short (preerable automated) design cycles. In this paper we have shown that this can be aieved by using accurate trained circuit models and accurate CAD tools in combination with reliable design methodology. Fig. 11. Cross-coupled ilter designed with the automated procedure C. Diplexer design Next, the two ilters must be combined with a three-port junction to orm a diplexer. The simplest diplexer design is aieved i a Y-junction is osen [7], but in most cases there are strong meanical restrictions on the shape o the junction su that T- junctions are more applicable. The procedure or adapting separate ilters and junction to orm a diplexer is as ollows 1. Connect the ilters to the junction and adjust the distance between the ilters and the junction until REFERENCES [1] G. L. Matthaei, L. Young, and E. M. T. Jones, "Microwave ilters, impedance-mating networks, and coupling structures," New York: McGraw-Hill, 1964 (Reprint Norwood: Arte House, 198) [2] S. B. Cohn, "Direct-coupled-resonator ilters," Proc. IRE vol. 45, pp , 1957 [3] K. Folgerø and J. Kocba, Yield-Driven Design o Direct-Coupled Waveguide Filters with Minimum Use o Full Wave EM Solvers European Microwave Con, Muni, October 23 [4] K. Folgerø, "Step-by-step procedure or design o waveguide ilters with HFSS," Ansot HFSS User Workshop, Los Angeles, January 21 [5] J. Kocba and K. Folgerø, Design Procedure For Waveguide Filters With Cross-Couplings, IEEE MTT-S Int. Microwave Symp. Seattle, June 22, [6] Wasp-Net: Manual and design examples", Microwave Innovation group, Bremen, Germany, 21 [7] A. Morini, and T. Rossi, Constraints to the optimum perormance and bandwidth limitations o diplexers employing symmetric three-port junctions, IEEE Trans Microwave Theory Te. vol 44, no 2, 1996, pp

5 Fig. 12. Metal-insert diplexer. Fig. 13. Folded diplexer

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