HIGH POWER FACTOR ELECTRONIC BALLAST OPERATING AT CRITICAL CONDUCTION MODE

Transcription

1 HIGH POWER FACTOR ELECTRONIC BALLAST OPERATING AT CRITICAL CONDUCTION MODE Msircio A. C6, Domingos S.L. Simonetti and J.L. Freitas Vieira Universidade Federal do Espirito Santo Departamento de Engenharia ElBtrica - LEPAC - PO Box: VitQia - ES - Brazil Fax: joseluiz@ele.ufes.br ABSTRACT - A high power factor electronic ballast, that exhibits low switching losses is presented in this paper. The proposed topology is based on a single power processing stage which provides high frequency voltage to the fluorescent lamps and high power factor to the utility line. The lamps are driven by a self-oscillating half-bridge series resonant converter of great simplicity and attendant low cost and increased reliability. High power factor is achieved by using a non-conventional boost converter operating at critical conduction mode. Theoretical analysis and experimental results for two 40W fluorescent lamps operating at SOkHz from 127V utility line have been obtained, which demonstrate the high efficiency and high power factor of this electronic ballast. 1 - INTRODUCTION Fluorescent lamps performance rises when the electronic ballast is used instead of magnetic ballast. When operating at high frequency, the following characteristics can be emphasized [ 1,2,3,4]: the luminous efficacy increases by about 10% which reduces the energy consumption, weight and size can be reduced, flickering as well as stroboscopic effects can be eliminated, and the audible noise fall to unnoticeable levels. In order to obtain a compact electronic ballast the operating frequency must be raised. However, at high frequency the soft commutation techniques are recommend to maintain high efficiency. The boost converter can operate as a pre-regulator stage at discontinuous conduction mode. In this case, the input current follows naturally the sinusoidal waveform of the input voltage, providing HPF to the utility line [7,8]. However two power processing stages increase the final cost, besides reducing the electronic ballast reliability. An interesting option to avoid these problems are the high power factor electronic ballast based on a single power processing stage [5,9,10,11,12,13]. This paper proposes a HPF electronic ballast using a single power processing stage operating at critical conduction mode. This mode provides the rms current reduction in the switches, compared to that of the discontinuous conduction mode presented in [ 131. This electronic ballast can be understood as being formed by two cascade stages. The first stage performs PFC, which is a non-conventional boost operating at critical conduction mode. The second one is a self-oscillation series-parallel resonant converter operating above the resonant frequency to provide zero voltage switching 2 - CONVERTER TOPOLOGY The power stage diagram of the proposed high power factor electronic ballast is shown in Fig. 1. High Power Factor (HPF) with low Total Harmonic Distortion (THD) is a feature specified by international regulations, and is also required by the utility companies. The advantages of HPF include reduction in the rms line current and line current harmonic distortion. In this way, the utility line can be more efficiently utilized [5]. Ch CI HPF can be obtained using two power processing stages. The first one is the pre-regulator stage, which converts the AC input voltage to a DC voltage with high power factor. The second stage transforms the DC voltage to a high frequency AC voltage that drives the fluorescent lamps. The Power Factor Correction (PFC) performed by that boost pre-regulator, operating at continuous conduction mode, is know as active PFC. This method provides nearly unity power factor with THD less than 5% [1,4,6]. Fig. 1 - Power stage diagram of the proposed HPF electronic ballast. The non-conventional boost converter, which operates as a PFC stage, is formed by the boost inductor L, and the transformer secondary which are within the fast diode bridge D, - D,. In this topology the boost inductor current never /96/$ IEEE 962

2 dwells at zero value, since there is always a path to the circulating current through the fast diode bridge. This characterizes the operation at critical conduction mode for the current during the full 6OHz cycle of the utility line. Then, the current naturally follows the sinusoidal shape of the input voltage, providing HPF. The differences between the current conduction modes of this converter and that presented in [13], can be verified from the simulation results shown in Fig. 2. This results were obtained from simplified circuits, for: V, = 1155senel, V,=19OV, f,= 6OHz and V,, = V,. To emphasize the details of the current conduction modes, a small switching frequency of looohz was chosen. a) I in Vi" 5) '.FA * * B *...* * k 1.h z.bi 3. k *.%a k?.k vi DI * * l * *... + ~.e.~ m *.%a 5. k 6.m 7.m 8. k 9.m Fig. 2 - Simplified circuits and dc link current waveforms for: (a) electronic ballast presented in [13] and (b) electronic ballast proposed in this paper. D1 VO The waveforms of Fig.2 shows the DC link current of this electronic ballast at critical conduction mode, in contrast to that of [ 131, which operates at discontinuous conduction mode. Furthermore, the boost inductor current as well as the transformer secondary current, of the electronic ballast described in [ 131, contain a DC value which leads to a more elaborate design due to core saturation. The DC link current frequency is twice the switching frequency. Therefore, the high harmonics currents are eliminated with a smaller input filter when compared with that proposed in [13]. The secondary with 1:2 turns ratio ensures critical conduction mode operation over full cycle of the utility line. The tertiary with 1:1.5 turns ratio provides the necessary voltage level to drive two 40W series connected fluorescent lamps and in addition ensures circuit isolation for the lamps. 3 - PRINCIPLE OF OPERATION To establish the principle of operation the following assumptions are made: - M, and M, operate with fixed frequency, 0.5 of duty cycle and 180" out of phase; - the primary transformer voltage is a square waveform with high frequency and amplitude equal to VJ2; - the V,, square wave has amplitude equal to V,, what is assured by the 1:2 turn ratio. It maintain current at critical conduction in the ac power supply during the full cycle. - the output voltage profile of the input rectifier is simply the rectified AC voltage; - the L, inductor voltage remains constant during a switching cycle, since the switching frequency is much greater than the supply frequency; - this ballast always operates as a boost converter, for the voltage V, is always larger than the AC peak voltage. This is ensured by an appropriated value. - the capacitance C, is sufficiently large to be considered a voltage source; - at steady-state the fluorescent lamp can be considered as a resistive load. Based on the above assumption, and considering a single switching period, this electronic ballast can be viewed as two simplified independent converters. The first one is obtained when the capacitance CO is replaced by the voltage source V,, and the transformer secondary is replaced by the high frequency square wave voltage source V,,,. The resulting converter is shown in Fig.3, which represents a non-conventional boost converter. This converter operates like a power factor correction stage at critical conduction mode. 963

3 I - Fig.3 - Simplified non-conventional boost converter operating in critical mode. The second one is obtained when the transformer tertiary is replaced by the high frequency square wave voltage source Vswb. The resulting converter is shown in Fig.4, which is a series-parallel resonant converter. This converter has been an attractive choice to be used in lighting electronic systems, because it ensures an appropriated voltage during the lamp ignition process, and maintain the steady-state rated current. 3ulAhge - linear increasing of i, through D, and D, (h, 5): At instant 4, the inductor current i, becomes zero. The voltage V,,, = V, turns the diodes D, and D, on. The inductor voltage V, = -V, maintains i, increasing linearly. 4WShge - linear decreasing of i,, through D, and D, (4, t,): At instant G, the inductor current i, reaches the value -Ibm. The voltage V,,, changes its value to -V,, and keeps the diodes D, and D, on. The inductor voltage V, = 2V,-V, maintains i, increasing linearly. This stages ends at instant t,, when i, becomes zero. I I 1st Staee Fig.4 - Simplified series-parallel resonant converter. Consequently, the proposed electronic ballast operation, can. be understood as the cascade operation of the two independent converters referred above. 2nd. Stage DI DI 4 - STAGES OF OPERATION AND WAVEFORMS Non-Conventional Boost Converter at Critical Conduction Mode At steady-state, a switching cycle of the converter operation is represented by four stages. The equivalent circuits of these stages are shown in Fig.5. Non-conventional boost converter operation at critical conduction mode is described as follow: 3rd. Stage l,&s@y - linear increasing of i,, through D, and D3 (to, tl): At instant to, the inductor current i, is equal to zero. The voltage V,,, = - V, turn the diodes D, and D, on. The inductor voltage V, = V, maintains i, increasing linearly. 2nLQage - linear decreasing of i,, through D, and D, (tl, tj: At instant t,, the inductor current i, reaches the value Ibm. The voltage V,,, changes its value to V,, and keeps the diodes D, and D, on. The inductor voltage V, = - (2Vo-Vi,) maintains i, decreasing linearly. 4th. Stage Fig.5 - Stages of operation of the non-conventional boost converter. The main waveforms of the non-conventional boost converter are shown in Fig.6. The variables are defined as follows: VO 964

4 t, - linear increasing time; td - linear decreasing time; I,, - i, peak value in a switching period i, - DC link current; Vim = lvpsin8( is the rectified input voltage, considered constant for a switching period; V, - DC link voltage, which is greater than the input peak voltage VpW. 5 - RELEVANT ANALYSIS The relevant characteristics of the proposed electronic ballast are defined by the: input current, power factor and THD. The main parameters to be determined are: the boost inductance and the resonant parameters. 5.1 Input Current Due to the high frequency input filter, the AC line current should be given by the instantaneous mean value of the boost inductor current, according to the following equation [8]. t W - -.mt I- - -I---+ t > where: 5.2 Input Power The input power is obtained from the following equation: where: From equations (1)- (4) the following expression results: Fig.6 - Non-conventional boost converter waveforms at critical mode Series-Parallel Resonant Converter This converter operates above the resonant frequency to provide zero voltage switching. If the series-parallel resonant circuit shown in Fig.4 is reflected to the transformer primary, this converter operation can be considered like the well-know self-oscillating half-bridge series-parallel resonant converter [ 14,151. The complete electronic ballast operation is established when the two independent converters referred above are operating in cascade. To obtain the MOSFETs currents and the transformer primary current the independent waveforms should be added Boost Inductance The boost inductance is obtained from equation (4), considering the output power is given by: P, = q.pi,, where q is the electronic ballast efficiency. Hence, the normalized boost inductance is defined by: where: The normalized boost inductance as function of a, with Po as the parameter, can be obtained from Fig.7. (7) 965

5 Lblk 0.025~ The THD of this electronic ballast, as function of a, is shown in Fig.9. / > a Fig.7 - Normalized boost inductance as a function of a with Po as a parameter. 5.4 Power Factor The power factor is defined by: PF = Pin VAC,.iAC,, Considering that the input voltage does not have harmonic components, the power factor can be given by: The proposed electronic ballast power factor as function of a, given by equation (9), is shown in Fig.8. PF 14 : : 8: 6: 4' 2:, H. + a Resonant Parameters Fig. 9 - THD as function of a. The resonant parameters are obtained considering the start-up and steady-state conditions, according to the equations described in [ RMS MOSFETs Current of the Non-Conventional Boost Converter The rms currents of the MOSFETs for the non-conventional boost converter, which are symmetrical, can be determined from the boost inductor current waveform, taking into account the transformer ratio. Therefore, the normalized MOSFET rms currents are described by: I& >a Fig. 8 - Electronic ballast power factor as function of a. IY,.M~- = - 1lhl.W ImlMZm, are the MOSFETs normalized currents (12) = a, is the base current (13) VO The rms normalized currents of the MOSFETs in this non-conventional boost converter, along with those proposed in [13], are shown in Fig Total Harmonic Distortion - THD As can be seen in Fig. 10, the MOSFETs conduction losses in the non-conventional boost converter presented in this Considering unit displacement factor, the THD can be paper are smaller than those of the topology proposed in [ 131. defined by: Considering that the conduction losses due to the rms current of the series-parallel output stage are the same, it can be concluded that this electronic ballast has smaller conduction THD= ri' losses in the MOSFETs.

6 7 - EXPERIMENTAL RESULTS An electronic ballast prototype has been built to meet the input data specifications. The complete diagram is shown in Fig. 1 1, whose parameters and components are the following: r..., ,= ' Fig RMS normalized currents of the MOSFETs for the non-conventional boost converter: IEn,), and for those proposed in [13]: G(*) and G(*). 6- DESIGN PROCEDURE AND EXAMPLE A design procedure of this electronic ballast, along with a practical example, is presented next. - = 550 mh, 80 turns on core EE 30114, IP6 - Thornton; - L, = 1.45 mh, 66/2 turns on core EE 30114, IP6 -Thornton; - Lf = 0.4 mh, 37 turns on core EE 20/10, IP6 - Thornton; - Transformer TI : 33/66/50 turns on core EE4U15, IP6 - Thornton; - Transformer T2 : 5/14/14 turns on core EE 20/10, IP6 - Thornton; - Input diode rectifier bridge, D, - D4 : 1N 4004; - Fast diode bridge D, - D,: SK3GF04 (Semikron), - MI, M, : IRFP 244 ( International Rectifier); - C,, = C, = 10 nf/ 1500V, C, = 100 nf/400v, (polypropylene); - CO = 220 pf/250v, C, = C, = 1 pf/250v, (polypropylene); The components value of the start-up, snubber, overvoltage and overcurrent circuits are specified in [ 131. DI DI -(>crm1 4 I a) Input Data: - rms AC input voltage: VAC = 127 V +/- 15%,60 Hz; - output power: Po = 80 W; - switching frequency: f, = 50 khz; - fluorescent lamp rated current: I,, = 0.35 A; - fluorescent lamp ignition voltage: Vi, = 1000 V; - efficiency: q t 92%. b) Selection of DC Link Voltage: As this electronic ballast always operates as a boost converter, the voltage V, must be larger than the maximum Fig. Overvoltage and overcurrent areuit 1 - The complete diagram of the proposed electronic ballast. AC Peak Vpmm = l8ov. In this case, = l9ov has Experimental waveforms have been obtained for: been selected. VAc=127 V, IA,=0.67 A. The input AC voltage and current, which demonstrate the high power factor of this electronic c) a and k Parameters: ballast, are shown in Fig.12. From equations (2) and (7) results: d) Boost Inductance, Power Factor and THD: and k= From equations (6) and Figures (8) and (9) results: ]a=540mh, PF=0.991 and THD=13.4%. e) Resonant Parameters: From equations presented in [13] results: C, = 89 nf, C, = 5.9 nf and Lf A.47 mh. f) RMS MOSFETS Currents in the Non-Conventional Boost Converter: From equations (1 1)-( 13) results: ZM,,M~- = 1.04A Fig Input voltage VAC (50 V/div) and input current I, (0,5 Ndiv), Time scale: 2ms/div. 967

7 The rectified input voltage and the DC link current are shown in Fig.13. The inductor boost current presenting an 120Hz envelope is shown in Fig.14. The MOSFET M, commutation can be seen in Fig.15, which shows the zero voltage switching. The following characteristics were obtained experimentally: q=94%, PF=0.99 e THD=14%. Fig Rectified input voltage V, (50 V/div) and the DC link current I, (0,5 Ndiv), Time scale: 2ms/div. Fig Boost inductor current at 120 Hz, i, (0,5 Ndiv), Time scale: 2ms/div. Fig MOSFET M, commutation, V, (50 V/div), I, (2 Ndiv), Time scale: 2ps/div. 8 - CONCLUSION This paper presented an electronic ballast with high power factor in a single power processing stage, operating at critical conduction mode. From the simplified analysis it is possible to describe the principle of operation of this electronic ballast, as that as two cascade independent converters. The first one is the non-conventional boost converter, operating at critical conduction mode. Then the input current naturally follows a sinusoidal envelope, resulting in a high power factor to the utility line. The second one, is the well known self-oscilation series resonant converter, operating above the resonant frequency to provides ZVS for the MOSFETs. The frequency of the DC link current as well as that of the line current is twice the switching frequency. So, the high harmonics currents are eliminated with a smaller input filter, when compared to the DCM electronic ballasts. Experimental results have been obtained for two 40W fluorescent lamps operating at 50kHz switching frequency and 127V line voltage, which demonstrate the high power factor and efficiency of this electronic ballast. REFERENCES [I] M. K. Kazimierczuk and W. Szaraniek, "Electronic Ballast for Fluorescent Lamps", IEEE-Transactions on Power Electronics, vol. 8, No. 4, October 1993, pp [2] Edward E. Hammer and Terry K. McGowan, "Characteristics of Various F40 Fluorescent Systems at 60Hz and High Frequency", Transactions on Industry Applications, Vol.IA-21, No. 1, JanuaryFebruary 1985, pp [3] E.C. Nho, K.H. Jee and G.H. Cho, "New Soft-Switching for High Efficiency Electronic Ballast with Simple Structure", Int. Joumal of Electronics, Vo1.71, No.3, 1991, pp [4] Laszlo Laskai and Ira J. Pitel, "Discharge Lamp Ballasting", IEEE-PESC'95, Tutorial 2, 1995, Atlanta, GA, USA. [5] Ed Deng and Slobodan Cuk, "Single Stage, High Power Factor, Lamp Ballast", IEEE-APEC Proc., 1994, pp [6] Jim Spangler and Anup K.Behera, "Power Factor Correction Used for Fluorescent Lamp Ballast", IEEE-IAS Proc., 1991, pp [7] J.M. Alonso, J. Dias, C. Blanco, and M. Rico, "A Smart-Lighting Emergency Ballast for Fluorescent Lamps Based on Microcomputer", IEEE-APEC Proc., 1993, pp [8] Kwang-Hwa Liu and Yung-Lin Lin, "Current Waveform Distortion in Power Factor Correction Circuits Employing Discontinuous-Mode Boost Converters", IEEE-PESC Proc., 1989, pp I.Takahashi, "Power Factor Improvement of a Diode Rectifier Circuit", IEEE-IAS Annual Meet. Proc., 1990, pp [IO] Laszio Laskai, Presad Enjeti and Ira J. Pitel, "A Unity Power Factor Electronic Ballast for Metal Halide Lamps", IEEE-APEC, 1994, pp [ll] Carmelo Licitra, Luigi Malesani, Giogio Spiazzi, Paolo Tenti and Antonio Testa, "Single-Ended Soft-Switching Electronic Ballast with Unit Power Factor", IEEE-APEC Proc., 1991, pp [I21 Ed Deng and Slobodan Cuk, "Single Switch, Unit Power Factor, Lamp Ballasts", IEEE-APEC Proc., 1995, pp [13] J.L. Freitas Vieira, Mfircio A. C6 and Lucian0 D. Zorzal, "High Power Factor Electronic Ballast Based on a Single Power Processing Stage", IEEE-PESC Proc., 1995, pp [14] M.I. Mahmoud, "Design Parameters for High frequency Series Resonance Energy Converters Used as Fluorescent Lamp Electronic Ballast", EPE Proc., 1989, pp [15] Peter N. Wood, "High Frequency Discharge Lamp Ballasts Using Power MOSFETs, IGBT's and High Voltage Monolithic Drivers", PCI hoc., 1989, pp