THREE PHASE FOUR WIRE VOLTAGE SOURCE PWM RECTIFIER WITH LOW INPUT CURRENT HARMONIC. M.Sc. Thesis by İbrahim GÜNEŞ

Transcription

1 İSTANBUL TECHNICAL UNIVERSITY INSTITUTE OF SCIENCE AND TECHNOLOGY THREE PHASE FOUR WIRE VOLTAGE SOURCE PWM RECTIFIER WITH LOW INPUT CURRENT HARMONIC M.Sc. Thesis by İbrahi GÜNEŞ Departent: Prograe: Electrical Engineering Electrical Engineering JANUARY 008

2 İSTANBUL TECHNICAL UNIVERSITY INSTITUTE OF SCIENCE AND TECHNOLOGY THREE PHASE FOUR WIRE VOLTAGE SOURCE PWM RECTIFIER WITH LOW INPUT CURRENT HARMONIC M.Sc. Thesis by İbrahi GÜNEŞ ( ) Date of subission : 8 Deceber 007 Date of defence exaination: 9 January 008 Supervisor (Chairan): Asst. Prof. Dr. Deniz YILDIRIM Mebers of the Exaining Coittee: Asst. Prof.Dr. Özgür ÜSTÜN (I.T.U.) Asst. Prof.Dr. Metin AYDIN (K.U.) JANUARY 008

3 İSTANBUL TEKNİK ÜNİVERSİTESİ FEN BİLİMLERİ ENSTİTÜSÜ ÜÇ FAZ DÖRT İLETKENLİ DARBE GENİŞLİK MODÜLASYONLU DÜŞÜK AKIM HARMONİKLİ GERİLİM KAYNAĞI TİPİ DOĞRULTUCU YÜKSEK LİSANS TEZİ İbrahi GÜNEŞ ( ) Tezin Enstitüye Verildiği Tarih : 8 Aralık 007 Tezin Savunulduğu Tarih : 9 Ocak 008 Tez Danışanı: Diğer Jüri Üyeleri: Yrd. Doç. Dr. Deniz YILDIRIM Yrd. Doç. Dr. Özgür ÜSTÜN (İ.T.Ü.) Yrd. Doç. Dr. Metin AYDIN (K.Ü.) OCAK 008

4 ACKNOWLEDGEMENTS I would like to express y sincerest thanks to Asst. Prof. Dr. Deniz YILDIRIM for his guidance, support, encourageent and valuable contributions during y graduate studies. His ipressive knowledge and technical skills has been a odel for e to follow. I express y deepest gratitude to y faily, y wife Kevser, y other Üran, y father Ali İhsan, y brother Gökhan, y brother-in-law Fatih for their support throughout. Without their endless love and encourageents, I would not coplete this thesis. Special appreciation goes to Bülent Üstüntepe, Osan Okay and Argun Yüzüşen for sharing their knowledge and valuable ties with e during experiental studies. I wish to thank to ENEL Enerji Elektronik A.S. and staff for their help throughout y graduate studies. İbrahi GÜNEŞ February 008 iii

5 TABLE OF CONTENTS ACKNOWLEDGEMENTS TABLE OF CONTENTS LIST OF ABBREVIATIONS LIST OF TABLES LIST OF FIGURES ÖZET SUMMARY iii iv vi vii ix xiii xiv 1. INTRODUCTION Rectifiers and Electric Power Quality Haronic Reduction Methods General Introduction to Three Phase PWM Rectifiers Basic Topologies and Characteristics Operation of the Voltage Source PWM Rectifier Control Methods Linear Current Control Deadbeat Control Hysteresis Control Resonant Filter Bank Controller Outline of the Thesis 13. RESONANT FILTER BASED INPUT CURRENT CONTROL OF THE THREE PHASE FOUR WIRE VOLTAGE SOURCE PWM RECTIFIER Introduction 15.. Resonant Filter Fors Ideal Resonant Filter Phase Delay Copensated Resonant Filter Daped Resonant Filter 0.3. Resonant Filter Bank Fors.3.1. Daped Resonant Filter Banks.3.. Phase Delay Copensated and Daped Resonant Filter Banks 4.4. Discrete Tie Ipleentation of Resonant Filter Banks 6.5. P+Resonant Controller 7.6. The Full Control Structure of the Four Wire PWM Rectifier 30 iv

6 3. SIMULATION RESULTS OF THE THREE PHASE FOUR WIRE VOLTAGE SOURCE PWM RECTIFIER Introduction Modeling of the Three Phase Four Wire Voltage Source Rectifier Siulation Results Trial and Error Based Voltage Controller Tuning Procedure Trial and Error Based Current Controller Tuning Procedure Siulation Results of the Four Wire Rectifier Under Highly Distorted Utility Siulation Results of the Four Wire Rectifier Under Distorted Utility Siulation Results of the Four Wire Rectifier Under Undistorted Utility Siulation Results of the Four Wire Rectifier Under Highly Distorted Utility with the Fundaental Frequency of 49,8 Hz EXPERIMENTAL PERFORMANCE INVESTIGATION OF THE THREE PHASE FOUR WIRE VOLTAGE SOURCE PWM RECTIFIER Introduction Hardware and Software Set Up of the Three Phase Four Wire Voltage Source PWM Rectifier Experiental Results CONCLUSION 76 REFERENCES 78 APPENDIX A : Basic specifications of Intelligent Power Module 81 APPENDIX B : Main features of Digital Signal Processor 8 v

7 LIST OF ABBREVIATIONS PWM IGBT UPS RMS EMC PI THD PLL IPM DSP ADC EVA EVB : Pulse Width Modulation : Insulated Gate Bipolar Junction Transistor : Uninterruptible Power Supply : Root Mean Square : Electroagnetic Copability : Proportional Integral : Total Haronic Distortion : Phase Locked Loop : Intelligent Power Module : Digital Signal Processor : Analog to Digital Conversion : Event Manager A : Event Manager B vi

8 LIST OF TABLES Page Table 1.1 Advantages and disadvantages of haronic reduction ethods. 3 Table 1. Advantages and disadvantages of control ethods. 13 Table 3.1 Syste paraeters utilized in the siulation odel Table 3. Syste level seiconductor device odel paraeters.. 34 Table 3.3 Resonant filter paraeters.. 35 Table 3.4 Voltage and current controller paraeters of the three phase four wire voltage source PWM rectifier. 37 Table 3.5 Haronic contents of the voltage-sources used in the siulations 37 Table 3.6 Perforance coparison of three resonant filter fors under highly distorted utility. 43 Table 3.7 Highly distorted input voltage haronic content Table 3.8 Input current haronic content for the variable daped and phase copensated resonant filter for under highly distorted utility.. 44 Table 3.9 Input current haronic content for the constant daped and phase copensated resonant filter for under highly distorted utility.. 44 Table 3.10 Input current haronic content for the constant daped resonant filter for under highly distorted utility Table 3.11 Perforance coparison of three resonant filter fors under distorted utility Table 3.1 Distorted input voltage haronic content Table 3.13 Input current haronic content for the variable daped and phase copensated resonant filter for under distorted utility. 49 Table 3.14 Input current haronic content for the constant daped and phase copensated resonant filter for under distorted utility. 49 Table 3.15 Input current haronic content for the constant daped resonant filter for under distorted utility Table 3.16 Perforance coparison of three resonant filter fors under undistorted utility Table 3.17 Distorted input voltage haronic content Table 3.18 Input current haronic content for the variable daped and phase copensated resonant filter for under undistorted utility. 54 Table 3.19 Input current haronic content for the constant daped and phase copensated resonant filter for under undistorted utility. 54 Table 3.0 Input current haronic content for the constant daped resonant filter for under undistorted utility Table 3.1 Perforance coparison of three resonant filter fors.. 59 Table 3. Highly distorted 49.8 Hz input voltage haronic content Table 3.3 Input current haronic content for the variable daped and phase copensated resonant filter for under highly distorted 49.8 Hz utility vii

9 Table 3.4 Table 3.5 Input current haronic content for the constant daped and phase copensated resonant filter for under highly distorted 49.8 Hz utility Input current haronic content for the constant daped resonant filter for under highly distorted 49.8 Hz utility Table 4.1 Experiental syste paraeters. 6 Table 4. Haronic content of the distorted utility Table 4.3 Coparison of experiental and siulation results Table A.1 Basic specifications of the PM50CLA Table B.1 Main features of the ezdsp F81 board. 8 Table B. Main features of the TMS30F81 DSP... 8 viii

10 LIST OF FIGURES Figure 1.1 Figure 1. Figure 1.3 Figure 1.4 Figure 1.5 Figure 1.6 Figure 1.7 Figure 1.8 : Active shunt filter... : Three-phase current-source PWM rectifier... : Three-phase voltage-source PWM rectifier... : Operation principle of the three-phase voltage-source PWM rectifier... : PWM pattern... : Changing V MOD through the PWM pattern... : Four-quadrant operation of the voltage-source PWM rectifier... : Current wavefors through the ains, the IGBTs, and the DC link.... : DC link voltage of the voltage-source PWM rectifier : Basic schee of a linear rotating frae current regulator. Page Figure 1.9 Figure 1.10 Figure 1.11 : Basic schee of a digital deadbeat current regulator Figure 1.1 : Basic schee of a hysteresis current regulator.. 1 Figure.1 : Transforerless four-wire voltage-source PWM rectifier. 16 Figure. : Bode plot of the ideal resonant filter for =1, K i =0, ω e =π 50 rad/sec. 18 Figure.3 : The gain characteristic of a daped resonant filter... 1 Figure.4 : The gain and phase characteristics of the daped resonant filter for =1, K i =0, ω e =π 50 rad/s, τ = Figure.5 : The gain and phase characteristics of the constant daped resonant filter bank for = {1, 3, 5, 7, 9}, K i =0, ω e =π 50 rad/s, τ = Figure.6 : The gain and phase characteristics of the variable daped resonant filter bank for = {1, 3, 5, 7, 9}, K i =0, ω e =π 50 rad/s, τ = Figure.7 : The gain and phase characteristics of the constant daped and phase copensated resonant filter bank for ={1, 3, 5, 7, 9}, K i =0,ω e =π 50 rad/sec, τ =5 10-3, φ = T s ωe.. 5 Figure.8 : The gain and phase characteristics of the variable daped and phase copensated resonant filter bank for ={1, 3, 5, 7, 9}, K i =0,ω e =π 50 rad/sec, τ = , φ = T s ωe.. 5 Figure.9 : Single phase current control block diagra of the four-wire PWM rectifier. 7 Figure.10 : The gain and phase characterisitics of the ideal P+Resonant filter controller for K p =1, =1, K i =0, ω e =π 50 rad/sec 8 Figure.11 : The gain and phase characteristics of the daped P+Resonant filter controller for K p =1, =1, K i =0, ω e =π 50 rad/s, τ = ix

11 Figure.1 : The gain and phase characteristics of the constant daped and phase copensated P+resonant filter bank for ={1, 3, 5, 7, 9}, K i =0, K p =1, ω e =π 50 rad/sec, τ =5 10-3, φ = T s ωe... 9 Figure.13 : The gain and phase characteristics of the variable daped and phase copensated P+resonant filter bank for ={1, 3, 5, 7, 9}, K i =0, K p =1, ω e =π 50 rad/sec, τ =5 10-3, φ = T s ωe Figure.14 : The control syste block diagra of the four-wire voltage source PWM rectifier.. 31 Figure 3.1 : Power stage of the three-phase four-wire PWM rectifier.. 33 Figure 3. : Scheatic diagra of the three-phase four-wire PWM rectifier for adjusting voltage controller gains Figure 3.3 : Scheatic diagra of the three-phase four-wire PWM rectifier for adjusting current controller gains.. 36 Figure 3.4 : Steady state highly distorted input phase voltages Figure 3.5 : Dc link voltage and reference value for the variable daped and phase copensated resonant filter for under highly distorted utility 39 Figure 3.6 : Positive and negative half dc link voltage for the variable daped and phase copensated resonant filter for under highly distorted utility 39 Figure 3.7 : Input phase voltage and current at the instant of loading for the variable daped and phase copensated resonant filter for under highly distorted utility.. 40 Figure 3.8 : Dc link voltage and reference value for the constant daped and phase copensated resonant filter for under highly distorted utility 41 Figure 3.9 : Input phase voltage and current at the instant of loading for the constant daped and phase copensated resonant filter for under highly distorted utility.. 41 Figure 3.10 : Dc link voltage and reference value for the constant daped resonant filter for under highly distorted utility.. 4 Figure 3.11 : Input phase voltage and current at the instant of loading for the constant daped resonant filter for under highly distorted utility... 4 Figure 3.1 : Steady state distorted input phase voltages Figure 3.13 : Dc link voltage and reference value for the variable daped and phase copensated resonant filter for under distorted utility Figure 3.14 : Input phase voltage and current under full load for the variable daped and phase copensated resonant filter for under distorted utility Figure 3.15 : Dc link voltage and reference value for the constant daped and phase copensated resonant filter for under distorted utility Figure 3.16 : Input phase voltage and current under full load for the constant daped and phase copensated resonant filter for under distorted utility Figure 3.17 : Dc link voltage and reference value for the constant daped resonant filter for under distorted utility x

12 Figure 3.18 : Input phase voltage and current under full load for the constant daped resonant filter for under distorted utility Figure 3.19 : Steady state undistorted input phase voltages Figure 3.0 : Dc link voltage and reference value for the variable daped and phase copensated resonant filter for under undistorted utility Figure 3.1 : Input phase voltage and current at the instant of loading for the variable daped and phase copensated resonant filter for under undistorted utility.. 51 Figure 3. : Dc link voltage and reference value for the constant daped and phase copensated resonant filter for under undistorted utility Figure 3.3 : Input phase voltage and current at the instant of loading for the constant daped and phase copensated resonant filter for under undistorted utility.. 5 Figure 3.4 : Dc link voltage and reference value for the constant daped resonant filter for under undistorted utility Figure 3.5 : Input phase voltage and current at the instant of loading for the constant daped resonant filter for under undistorted utility. 53 Figure 3.6 : Steady state highly distorted 49.8 Hz input phase voltages Figure 3.7 : Dc link voltage and reference value for the variable daped and phase copensated resonant filter for under highly distorted 49.8 Hz utility Figure 3.8 : Input phase voltage and current at the instant of loading for the variable daped and phase copensated resonant filter for under highly distorted 49.8 Hz utility Figure 3.9 : Dc link voltage and reference value for the constant daped and phase copensated resonant filter for under highly distorted 49.8 Hz utility Figure 3.30 : Input phase voltage and current at the instant of loading for the constant daped and phase copensated resonant filter for under highly distorted 49.8 Hz utility. 57 Figure 3.31 : Dc link voltage and reference value for the constant daped resonant filter for under highly distorted 49.8 Hz utility. 58 Figure 3.3 : Input phase voltage and current at the instant of loading for the constant daped resonant filter for under highly distorted 49.8 Hz utility Figure 4.1 : The syste block diagra of the experiental set up... 6 Figure 4. : The electrical power circuitry of the overall syste... 6 Figure 4.3 : Main parts of the experiental set up Figure 4.4 : The filtering eleents of the experiental set up Figure 4.5 : Main board of the experiental syste Figure 4.6 : The flowchart of the DSP progra Figure 4.7 : Input currents and dc link voltage wavefors of the constant daped resonant filter bank case at the instant of full load transition Figure 4.8 : Steady state input currents and dc link voltage wavefors of the constant daped resonant filter bank case under full load Figure 4.9 : Steady state input current and voltage wavefors of the constant daped resonant filter bank case under full load xi

13 Figure 4.10 : Input currents and dc link voltage wavefors of the variable daped and phase copensated resonant filter bank case at the instant of full load transition Figure 4.11 : Steady state input currents and dc link voltage wavefors of the variable daped and phase copensated resonant filter bank case under full load Figure 4.1 : Steady state input current and voltage wavefors of the variable daped and phase copensated resonant filter bank case under full load operation... 7 Figure 4.13 : Input power and the power factor of the constant daped resonant filter bank case under full load operation Figure 4.14 : Input voltage haronic content of the constant daped resonant filter bank case under full load operation Figure 4.15 : Input current haronic content of the constant daped resonant filter bank case under full load operation Figure 4.16 : Input power and the power factor of the variable daped and phase copesated resonant filter bank case under full load operation Figure 4.17 : Input voltage haronic content of the variable daped and phase copensated resonant filter bank case under full load operation Figure 4.18 : Input current haronic content of the variable daped and phase copensated resonant filter bank case under full load operation xii

14 ÜÇ FAZ DÖRT İLETKENLİ DARBE GENİŞLİK MODÜLASYONLU DÜŞÜK AKIM HARMONİKLİ GERİLİM KAYNAĞI TİPİ DOĞRULTUCU ÖZET Yarı iletken elean teknolojisindeki gelişeler ile birlikte yüksek güçlü, darbe genişlik odülasyonlu, gerili kaynağı tipi doğrultucuların kullanıı giderek yaygınlaşaktadır. Düşük giriş akı haroniği, ayarlanabilir giriş güç faktörü, ve çift yönlü güç aktarıı özellikleri ile üç fazlı, dört iletkenli, darbe genişlik odülasyonlu, gerili kaynağı tipi doğrultucular, kesintisiz güç kaynağı ve otor sürücü uygulaalarında yaygın bir şekilde kullanılaktadır. Literatürde üç fazlı, dört iletkenli, darbe genişlik odülasyonlu, gerili kaynağı tipi bir doğrultucunun denetlenesini sağlayan çeşitli kontrol yönteleri yer alaktadır. Bu tez çalışasında, üç fazlı, dört iletkenli, darbe genişlik odülasyonlu, gerili kaynağı tipi bir doğrultucunun yüksek başarıla denetlenesini sağlayan kontrol yöntei geliştiriliştir. Bu yöntede, DC bara geriliini kontrol etek için gerili çevriinde, DC kazancı yüksek oransal-integral (PI) denetleyici kullanılaktadır. Akı çevriinde, her bir fazın akıını kontrol etek için durağan eksende teel bileşen ve haronik bileşenlerden oluşan rezonans süzgeç grubu kullanılaktadır. Rezonans süzgeçlerinin tasarıı ve uygulaa kolaylıkları ile ilgili detaylı bilgiler verilektedir. Akı kontrolünün dinaik başarıını arttırak aacı ile rezonans süzgeç grubuna paralel bağlı orantısal bir kazanç ekleniştir. Ayrıca, şebeke gerilii ileri besleesi kullanılarak sistein durağan ve dinaik başarıı iyileştirilektedir. Üç fazlı, dört iletkenli, darbe genişlik odülasyonlu, gerili kaynağı tipi doğrultucunun durağan ve dinaik başarıı farklı çalışa koşulları için ayrıntılı olarak inceleniştir. Deneti yönteinin başarıı teori, bilgisayarla benzeti ve deneysel çalışalarla doğrulanıştır. xiii

15 THREE PHASE FOUR WIRE VOLTAGE SOURCE PWM RECTIFIER WITH LOW INPUT CURRENT HARMONIC SUMMARY As a result of treendous developents in the seiconductor industry, voltage source PWM rectifiers are becoing cheaper and available at increased power levels. Due to its several advantages such as low input current haronic, adjustable input power factor, and four quadrant operation, PWM rectifiers are widely used in variable-speed drives and uninterruptible power supplies. There exist several control ethods for three-phase four-wire voltage-source PWM rectifiers in the literature. In this study, a new control ethod is proposed for high perforance operation of three-phase four-wire voltage-source PWM rectifiers. In this ethod, a PI type controller with high dc gain is used for regulating the DC bus voltage. For controlling the input currents of each phase resonant banks are used. Resonant banks are coposed of parallel connected resonant filters for fundaental and haronic coponents. Detailed inforations for designing optiu resonant filters are given in the thesis. In order to iprove the dynaic and steady-state perforance of the current controller, a proportional gain is connected parallel to the resonant filter bank. Also, input voltage feedforward is used for iproving the dynaic and steady-state perforance of the rectifier. The steady-state and dynaic perforance of the three-phase four-wire voltagesource PWM rectifier is tested under different utility conditions. The proposed control ethod is proven by eans of theory, siulations, and experients. xiv

16 1. INTRODUCTION 1.1. Rectifiers and Electric Power Quality In this thesis, three-phase four-wire voltage-source PWM rectifiers are investigated. As a result of treendous developents in the seiconductor industry, PWM rectifiers are becoing cheaper and available at increased power levels. Today, PWM rectifiers are widely used in industrial applications, such as variable-speed drives and uninterruptible power supplies (UPSs). This is advantageous because using power electronic equipents result in high efficient and high perforance operation. However, using the power electronics starts a new dilea. Since all power electronic circuits behave as nonlinear loads, haronic currents are injected into the grid. Most of the power electronic equipents are a source of current haronics, which results in increase in reactive power and power losses in transission lines. The haronics also cause electroagnetic interference and, soeties dangerous resonances. They have negative influence on the control and autoatic equipent, protection systes, and other electrical loads, resulting in reduced reliability and availability. Moreover, nonlinear loads and nonsinusoidal currents produce nonsinusoidal voltage drops across the network ipedances, so that nonsinusoidal voltages appear at several points of the ains. It results in overheating of transission line, transforers and generators due to the increased copper losses. 1.. Haronic Reduction Methods Reduction of haronic content in line current to a few percent allows avoiding ost of the entioned probles above. Restrictions on current and voltage haronics aintained in any countries through IEEE and IEC /IEC standards, are associated with the popular idea of clean power. Today, several techniques are used for reducing the line side haronics. The ost popular techniques used for reducing haronics are; 1

17 Adding Passive Filters Using Multipulse Rectifiers Using Active Filters Using PWM Rectifiers The traditional ethod of current haronic reduction involves adding passive LC filters. These filters are connected parallel to the grid. Filters are usually constructed as series-connected legs of capacitors and inductors. The nuber of legs depends on nuber of filtered haronics (5 th, 7 th, 11 th, and 13th). The ain advantages of passive filters are their siplicity and low cost. On the other hand, it has any disadvantages. These filters are designed for a particular application, and the filter eleents are heavy and bulky. There exists a risk of resonance proble at the grid. Beside, these filters consue reactive power which results in extra cost for the user. Multipulse rectifiers are also used for reducing haronics. Although it is easy to ipleent, it possesses several disadvantages such as, bulky and heavy transforer, increased voltage drop, and increased haronic currents at non-syetrical load or line voltages. Active filters, Figure 1.1, are used as a better alternative of the passive filters. They have better dynaics responses and they can control the haronic and fundaental currents. Figure 1.1: Active Filter Active filters provide copensation of fundaental reactive coponents of load current, load syetrization, fro grid point of view, and haronic copensation

18 uch better than passive filters. In spite of its advantages, active filters possess certain disadvantages such as, coplex control, switching losses and EMC probles. PWM rectifiers are the ost effective way of reducing line side haronics. As a result of treendous developents in the seiconductor industry, they are becoing cheaper and available at increased power levels. Table 1.1 Advantages and disadvantages of haronic reduction ethods 1.3. General Introduction to Three Phase PWM Rectifiers During the past twenty years, the interest in rectifying units has been rapidly growing ainly due to the increasing concern of the electric utilities and end users about the haronic pollution in the power syste. As a result, PWM rectifiers have been of particular interest and they have becoe attractive especially in industrial variable speed drive and UPS applications in the power range fro a couple of kilowatts up to several egawatts Basic topologies and characteristics PWM rectifiers are built with seiconductors with gate-turn-off capability. The gateturn-off capability allows full control of the rectifier, because switches can be switched ON and OFF whenever it is required. This allows the coutation of the switches hundreds of ties in one period, which is not possible with line coutated rectifiers, where thyristors are switched ON and OFF only once a cycle. This feature has the following advantages; 3

19 the current or voltage can be odulated (Pulse Width Modulation or PWM), generating less haronic contents power factor can be controlled, and even it can be ade leading they can be built as voltage-source or current-source rectifiers the reversal of power in thyristor rectifiers is by reversal of voltage at the dc bus, on the other hand, PWM rectifiers can be ipleented for both, reversal of voltage or reversal of current. There are two ways to ipleent three-phase PWM rectifiers; as a current-source rectifier, where power reversal is obtained by DC voltage reversal as a voltage-source rectifier, where power reversal is obtained by current reversal at the dc bus Figure 1. and Figure 1.3 shows the basic circuits for these two topologies. Figure 1.: Three-Phase Current-Source PWM Rectifier Figure 1.3: Three-Phase Voltage-Source PWM Rectifier 4

20 1.3.. Operation of the Voltage Source PWM rectifier The voltage-source rectifier is by far, the ost widely used type of PWM rectifiers. The basic operation principle of the voltage-source rectifier consists on keeping the DC bus voltage at a desired reference value, using a feedback control loop as shown in Figure 1.4. To accoplish this task, the DC bus voltage is easured and copared with a DC voltage reference VREF. Figure 1.4: Operation Principle of the Three-Phase Voltage-Source PWM Rectifier When the current I D is positive, PWM rectifier is in rectifier operation. In this ode of operation the dc bus capacitor C D is discharged due to the positive I D, and the error signal ask the Control Block for ore power fro the AC supply. Inversely, when I D becoes negative (inverter operation), the capacitor C D is overcharged, and the error signal ask the control to discharge the capacitor and return power to the AC supply. The Pulse Width Modulation consists on switching the switches ON and OFF, following a preestablished teplate. Particularly, this teplate could be a sinusoidal wavefor of voltage or current. The PWM pattern has a fundaental signal V MOD, with the sae frequency of the power source, so the rectifier works properly. Changing the aplitude of this fundaental, and its phase shift with respect to the ains, the rectifier can be controlled to operate in the four quadrants. For exaple, the odulation of one phase could be as the one shown in Figure 1.5. The aplitude of the V MOD in Figure 1.5 is proportional to the aplitude of the teplate. 5

21 Figure 1.5: PWM pattern Figure 1.6: Changing V MOD through the PWM pattern The interaction between V MOD and V (source voltage) can be seen through a phasor diagra. This interaction perits to understand the four-quadrant capability of this rectifier. In the Figure 1.7, the four-quadrant operation is clearly explained. I S in Figure 1.7 flows through the seiconductors in the way shown in Figure 1.8. During the positive half cycle, the transistor T N, connected at the negative side of the DC bus is switched ON, and the current i s begins to flow through T N (i Tn ). The current returns to the ains and coes back to the switches, closing a loop with another phase, and passing through a diode connected at the sae negative terinal of the DC bus. The current can also go to the DC load (inversion) and return through another transistor located at the positive terinal of the dc bus. 6

22 Figure 1.7: Four-quadrant operation of the voltage-source PWM rectifier a) voltage-source PWM rectifier b) rectifier operation at unity power factor c) inverter operation at unity power factor d) capacitor operation at zero power factor e) inductor operation at zero power factor 7

23 When the transistor T N is switched OFF, the current path is interrupted, and the current begins to flow through the diode D P, connected at the positive terinal of the DC bus. This current, called i Dp in Figure 1.8, goes directly to the DC bus, helping in the generation of the current i dc. The current i dc charges the capacitor C D and perits the rectifier to produce DC power. The inductances L S are very iportant in this process, because they generate an induced voltage which allows the conduction of the diode D P. Siilar operation occurs during the negative half cycle, but with T P and D N, Figure 1.8. Figure 1.8: Current wavefors through the ains, the IGBTs, and the DC bus To have full control of the operation of the rectifier, six antiparallel connected diodes ust be polarized negatively at all values of instantaneous AC voltage supply. Otherwise diodes will conduct, and the PWM rectifier will behave like a coon diode rectifier bridge. The way to keep the diodes blocked is by ensuring a DC bus voltage higher than the peak DC voltage generated by the diodes alone, as shown in Figure 1.9. In this way, the diodes reain polarized negatively, and they only will conduct when at least one transistor is switched ON, and favorable instantaneous AC voltage conditions are given. In the Figure 1.9 V D represents the capacitor DC voltage, which is kept higher than the noral diode bridge rectification value V BRIDGE. 8

24 Figure 1.9: DC bus voltage of the Voltage-source PWM rectifier Control Methods Voltage-source PWM rectifiers allow a full control over both active and reactive power exchanges between the AC ains and DC source. Different control techniques have been discussed to shape the input current wavefors of the voltage-source PWM rectifiers Linear Current Control The conventional version of the linear current controller perfors a sine, triangle PWM voltage odulation technique. Linear current control technique provides an unsatisfactory perforance level as far as PWM rectifier applications are concerned. This is ainly due to the liitation of the achievable regulator bandwidth which is iplied by the necessity of sufficiently filtering the ripple in the odulating signal. This necessity copels one to keep the loop gain crossover frequency well below the odulation frequency. This reflects in a poor rejection of the disturbances injected into the current control loop, ainly due to the AC line voltage at the fundaental frequency. To overcoe this liitation, recent versions of the linear current controllers eploy reference frae transforations [1-3]. Control variables are transfored into the rotating frae according to the schee represented in Figure The ain advantage of such a solution is that the fundaental haronic coponents of voltage and current signals appear constant to the current regulator. As a consequence, the rejection of this disturbance is uch ore effective. On the other hand, the bandwidth liitation of the PI regulators, which reains unchanged, 9

25 still iplies significant errors in the tracking of the high order haronic coponents of the current reference. Figure 1.10: Basic schee of a linear rotating frae current regulator [3] Deadbeat Control The deadbeat control ethod can only be ipleented on a digitally controlled syste. In order to apply the deadbeat control, the atheatical odel of the syste ust be known. Using the reference and feedback signals, and eploying the syste odel, the control signal that forces the input current error to zero in finite nuber of sapling cycles is calculated and applied to the odulator. When the syste odel is exactly known, the error due to changes in the state variables is driven to zero in ideally one step, but in practical case it takes two steps. So, the deadbeat controller has a very fast response and high control bandwith. However, the deadbeat control ethod, which is dynaically very stable for a well defined syste, is highly affected by the syste paraeters. Even a sall change in the paraeters can ake the syste unstable. Beside of this, coputational and easureent delays effects the control perforance greatly. Soe techniques are given in the literature for copensating the coputational delays [4-6]. The basic schee of a deadbeat current regulator can be seen in Figure

26 Figure 1.11: Basic schee of a digital deadbeat current regulator [4] Hysteresis Control The ain goal of the hysteresis control is keeping the current error in a specified hyteresis band. In spite of its siplicity, good accuracy and high robustness, this control technique exhibits several unsatisfactory features [7]. The ain one is that it produces a varying odulation frequency for the power converter. Many iproveents to the original control structure have been suggested by industrial applications [8-9]. First of all, phase current decoupling techniques have been devised [10]. Secondly, fixed odulation frequency has been achieved by a variable width of the hysteresis band as function of the instantaneous input current [11]. Figure 1.1 shows the siplified schee of the ipleentation of such a controller. As can be seen, the controller odifies the hysteresis band by suing two different signals. The first is the filtered output of a PLL phase coparator β 1, and the second is the filtered output of a band estiation circuit β. The band estiator ipleents a feedforward action that helps the phase locked loop, PLL, based circuit to keep the switching frequency constant, in this way, the output of the PLL circuit only provides the sall aount of the odulation of the hysteresis band which is needed to guarantee the phase lock of the switching pulses with respect to an external clock signal. This also ensures the control of the utual phase of the odulation pulses. All of these provisions have allowed a substantial iproveent in the perforance of the hysteresis current controller, as is discussed in [1]. 11

27 Figure 1.1: Basic schee of a hysteresis current regulator [1] Resonant Filter Bank Controller Resonant filters have recently reeerged as a focus in the literature with the recognition that any rotating frae controllers can be transfored to an equivalent stationary frae syste. This reoves the need for rotating frae transforations and sine tables, [13], and has led to reduced coplexity for applications such as current regulators, [13-14], active filters, [15-16], and UPS systes [17]. The ethod is based on copensator assignent for each frequency of interest. Thus, the controller has a parallel structure. Although various copensator types are possible, the type shown in the following equation is ost widely utilized [18-1]. Ki s GC ( s) = (1.1) s + ( ω ) e In the resonant filter copensator of Eq. 1.1, K i is the integral gain, and ω e is the frequency of interest where infinite gain is deanded. If the resonant filter controller is tuned at the fundaental frequency (=1), due to infinite gain and zero phase at the fundaental frequency, the steady state error will be zero at fundaental frequency. As the controller has zero gain at all other frequencies, the controller does not influence other frequency coponents. Thus, for each frequency of interest a resonant filter at that frequency should be utilized. 1

28 Because of its siplicity, perfect accuracy and high robustness, in this thesis, resonant filter banks are used for input current regulation of Three-Phase Four-Wire Voltage-Source PWM Rectifier. As the ethod does not require positive, negative, zero sequence separation and it is easy to ipleent, it has been found favorable over other ethods briefly discussed above. Table 1.1: Advantages and disadvantages of control ethods 1.4. Outline of the Thesis In this thesis, the Three-Phase Four-Wire Voltage-source PWM Rectifier syste is investigated in detail. In this topology, neutral line of the supply is directly connected to the centre point of the DC bus using centre tapped capacitors. With this property, this topology can be easily used in a three-phase transforerless UPS syste. The neutral connection in the topology decouples the three phases and allows individual control of each phase.with the ipleentation of the resonant type filters in the current controller, the current tracking perforance of the voltage-source PWM rectifier becoes perfect. Designing the resonant filter bank with proper resonant filter coponents such as the fundaental coponent, 3 rd, 5 th, 7 th, 9 th, 11 th, etc. the input current coposed of ainly with the desired fundaental coponent, and less haronic coponent, even if the input voltage is highly distorted. All these control algoriths have been ipleented on a 15 kva syte using a digital signal processor, and the theory has been verified experientally. The organization of the thesis is given as follows; In the second chapter, the input current control of the four-wire voltage-source PWM rectifier is explained clearly. The iportant points in designing the optiu resonant 13

29 filter for that can achieve high perforance are clarified. Later on, the ipleentation of these resonant filters in a fixed point digital signal processor is explained. In the third chapter, the detailed odelling and coputer siulation of a 15 kva, 50 Hz, four-wire PWM rectifier is provided. Following the establishent of the siulation odel, the controller tuning procedure is described. Then the rectifier control perforance is investigated for steady state and dynaic operating conditions. The superior steady state and dynaic perforance of the proposed control ethod is illustrated by eans of coputer siulations involving steady state and dynaic loading conditions. The fourth chapter of the thesis explains the hardware setup and the experiental studies of the 15 kva syste. In this chapter the experiental syste hardware and software are discussed in detail. The steady state perforance under various load conditions and dynaic perforance under loading transient conditions are shown via laboratory experients. Correlation with the coputer siulation and experiental results are provided. The fifth chapter suarizes the research results, and concludes the thesis. Finally, recoendations for future work on the study subject of this thesis are given. 14

30 . RESONANT FILTER BASED INPUT CURRENT CONTROL OF THE THREE PHASE FOUR WIRE VOLTAGE SOURCE PWM RECTIFIER.1. Introduction Three-phase AC-DC-AC converters have been used for any years in UPSs and induction otor drives. Three-phase induction otors ay be driven fro a three wire source and assuing galvanic isolation is not required, a transforerless three wire AC-DC-AC converter is coonly used. On the other hand, a double conversion UPS can operate with a three wire input rectifier, but ust have a fourwire output. A three-phase transforer is needed on the front or back end of the converter even when galvanic isolation is not required. The load neutral connection of the UPS is directly connected to the star point of the supply thus preventing the load neutral fro floating and providing a path for fault currents to flow via the earth connection. The presence of the three-phase transforer increases the cost, size and the weight of the UPS. Reoval of the transforer in a three-phase UPS would lead to copact converters with significant savings in weight and cost. However, reoval of the transforer would lead to haronics flowing in the supply neutral, ainly third haronic of the supply frequency. If a four-wire PWM rectifier is used, the transforer can be reoved while allowing sinusoidal currents to be drawn fro the ains at unity power factor. This chapter focuses on the input current control of the four-wire PWM rectifier. The current controller of each phase is coposed of parallely connected resonant filters, one for fundaental, and the others for haronic coponents. In the voltage controller, a PI type controller is used. With this controller structure the four-wire PWM rectifier exhibits superior steady state and dynaic perforance under all practical operating conditions. 15

31 InpR L1 L4 R1 C4 InpS L L5 R InpT L3 L6 R3 C5 C 10u C1 C C3 R4 R5 R6 Figure.1: Transforerless four-wire voltage-source PWM rectifier [0].. Resonant Filter Fors A resonant filter consists of a transfer function which has very large gain and zero phase delay at the desired frequency. The desired frequency is called as the resonant frequency. Frequencies other than the resonant frequency, the gain of the transfer function becoes very sall and the phase angle becoes negligible. The siple linear proportional integral, PI, type controllers are prone to known drawbacks including the presence of steady state error in the stationary frae and the need to decouple phase dependency in three-phase systes altough they are relatively easy to ipleent []. Exploring the siplicity of PI controllers and to iprove their overall perforance, any variations have been proposed in the literature including the addition of a grid voltage feedforward path, ultiple state feedbacks and so on. Generally, these variations can expand the PI controller bandwith, but unfortunately they also push the syste towards their stability liits. Alternatively for three-phase systes, synchronous frae PI control with voltage feedforward can be used, but it usually requires ultiple frae transforations. Overcoing the coputational burden and still achieving virtually siilar frequency response characteristics as a synchronous frae PI controller, resonant controllers are eployed for reference tracking in the stationary frae [14, 1]. Resonant controllers are conceptually siilar to an integrator whose infinite DC gain forces the DC steady state error to zero. 16

32 ..1. Ideal Resonant Filter In the three-phase four-wire voltage-source PWM rectifier application, the resonant filters are eployed in the current controller. It is constructed in the stationary frae (without coordinate transforations) and has a superior perforance siilar to the linear PI regulator in synchronous frae [14]. The ideal resonant filter has a transfer function as given in Eq..1. Ki s GC ( s) = (.1) s + ( ω ) e In Eq..1, K i is the integral gain, and ω e is the frequency of interest where infinite gain is deanded. If the resonant filter controller is tuned at the fundaental frequency (=1), due to infinite gain and zero phase at the fundaental frequency, the steady state error will be zero at fundaental frequency. It can be atheatically derived by transforing a synchronous frae PI controller to the stationary frae without consideration of the redundant cross coupling ters, and has an infinite gain at the controller s resonant frequency ω e, which is chosen to be the line fundaental frequency (π 50 rad/s). Transforing PI controllers in both positive and negative sequence synchronous fraes of a three-phase syste to the stationary frae, using either frequency doain or tie doain technique, the final stationary controller of Eq..1 can be obtained, and the cross coupling ters generated fro positive and negative sequence synchronous fraes would cancel each other if the sae PI paraeters are eployed in both synchronous fraes. Therefore, the resonant filter would achieve zero steady state error for both positive and negative sequence coponent regulations in principle with infinite gains at fundaental frequency. The ideal resonant filter has an infinite gain and zero phase at the resonant frequency (ω e ). A typical bode plot for the ideal resonant filter is shown in Figure.. 17

33 Figure.: Bode plot of the ideal resonant filter for =1, K i =0, ω e =π 50 rad/sec The phase relation is such that below the resonant frequency the resonant filter provides 90 o leading copensation while above the resonant frequency the filter provides 90 o lagging copensation. The resonant frequency is selected to be the frequency of the AC signal to be controlled. The gain is optiized by considering the syste dynaic and steady state perforance requireents. For instance, if a resonant filter is used at the fundaental frequency for controlling the input current of a three-phase rectifier, the resonant filter controller precisely controls the fundaental coponent of the current at the desired phase and agnitude. However, the controller can not copensate the haronic current coponents other than the fundaental frequency, because the controller provides nearly zero gain for these coponents. This proble can be solved by using resonant filters for each doinant haronic frequency. Thus, depending on the existing haronic coponents, the resonant frequency ultiplier is selected. Resonant filter banks are fored by connecting resonant filters at the required frequency in parallel. In the three-phase four-wire voltage-source PWM rectifier application, due to the occurrance of the fourth wire, neutral wire, each phase can be controlled independently. The outer control loop consists of a PI type DC bus voltage 18

34 controller. The output of the voltage controller is ultiplied with the sine references for obtaining the current references of each phase. For each phase, one identical resonant filter bank is utilized. Thus, the fundaental coponent is controlled without involving positive-negative-zero sequence decoposition and coplex controllers which is the conventional approach [3]. The resonant filter controller provides an easy design and ipleentation task with respect to the synchronous frae based controller or other coplex control algoriths.... Phase Delay Copensated Resonant Filter Although theoretically, the ideal resonant filter would achieve zero steady state error at the resonant frequency, there could be practical probles during its ipleentation as it is sensitive to syste delays and frequency variations. The practical ipleentation of the resonant filters involves ore detailed structure than the ideal structure given in Eq..1. If the syste delay, which is the total delay originating fro the individual delays of the rectifier syste stages, is ignored, the controller perforance decreases especially at high frequencies. Practically, the easureent, signal processing, and the PWM units introduce delays that can have significant influence on the controller perforance. The easureent delay (τ ea ) occurs at the stage of the easureent and signal conditioning of the voltage and current feedback signals. The sapling delay (τ sap ) is ainly due to the A/D conversion tie. The PWM delay (τ PWM ) is due to the discrete nature of the rectifier. The su of all these delays is called as the total delay τ T. τ τ + τ + τ T = (.) ea sap PWM This total delay results in a phase shift, φ which varies with the frequency. It can be obtained as a function of the total syste delay and the resonant frequency (ω e ) of the controller, as given in Eq..3. Since the phase shift is directly proportional to the frequency, it is uch ore iportant at high frequencies. The ideal transfer function of the resonant filter, given in Eq..1 should be odified such that the effect of delay is copensated, the control signals should be phase advanced by φ. Since Eq..1 is the Laplace transfor of cos (ω e t), the phase shift should be considered as a process of adding a phase advance angle to the resonant ter, iplying the cosine function angle should be advanced. Thus the delay copensated tie doain 19

35 equivalent of the resonant filter is cos (ω e t + φ ) and the Laplace transfor of the phase advanced cosine ter shown in Eq..4 gives the odified resonant filter as in Eq..5. In this anner, the transfer function of the phase shifted resonant filter controller is obtained. φ = τ ω (.3) T e L s cos( φ { ( )} ) ωe sin( φ ) K cos + = i et φ Ki s + ( ωe) ω (.4) G ( s) C s cos( φ ) ωe sin( φ ) s + ( ωe ) = Ki (.5)..3. Daped Resonant Filter The odified transfer function of the resonant filter given in Eq..5 can fix the probles due to the delays in the total syste, but it still involves difficulties both fro perforance and ipleentation point of view. Since the above for is a lossless resonant filter, the gain of the filter at the resonant frequency is very high and the sidebands are very sall. This iplies that the controller is highly selective and can only track a reference exactly at the designed value of ω e. However, in certain applications the frequency of the reference signal is not constant. In such a case, the gain and the phase of the resonant filter changes very rapidly, which results in instabilities in the controller perforance. For the four-wire PWM rectifier application, the input voltage frequency can change in a wide range. Typically for a three-phase rectifier, the input voltage frequency variation liit is ±.5 Hz for 50 Hz utility grid. So the fundaental frequency of the control signal applied to the resonant filter can change a lot. Therefore, the bandwidth of the resonant filter should be widened for iproved tracking over a specified frequency range. Beside of the entioned practical probles above, ipleentation of a resonant filter with a fixed point processor is another big proble. The coefficients of a lossless discrete tie resonant controller ay becoe extree values, soe of the coefficients approaches to zero and soe others approaches to one. Since a nuber can be represented with a finite word length in a fixed point processor, the coefficients of the discrete tie ipleented filter becoe an issue as loss of significance occurs in ipleenting the resonant controller. In a fixed point 0

36 processor soe sall nubers are lost. As a result, the gain and the agnitude of the resonant controller becoe significantly different than the evaluated values. The resonant filter fors in Eq..1 or Eq..5 can not be used because of the above entioned probles. The original resonant filter for is odified to solve these probles. The daping of the resonant filter is increased. The new for of the daped resonant filter is given in Eq..6. In this equation τ is the daping constant and ω e is the resonant frequency of the daped resonant filter. G C K i τ ωe s ( s) = (.6) s + τ ω s + ( ω ) e e Selectivity function, Eq..7, is defined in [4] for describing the daping. In this equation the resonant frequency is ω e and the corner frequency where the gain drops to 70.7 % is ω e ±0.5 ω as shown in Figure.3 which is drawn by utilizing Eq..7. The gain curve of the resonant filter is practically syetric with respect to the resonant frequency ω e and the gain at ω e ω and ω e ω is practically the sae. Figure.3: The gain characteristic of a daped resonant filter resonant frequency ωe S = = (.7) bandwidth ω The resonant filter daping τ is related to the resonant filter corner frequency. 1

37 ω = τ ω (.8) e The selectivity can be expressed in ters of the filter daping by subtituting Eq..8 in Eq S = τ (.9) Coparing the ideal resonant filter of Figure Eq..1 with a resonant filter with the daping value of of which the gain characteristic is shown in Figure.4, it can be seen that both the gain and selectivity decrease significantly. However, the filter becoes ore effective over a wider operating frequency range. In order to copensate for the gain loss at the resonant frequency, the integral gain should be increased to a sufficient value. Figure.4: The gain and phase characteristics of the daped resonant filter for =1, K i =0, ω e =π 50 rad/s, τ = Resonant Filter Bank Fors.3.1. Daped Resonant Filter Banks In the three-phase four-wire PWM rectifier application, current controller contains a fundaental frequency resonant filter and a specific nuber of haronic frequency

38 resonant filters in each phase. These resonant filters are parallel connected to each other and they fored the resonant filter bank. The outputs of these resonant filter banks produce pw signal for each phase. The daping coefficients of the resonant filters in the resonant filter bank ay be equal or different than each other. Figure.5 shows the daped resonant filter bank gain and phase characteristics for the case that the daping coefficient reains the sae for all the frequencies. This iplies that for increasing frequencies the frequency sideband reains the sae. For this case, the higher frequency resonant filters becoe highly selective and in practice if the easured haronics are not exactly at the selected resonant frequency, the filter perforance becoes unsatisfactory. On the contrary, for the case where the daping coefficients increases with the increasing filter frequency, the filter perforance does not affected by the frequency. Figure.6 shows the resonant filter bank with the daping coefficient equals to ( τ). Figure.5: The gain and phase characteristics of the constant daped resonant filter bank for ={1, 3, 5, 7, 9}, K i =0, ω e =π 50 rad/s, τ =

39 Figure.6: The gain and phase characteristics of the variable daped resonant filter bank for ={1, 3, 5, 7, 9}, K i =0, ω e =π 50 rad/s, τ = Phase Delay Copensated and Daped Resonant Filter Banks The final transfer function of the resonant filter can be obtained by considering the phase delay copensation and the daping in a single filter for. In practical applications τ is sall and typically τ <<1. With this assuption, if the equations Eq..5 and Eq..6 are odified, the final transfer function of the resonant filter is obtained, Eq..10. G C ( s cos( φ ) ω sin( φ )) K i τ ωe e ( s) = (.10) s + τ ω s + ( ω ) e e The transfer function in Eq..10 is the practically necessary for of a resonant filter. The phase copensation angle of the filter φ ust be equal to the phase delay of the syste and the daping constant τ is chosen by considering the syste requireents. These paraeters, φ and τ, are the ain paraeters of the resonant filter that akes it practically applicable. Eq..10 is a basic practical resonant filter structure and its five paraeters K i,, ω e, τ, and φ are the variables to be utilized in shaping the filter characteristics in the input current controller design procedure of the four-wire PWM rectifier. Figure.7 shows the resonant filter bank gain and 4

40 phase characteristics for the constant daped and phase copensated case. Whereas, Figure.8 shows the variable daped and phase copensated case. Eq..10 is the final for of the resonant filter structure which will be utilized in this thesis for the purpose of controlling the input current of the three-phase four-wire PWM rectifier. The controller will be ipleented with a digital signal processor, DSP. Thus, the filter structure ust be converted to the discrete tie for which involves the discrete doain rather than continuous doain. Figure.7: The gain and phase characteristics of the constant daped and phase copensated resonant filter bank for = {1, 3, 5, 7, 9}, K i =0, ω e =π 50 rad/sec, τ = , φ = T s ω e Figure.8: The gain and phase characteristics of the variable daped and phase copensated resonant filter bank for = {1, 3, 5, 7, 9}, K i =0, ω e =π 50 rad/sec, τ = , φ = T s ω e 5

41 6.4. Discrete Tie Ipleentation of Resonant Filter Banks Usually controllers are designed in the continuous doain for the continuous tie systes. Therefore, it is coon to design the resonant filter in the continuous doain. However, in order to adapt the controller to a digital signal processor, the controller structure should be expressed in the discrete doain where the discrete tie odel easily leads to discrete tie equations to be utilized in the real tie ipleentation of the controller. There are several ethods to provide transforation between continuous doain and discrete doain. Due to its easiness and high accuracy, the Tustin transforation ethod, Eq..11, will be used in the transforations. In the transforation, the effect of warping is copensated for via the prewarping coefficient A which is defined in Eq..1. The prewarping coefficient iproves the accuracy of the transforation. By utilizing the Tustin transforation, Eq..10 is transfored to Eq..13 where X(z) is the input of the resonant filter and Y(z) is the output of the resonant filter. The coefficients of Eq..13 are given in Eq..14 through Eq..18. A z z s + = 1 1 (.11) = tan S e e T A ω ω (.1) ) ( ) ( ) ( = = z b z b z a z a a z X z Y z G C (.13) ( ) 0 ) ( ) sin( ) cos( e e e e i A A A K a ω ω τ φ ω φ ω τ + + = (.14) 1 ) ( ) sin( 4 e e e e i A A K a ω ω τ φ ω ω τ + + = (.15) ( ) ) ( ) sin( ) cos( e e e e i A A A K a ω ω τ φ ω φ ω τ = (.16) ( ) 1 ) ( ) ( e e e A A A b ω ω τ ω + + = (.17)

42 b A τ ωe A + ( ωe ) = (.18) A + τ ω A + ( ω ) e e If the discrete doain transfer function of the resonant filter given in Eq..11 is transfored to discrete tie equations, the discrete response of the filter as shown in Eq..19 is obtained. In Eq..19, x[k] represents the output voltage error at the kt s interval, and y [k+1] represents the th resonant controller output to be applied to the PWM unit at the (k+1)t s interval. y [ k + ] = a x[ k] + a x[ k 1] + a x[ k ] b y [ k] b y [ k 1] (.19).5. P+Resonant Controller In order to iprove the rectifier current control loop bandwidth and dynaic perforance, a proportional feedback control loop can be added to the syste. The proportional controller is connected in parallel with the resonant filter bank. The total feedback control structure becoes one shown in Figure.9. The current error signal, which is the difference between the current reference produced by the DC bus voltage controller and the easured current, is ultiplied with a proportional gain (K p ) and added to the rectifier reference signal as shown in Figure.9. The added proportional gain ainly iproves the rectifier perforance during loading transients where the current error is large. Additionally, it enhances the current controller bandwidth. By using the proportional ter, current controller can copensates for the current haronic coponents higher than the largest frequency coponent of the resonant filter bank. As a result, the perforance of the rectifier iproves and the input current THD reduces. Figure.9: Single phase current control block diagra of the four-wire PWM rectifier 7

43 Since the proportional control eleent is in parallel with the resonant filter bank, the gain and phase characteristics of the resonant filter provided in the previous section are odified. The gain and phase characteristic of the ideal P+Resonant controller is given in Figure.10. The gain characteristics of both the resonant filter and the P+Resonant filter are the sae, but the phase characteristics are different. Figure.11 shows the gain and phase characteristics of the daped resonant filter with proportional gain. Siilar to the ideal resonant filter case, in the daped resonant filter case, the proportional gain iproves the phase characteristic of the controller. The proportional gain also iproves the resonant controller gain (K i +K p ). Figure.10: The gain and phase characteristics of the ideal P+resonant filter controller for K p =1, =1, K i =0, ω e =π 50 rad/sec 8

44 Figure.11: The gain and phase characteristics of the daped P+Resonant filter controller for K p =1, =1, K i =0, ω e =π 50 rad/s, τ 1 = The odified controller gain and phase characteristics of the P+resonant filter bank can be seen in Figure.1. The gain characteristics iprove slightly at higher frequencies. On the other hand the phase characteristic of the new structure changes significantly. The phase characteristic of the forer figures vary between -180 o and 180 o while those of the latter vary between -90 o and 90 o. Figure.1: The gain and phase characteristics of the constant daped and phase copensated P+resonant filter bank for = {1, 3, 5, 7, 9}, K i =0, K p =1, ω e =π 50 rad/sec, τ = , φ = T s ω e 9

45 Figure.13: The gain and phase characteristics of the variable daped and phase copensated P+resonant filter bank for = {1, 3, 5, 7, 9}, K i =0, K p =1, ω e =π 50 rad/sec, τ = , φ = T s ω e.6. The Full Control Structure of the Four Wire PWM Rectifier In this thesis the three-phase four-wire voltage-source PWM rectifier is controlled by eploying cascade controller structure. The outer loop is the voltage loop controller. The voltage controller is chosen as a PI type controller. Since the input variables of the voltage controller are DC signals, a PI type controller achieves high perforance. The inner control loop is the current controller. P+resonant filter banks are used in the current controller. Also an input voltage feedforward signal is added for iproved tracking perforance. The control syste block diagra is shown in Figure.14 in detail. In the diagra the control structure is priarily shown for one phase and the other phases also have identical controller structure. 30

46 Current Reference Figure.14: The control syste block diagra of the four-wire voltage-source PWM rectifier 31

47 3. SIMULATION RESULTS 3.1. Introduction In this chapter, the three-phase four-wire voltage-source pw rectifier with the control ethod proposed in Chapter is investigated by eans of coputer siulation. A three-phase four-wire rectifier syste with the power rating of 15 kva is odeled in the siulation software. The supply voltage of the three-phase rectifier is 0V and 50Hz. The DC bus voltage produced by the rectifier is chosen as 750 Vdc. Firstly the power stage of the rectifier is odeled in the siulation, and then the proposed control algoriths are ipleented. The paraeters of the resonant filter bank and the PI type voltage controller are tuned by trial and error. Using this odel, steady state and dynaic perforance of the three-phase rectifier is investigated under different operating conditions. 3.. Modeling of the Three Phase Four Wire Voltage Source PWM Rectifier Syste The syste is odeled by using the Ansoft-Siplorer coputer siulation package progra [5]. Because of its flexibility in connecting power eleents with the digital control blocks, Ansoft-Siplorer is chosen as the siulation progra. In this progra, the power stage is fored by picking and placing the required coponents on a graphic window, which is called scheatic diagra. Then, PWM signals are produced by using control blocks. These PWM signals are used for switching the power switches, IGBT s. In the siulation, the switching devices are odeled by syste level odels of these eleents. The siulation results are displayed on a new graphic window. Beside of displaying output wavefors in a graphic window, Day Post Processor odule of the Siplorer can be used for analysing these wavefors in detail. The input power factor and the input current THD value can be calculated by using Day Post Processor. 3

48 The siulation odel circuit diagra of the power stage of the three-phase four-wire PWM rectifier is given in Figure 3.1. The utility grid is odeled as a three-phase Y- connected AC voltage-source with 0Vrs/phase (380Vrs line-to-line) and 50Hz ratings. 5 th, 7 th, 11 th, and 13 th haronic voltage-sources with different agnitudes are connected serially to the utility grid for obtaining a distorted utility odel. An LCL filter is used for the input filter of the three-phase rectifier. In the LCL filter, a daping resistor is connected serially to the input filter capacitance for passive daping of the resonance at the filter resonance frequency. The additional paraeters used in the syste are listed in Table 3.1. Table 3.1: Syste paraeters utilized in the siulation odel Grid side Inductor ( L1,L,L3 ) 40 µh Rectifier side Inductor ( L4,L5,L6 ) 800 µh LCL Filter Filter Capacitor ( C1,C,C3 ) 10 µf Daping Resistor ( R4,R5,R6 ) 5 Ω Inductance Resistor ( R1,R,R3 ) 10 Ω DC bus DC Voltage 750 V DC bus Capacitors ( C4,C5 ) 4 F Load Load Resistance 37.5 Ω Figure 3.1: Power stage of the Three-Phase Four-Wire PWM rectifier ake figure large The coputer siulations of the syste shown in Figure 3.1 are siulated for various operating conditions for the purpose of perforance investigation. The siulations are carried out by utilizing sufficiently sall integration step size in 33

49 order to iniize the coputational errors. The paraeters of the syste level switching device odels utilized in the siulations are given in Table 3.. The iniu integration step size is selected as 100 ns so that the siulation results are accurate. Table 3.: Syste level seiconductor device odel paraeters Device Forward Voltage (V) Bulk Resistance (Ω) Blocking Resistance (kω) Diode IGBT In the siulation odel, control algoriths are ipleented by using the control blocks of Siplorer. The control blocks of Siplorer are executed at a preset rate which could be different than the integration step size. In the three-phase rectifier syste, current control loop operates in every PWM cycle and the voltage control loop operates once in eight PWM cycle Siulation Results In this section, the perforance of the three-phase four-wire voltage-source PWM rectifier syste is evaluated by eans of the coputer siulations. Firstly, the gains of the voltage loop PI controller are found by the trial and error ethod. Then the resonant filter bank based controllers are designed and their control paraeters are found by the trial and error ethod. Once all the controllers are tuned, the design procedure is finished. Following the copletion of the controller design stage, the steady state perforance of the rectifier syste is evaluated under various utility grid conditions Trial and Error Based Voltage Controller Tuning Procedure For adjusting the voltage loop controller gains, the scheatic diagra of Figure 3. is used. In this scheatic diagra the utility grid is pure sine wave with low haronic content, and the current controller is chosen as a daped P+Resonant controller with only the fundaental frequency coponent. In the siulation, the switching frequency is set as 1.8 khz. The sapling rate of the DC bus voltage is chosen as 1.6 khz. If the voltage controller has a low bandwith, the voltage and current controller loops can be decoupled fro each other. This is the reason for choosing the voltage sapling rate very low. 34

50 Figure 3.: Scheatic diagra of the Three-Phase Four-Wire PWM Rectifier for adjusting voltage controller gains Since the voltage controller has a low bandwith, adjusting the controller gains are very easy. After a few siulations, taking care of the overshoot in the DC bus voltage and the settling tie, the proportional and integral gains can be adjusted Trial and Error Based Current Controller Tuning Procedure After adjusting the voltage controller gains, for adjusting the current loop controller gains, the scheatic diagra of Figure 3.3 is used. The sapling rate of the input currents is chosen as 1.8 khz, which is equal to the switching frequency. Resonant filter blocks that can be seen in Figure 3.3 (Res_50, Res_50 etc) are designed using the following equation; ( s cos( φ ) ω sin( φ )) Ki τ ωe GC ( s) = (3.1) s + τ ω s + ( ω e e e) The paraeters of the resonant filters are listed in Table 3.3 Table 3.3: Resonant Filter Paraeters K i Integral Gain 0 τ Daping Coeffient ω e Fundaental Frequency 100 π (rad/s) φ Phase Delay Angle.815 (degrees) 35

51 Figure 3.3: Scheatic diagra of the Three-Phase Four-Wire PWM Rectifier for adjusting current controller gains The resonant filters in the current controller of each phase are designed with the sae integral gain and paraeters at the beginning. The outputs of these filters are ultiplied by different gains. For the purpose of adjusting these gains, a highly distorted utility grid is used in Figure 3.3. In the first siulation only the proportional gain of the fundaental coponent resonant filter is nonzero. In this siulation, the gain of the fundaental coponent and the proportional gain of the controller are tuned. Then checking the haronic spectru of the input current, the proportional gain of the next resonant filter is added. Since each resonant filter is decoupled fro each other, the resonant filter gains are tuned in a sequence. Once the resonant filter for this haronic is built and tuned, this haronic is suppressed and the next doinant haronic is taken into consideration. This procedure continues until the highest intended haronic is suppressed to a sufficient degree. The ethod of tuning each filter coponent is based on the degree of iproveent. Every filter paraeter is increased to a point where it is no ore beneficial to further increase the 36

52 paraeter. The final paraeters of the voltage and the current controller are listed in Table 3.4. Table 3.4: Voltage and Current Controller Paraeters of the Three-Phase Four-Wire Voltage-source PWM Rectifier Voltage Controller Current Controller Proportional Gain 0.4 Integral Gain 100 Sapling Tie Proportional Gain us 50Hz_Gain Hz_Gain Hz_Gain Hz_Gain Hz_Gain 0.75 Sapling Tie us After adjusting the gains of the all resonant filters, the perforance of the threephase four-wire voltage-source PWM rectifier is tested by siulating the scheatic in Figure 3.3 under highly distorted, distorted, and undistorted utility operation. The rs values of the added voltage sources with different frequencies, for siulating the highly distorted, distorted and undistorted utility is given in Table 3.5. In all these siulations, daped and phase copensated type of the resonant filters, Eq. 3.1, are used in the current controllers at the beginning. Also, in order to see the perforance differences between different types of resonant filter fors, the scheatic in Figure 3.3 is siulated again using the resonant filter fors in Eq. 3. and Eq Table 3.5: Haronic contents of the voltage-sources used in the siulations Highly Distorted Utility Distorted Utility Undistorted Utility Fundaental Frequency, 50 Hz 0 V rs (100 %) 5 th haronic coponent 3.64 % 7 th haronic coponent 3.64 % 11 th haronic coponent 5.45 % 13 th haronic coponent 5.45 % Fundaental Frequency, 50 Hz 0 V rs (100 %) 5 th haronic coponent 1.8 % 7 th haronic coponent 3.18 % 11 th haronic coponent - 13 th haronic coponent 1.8 % Fundaental Frequency, 50 Hz 0 V rs (100 %) 5 th haronic coponent - 7 th haronic coponent - 11 th haronic coponent - 13 th haronic coponent - 37

53 38 ( ) ) ( ) sin( ) cos( ) ( e e e e i C s s s K s G ω ω τ φ ω φ ω τ + + = (3.) ) ( ) ( e e e i C s s s K s G ω ω τ ω τ + + = (3.3) Siulation Results of the Four Wire Rectifier under Higly Distorted Utility The siulation odel of the Three-Phase Four-Wire Voltage-source PWM Rectifier, Figure 3.3, is siulated using the paraeters in Table 3.3 and Table 3.4. The results of the siulations for the highly distorted utility are given in Figure 3.5 through Figure Highly distorted three-phase input voltage wavefors can be seen in Figure 3.4. With these fundaental and haronic coponents listed in Table 3.5, input voltage total haronic distortion, V THD, equals to 9.7 %. Figure 3.4: Steady state highly distorted input phase voltages

54 Figure 3.5: DC bus voltage and reference value for the variable daped and phase copensated resonant filter for under highly distorted utility Figure 3.6: Positive and negative half DC bus voltage for the variable daped and phase copensated resonant filter for under highly distorted utility 39

55 Figure 3.7: Input phase voltage and current at the instant of loading for the variable daped and phase copensated resonant filter for under highly distorted utility In Figure 3.5 to 3.7, the siulation results of the four-wire PWM rectifier eploying variable daped, phase copensated resonant filter for in the current controller under highly distorted utility can be seen. In Figure 3.5, the DC bus voltage reference and the easured value can be seen. As it can be seen fro Figure 3.5, when the syste is fully loaded at t = 130 s, the rectifier DC bus voltage decreases only 30 volts, and the proposed control algorith copensates this voltage decrease in less than two fundaental cycles. In Figure 3.6, the DC voltage wavefors of the two split capacitors that for the DC bus can be seen. The DC voltage levels of the two capacitors are equal to each other. In the transforerless UPS application, if the voltage levels of the two split capacitors are not equal to each other, DC offset occurs on the inverter output voltage wavefors. Thus, it is very iportant to assure the equality of the DC voltage levels of the two split capacitors in the four-wire PWM rectifier. In Figure 3.7, the rectifier input voltage and current wavefors at the instant of load transition can be seen. The rectifier input voltage and current is easured by volteter VM and aeter AM1 in Figure 3.. The controller reacts to the load change very rapidly and the input current stabilizes in a few iliseconds. 40

56 Figure 3.8: DC bus voltage and reference value for the constant daped and phase copensated resonant filter for under highly distorted utility Figure 3.9: Input phase voltage and current at the instant of loading for the constant daped and phase copensated resonant filter for under highly distorted utility In Figure 3.8 and 3.9, the siulation results of the four-wire PWM rectifier eploying constant daped, phase copensated resonant filter for in the current controller can be seen. In this case, the daping coefficient of each resonant filter for 41

57 different frequencies is chosen as equal to each other. As a result of this, the perforances of the resonant filters with higher resonance frequencies becoe unsatisfactory. As it can be seen in Figure 3.9, the ost doinant haronic frequencies of the input currents are the 11 th and the 13 th haronic coponents. Figure 3.10: DC bus voltage and reference value for the constant daped resonant filter for under highly distorted utility Figure 3.11: Input phase voltage and current at the instant of loading for the constant daped resonant filter for under highly distorted utility 4

58 In Figure 3.10 and 3.11, the siulation results of the four-wire PWM rectifier eploying only the constant daped resonant filter for in the current controller can be seen. Since the resonant filters used in the current controller of the rectifier are not phase copensated, phase delay of the syste results in a poor steady state perforance of the four-wire PWM rectifier. As it can be seen in Figure 3.11, all the haronic coponents are doinant in the three-phase input current wavefors. Although the input current wavefor is highly distorted, the dc bus voltage regulation is sufficient due to the high dc gain of the PI voltage controller. Table 3.6: Perforance coparison of three resonant filter fors under highly distorted utility Variable daped phase copensated Constant daped phase copensated Power Factor, PF Input Current Total Haronic Distortion, I THD Input Voltage Total Haronic Distortion, V THD % 9.7 % % 9.7 % Constant daped % 9.7 % In Table 3.6, the perforance coparison of the siulations with the three different resonant filter fors can be seen. The power factor values of the all three fors are in acceptable liits. However, the input current THD values of the constant daped, phase copensated for and the only constant daped for are very high. For the variable daped, phase copensated resonant filter case; the input current THD value is saller than the input voltage THD value. The input voltage haronic coponents and the current haronic coponents for different resonant filter fors can be seen in Table 3.7 through Table Table 3.7: Highly distorted input voltage haronic content Frequency (Hz) Percentage (%)

59 Table 3.8: Input current haronic content for the variable daped and phase copensated resonant filter for under highly distorted utility Frequency (Hz) Percentage (%) Table 3.9: Input current haronic content for the constant daped and phase copensated resonant filter for under highly distorted utility Frequency (Hz) Percentage (%) Table 3.10: Input current haronic content for the constant daped resonant filter for under highly distorted utility Frequency (Hz) Percentage (%) Siulation Results of the Four Wire Rectifier under Distorted Utility In Figure 3.13 and 3.14, the siulation results of the four-wire PWM rectifier eploying variable daped, phase copensated resonant filter for in the current controller under distorted utility can be seen. In Figure 3.13, the DC bus voltage reference and the easured value can be seen. In Figure 3.14, the rectifier input voltage and current wavefors under full load can be seen. 44

60 Figure 3.1: Steady state distorted input phase voltages Figure 3.13: DC bus voltage and reference value for the variable daped and phase copensated resonant filter for under distorted utility 45

61 Figure 3.14: Input phase voltage and current under full load for the variable daped and phase copensated resonant filter for under distorted utility In Figure 3.15 and 3.16, the siulation results of the four-wire PWM rectifier eploying constant daped, phase copensated resonant filter for in the current controller under distorted utility can be seen. Figure 3.15: DC bus voltage and reference value for the constant daped and phase copensated resonant filter for under distorted utility 46

62 Figure 3.16: Input phase voltage and current under full load for the constant daped and phase copensated resonant filter for under distorted utility In Figure 3.17 and 3.18, the siulation results of the four-wire PWM rectifier eploying only the constant daped resonant filter for in the current controller under distorted utility can be seen. Figure 3.17: DC bus voltage and reference value for the constant daped resonant filter for under distorted utility 47

63 Figure 3.18: Input phase voltage and current under full load for the constant daped resonant filter for under distorted utility In Table 3.11, the perforance coparison of the siulations with the three different resonant filter fors can be seen. As in the highly distorted utility case, the power factor values of the all three fors are in acceptable liits again. However, the input current THD values of the constant daped, phase copensated for and the only constant daped for are very high. For the variable daped, phase copensated resonant filter case; the input current THD value is saller than 4 %. Table 3.11: Perforance coparison of three resonant filter fors under distorted utility Variable daped phase copensated Constant daped phase copensated Power Factor, PF Input Current Total Haronic Distortion, I THD Input Voltage Total Haronic Distortion, V THD % 4.1 % % 4.1 % Constant daped % 4.1% 48

64 Table 3.1: Distorted input voltage haronic content Frequency (Hz) Percentage (%) Table 3.13: Input current haronic content for the variable daped and phase copensated resonant filter for under distorted utility Frequency (Hz) Percentage (%) Table 3.14: Input current haronic content for the constant daped and phase copensated resonant filter for under distorted utility Frequency (Hz) Percentage (%) Table 3.15: Input current haronic content for the constant daped resonant filter for under distorted utility Frequency (Hz) Percentage (%) Siulation Results of the Four Wire Rectifier under Undistorted Utility In Figure 3.0 and 3.1, the siulation results of the four-wire PWM rectifier eploying variable daped, phase copensated resonant filter for in the current controller under undistorted utility can be seen. In Figure 3.0, the DC bus voltage reference and the easured value can be seen. In Figure 3.1, the rectifier input voltage and current wavefors at the instant of load transition can be seen. 49

65 Figure 3.19: Steady state undistorted input phase voltages Figure 3.0: DC bus voltage and reference value for the variable daped and phase copensated resonant filter for under undistorted utility 50

66 Figure 3.1: Input phase voltage and current at the instant of loading for the variable daped and phase copensated resonant filter for under undistorted utility In Figure 3. and 3.3, the siulation results of the four-wire PWM rectifier eploying constant daped, phase copensated resonant filter for in the current controller under undistorted utility can be seen. Figure 3.: DC bus voltage and reference value for the constant daped and phase copensated resonant filter for under undistorted utility 51

67 Figure 3.3: Input phase voltage and current at the instant of loading for the constant daped and phase copensated resonant filter for under undistorted utility In Figure 3.4 and 3.5, the siulation results of the four-wire PWM rectifier eploying only the constant daped resonant filter for in the current controller under undistorted utility can be seen. Figure 3.4: DC bus voltage and reference value for the constant daped resonant filter for under undistorted utility 5

68 Figure 3.5: Input phase voltage and current at the instant of loading for the constant daped resonant filter for under undistorted utility In Table 3.16, the perforance coparison of the siulations with the three different resonant filter fors can be seen. As in the distorted utility case, the power factor values of the all three fors are in acceptable liits again. By using the variable daped, phase copensated resonant filter fors in the current controller, the fourwire voltage-source PWM rectifier exhibits superior perforance. The input current THD value is saller than 3.5 %. Table 3.16: Perforance coparison of three resonant filter fors under undistorted utility Variable daped phase copensated Constant daped phase copensated Power Factor, PF Input Current Total Haronic Distortion, I THD Input Voltage Total Haronic Distortion, V THD % 1.3 % % 1.3 % Constant daped % 1.3 % 53

69 Table 3.17: Undistorted input voltage haronic content Frequency (Hz) Percentage (%) Table 3.18: Input current haronic content for the variable daped and phase copensated resonant filter for under undistorted utility Frequency (Hz) Percentage (%) Table 3.19: Input current haronic content for the constant daped and phase copensated resonant filter for under undistorted utility Frequency (Hz) Percentage (%) Table 3.0: Input current haronic content for the constant daped resonant filter for under undistorted utility Frequency (Hz) Percentage (%) Siulation Results of the Four Wire Rectifier under Highly Distorted Utility with the Fundaental Frequency of 49.8 Hz In a three-phase rectifier application, usually the input voltage frequency is constant. However for soe unusual cases, the input voltage frequency can change slightly. For instance, in a three-phase UPS, usually the input voltage frequency tolerance is set as 50±.5 Hz. In this frequency range, the rectifier unit of the UPS ust operate without any proble. The resonant filters in the current controller of the three-phase four-wire rectifier are designed for specific frequency coponents. The gain of the filters at the resonant 54

70 frequency is very high, and for the other frequencies it is negligible. And if the input voltage frequency changes slightly, the gain of the each resonant filter decreases, as a result the perforance of the resonant filter bank degrades. This is the ain reason for using daped resonant filter fors in the resonant filter bank. In order to see the perforance difference between the different fors of the resonant filters, the scheatic in Figure 3.3 is siulated with the input voltage fundaental frequency is set as 49.8 Hz. Each haronic coponent frequency is set as ( 49.8) Hz. Siulation results are given in Figure 3.7 to 3.3. Figure 3.6: Steady state highly distorted 49.8 Hz input phase voltages 55

71 Figure 3.7: DC bus voltage and reference value for the variable daped and phase copensated resonant filter for under highly distorted 49.8 Hz utility Figure 3.8: Input phase voltage and current at the instant of loading for the variable daped and phase copensated resonant filter for under highly distorted 49.8 Hz utility 56

72 As it can be seen in Figure 3.8, the perforance of the variable daped and phase copensated resonant filter is sufficient. The input current THD value is saller than the input voltage THD value again. Figure 3.9: DC bus voltage and reference value for the constant daped and phase copensated resonant filter for under highly distorted 49.8 Hz utility Figure 3.30: Input phase voltage and current at the instant of loading for the constant daped and phase copensated resonant filter for under highly distorted 49.8 Hz utility 57

73 For the consant daped and phase copensated filter for, since the selectivity of the resonant filters for the 11 th and 13 th haronics increases, the perforance of these filters are insufficient. As it can be seen in Figure 3.30, the 11 th and 13 th haronics are the ost doinant coponents in the input current wavefor. Figure 3.31: DC bus voltage and reference value for the constant daped resonant filter for under highly distorted 49.8 Hz utility Figure 3.3: Input phase voltage and current at the instant of loading for the constant daped resonant filter for under highly distorted 49.8 Hz utility 58

74 For the constant daped resonant filter case, all haronic coponents are doinant. As it can be seen in Figure 3.3, the input current wavefor is uch distorted. Table 3.1: Perforance coparison of three resonant filter fors Power Factor, PF Input Current Total Haronic Distortion, Input Voltage Total Haronic Distortion, I THD V THD Variable daped % 9.7 % phase copensated Constant daped % 9.7 % phase copensated Constant daped % 9.7 % Table 3.: Highly distorted 49.8 Hz input voltage haronic content Frequency (Hz) Percentage (%) Table 3.3: Input current haronic content for the variable daped and phase copensated resonant filter for under highly distorted 49.8 Hz utility Frequency (Hz) Percentage (%) Table 3.4: Input current haronic content for the constant daped and phase copensated resonant filter for under highly distorted 49.8 Hz utility Frequency (Hz) Percentage (%)

75 Table 3.5: Input current haronic content for the constant daped resonant filter for under highly distorted 49.8 Hz utility Frequency (Hz) Percentage (%)

76 4. EXPERIMENTAL PERFORMANCE INVESTIGATION OF THE THREE PHASE FOUR WIRE VOLTAGE SOURCE PWM RECTIFIER 4.1. Introduction In this chapter, the experiental perforance investigation of the three-phase fourwire voltage-source PWM rectifier is conducted. The four-wire PWM rectifier, with the proposed control ethod, perforance is experientally investigated under steady-state and dynaic operating conditions. Thus, the theoretical study results of Chapter 3, and the coputer siulation studies of Chapter 4 are verified by the experiental work in this chapter. First the experiental syste hardware and software are described, and then the perforance investigations are given in detail. 4.. Hardware and Software set up of the Three Phase Four Wire Voltage Source PWM Rectifier The four-wire PWM rectifier is designed and anufactured at ENEL Enerji Elektronik San. Tic. A.S. laboratory. The block diagra of the experiental setup is given in Figure 4.1. The electrical power circuitry of the overall syste is shown in Figure 4.. The paraeters, which are utilized in the experiental setup and experients, are given in Table 4.1. Figure 4.3 through Figure 4.5 shows the laboratory prototype of the three-phase four-wire voltage-source PWM rectifier. 61

77 Figure 4.1: The syste block diagra of the experiental setup. Figure 4.: The electrical power circuitry of the overall syste. Table 4.1: Experiental syste paraeters Rectifier Rated Power 15 kva Input Frequency 50 Hz DC bus Voltage 750 Vdc Capacitors 4 F, 500 Vdc (Center-tapped) Grid Side Inductance 40 µh LCL Filter Converter Side Inductance 800 µh Filter Capacitor 10 µf Daping Resistor 5 Ω - 10 W Loading Circuit Three-Phase inverter load 10 Ω (each phase) 6

78 Figure 4.3: Main parts of the experiental set up. In Figure 4.3, the ain board, isolation board, PM50CLA10 odule, DC bus capacitors, busbar design and the switch ode power supply, SMPS, of the experiental set up can be seen. In Figure 4.4, input LCL filter, output LC filter, and the input current sensor board of the experiental set up can be seen. In Figure 4.5, the ain board and the ezdsp F81 board can be seen. The ain board supplies the scaled and filtered feedback signals to the ezdsp F81 board. Figure 4.4: The filtering eleents of the experiental set up. 63

79 Figure 4.5: Main board of the experiental syste. In the experiental syste, in order to protect the power switches a three-phase 30 A autoatic fuse and a three-phase 35 A circuit breaker are used at the utility grid connection point. A soft start circuit is used in the experiental syste for precharging the DC bus capacitors. Since the capacitance values of the DC bus capacitors are very high, they ay result in a high inrush current when the utility grid is connected to the syste by turning on the circuit breakers. In order to protect the diodes of the IPM fro this inrush current, the DC bus capacitors are precharged by using the soft start circuit. Before turning on the circuit breaker of the syste, the soft start switch ust be turned on. Then, after a few seconds, the soft start switch is turned off and the circuit breaker is turned on. The four-wire PWM rectifier is built by using Intelligent Power Modules, IPM s. Mitsubishi brand IPM odules, PM50CLA10 are utilized in the experiental syste. Intelligent Power Modules are advanced hybrid power devices that cobine high speed, low loss IGBTs with optiized gate drive and protection circuitry. Highly effective over-current and short-circuit protection is realized through the use of advanced current sense IGBT chips that allow continuous onitoring of power device current. Syste reliability is further enhanced by the IPM s integrated over teperature and under voltage lock out protection. The basic specifications of the intelligent power odule, PM50CLA10 are suarized in Appendix A. 64

80 The DC bus of the four-wire rectifier is constructed by utilizing a planar bus structure to obtain low parasitic inductance on the DC bus. Three aluinu plates are ounted on the DC bus capacitors for positive, negative, and the neutral DC bus rails. Between the positive and the negative busbar an insulation aterial having an insulation level of 3.3 kv is inserted. A snubber capacitor is installed between the positive and negative voltage terinal of the each IPM, so that the switching stress on the IGBTs due to the parasitic inductances is reduced. The utilized snubber capacitors in the syste have a. µf capacitance and 1000 V voltage rating. The isolation board is used for isolating the PWM signals produced by the ain board. The isolated PWM signals are directly applied to the PWM inputs of the IPM odule. Beside of isolating the PWM signals, the isolation board converts the PWM signals with +5/0 voltage levels produced by the ain board to an aplified +15/0 voltage levels, which is required for the IGBT turn-on and turn-off operations of the IPM. In the perforance tests, the four-wire PWM rectifier is loaded by the three-phase inverter which can be seen in Figure 4.. Since 750 Vdc and 0 A rated resistive load does not exist in the laboratory set up, a three-phase inverter load is used in the experiental syste. Each phase of the inverter is loaded by a 5 kw rated load bank, and each load bank consists of 5 parallel connected resistances which can be turned on and off via anual switches. The three-phase input voltages, input currents, and the DC bus voltage are easured for controlling the four-wire PWM rectifier experiental syste. The rectifier input currents are easured by utilizing the Hall Effect based current sensor, LA 55-P/SP1 odel anufactured by LEM. All the easured signals are scaled and filtered by utilizing basic operational aplifier circuits and passive noise filters in the ain board. A digital signal processor, TMS30F81, is used for controlling the four-wire PWM rectifier in the ain board. In the DSP unit, analog to digital conversions, control block calculations, and PWM output signal generation are carried out. In the experiental set up, the ezdsp F81 board anufactured by Spectru Digital Inc., is eployed. The features of the ezdsp F81 board and TMS30F81 are given in Appendix B. 65

81 The digital signal processor executes the control loops using the ADC feedback signals, and then it generates the necessary PWM signals. The control and PWM cycles are synchronized and they are executed at the sae rate. These goals are achieved by using the two event anagers, EVA and EVB, of the DSP. The TMS30F81 DSP chip utilized in controlling the UPS syste has event anagers which have tiers that can be used for generating the PWM signals. With each one of the, by utilizing a tier, three coparators and dead-tie generators, threephase odulation signals can be converted to six PWM logic signals. The control algoriths are executed within the event anager tier interrupt period. When the interrupt occurs, the DSP processes the interrupt service routine function to generate the PWM output signals for the IGBT switches. The flow chart of the interrupt service routine is shown in Figure 4.6. As seen fro the figure, the interrupt service routine starts with the A/D conversion of the feedback signals. Then, picking up the stored A/D signals fro the allocated eory locations, the signals are scaled and noralized to appropriate values. The voltage error signal, which is the difference between the reference DC bus voltage and the easured DC bus voltage, is fed to the PI type voltage controller. The current references are obtained by ultiplying the output of the PI controller with the sinusoidal signals, which are synchronized with the three input phases. These synchronized signals are produced by the phase lock loop, PLL. Using the synchronized signals guarantees the high input power factor of the four-wire PWM rectifier. The current error signals, which are the difference between the current references and the easured input currents, are fed to the discrete tie ipleented resonant filter bank of each phase. The outputs of the resonant filter banks for the reference PWM signals for each phase. After selecting the odulation ethod, the PWM unit of the DSP generates the rectifier gate logic signals. In the experients, the switching frequency is chosen as 1.8 khz, and the dead-tie is set to µs for the copleentary PWM logic signals, which is equal to the iinu dead tie requireent of the IPM odule. 66

82 Figure 4.6: The flowchart of the DSP progra 67

83 4.3. Experiental Results In this section, the perforance of the three-phase four-wire voltage-source PWM rectifier is evaluated experientally. Firstly, the voltage controller and the current controllers are prograed using the paraeters that are obtained in the previous chapter. Following the copletion of the controller design stage, the steady-state and the dynaic perforance of the four-wire rectifier is evaluated under 0 V rs, 50 Hz distorted utility. The haronic content of the distorted utility is given in Table 4.. Table 4.: Haronic content of the distorted utility 5 th haronic coponent.1 % 7 th haronic coponent 3. % 11 th haronic coponent 1.1 % 13 th haronic coponent 1.8 % The control syste paraeters which were tuned by eans of coputer siulations in the previous chapter are utilized in the experiental stage. The perforance of the controller for these paraeters are experientally evaluated and copared to the perforance obtained by siulations. Throughout the experiental studies the wavefor and power quality easureents are conducted with a LeCroy Waverunner 6100A four channel 1 GHz oscilloscope, and a Fluke 43B power analyzer. Current and voltage wavefor easureents are conducted with the LeCroy Waverunner 6100A, and the Fluke 43B is used for obtaining the haronic spectru, voltage-current THD, and the power factor values of the four-wire PWM rectifier. For the three-phase current easureent with Lecroy Waverunner 6100A, three Lecroy CP500 current transducers are used. For the isolated DC bus easureent, Lecroy ADP 305 voltage sensor is used. The experiental results of the three-phase four-wire voltage-source PWM rectifier are shown in Figure 4.7 through Figure In Figure 4.7, Figure 4.8, Figure 4.10, and Figure 4.11 CH1 is the total DC bus voltage wavefor, CH is the input phase-r current wavefor, CH3 is the input phase-s current wavefor, and finally CH4 is the input phase-t current wavefor. In Figure 4.9, and Figure 4.1 CH1 is the input phase-r voltage wavefor, and CH is the input phase-r current wavefor. 68

84 Figure 4.7: Input currents and DC bus voltage wavefors for the constant daped resonant filter bank case at the instant of full loading Figure 4.8: Steady state input currents and DC bus voltage wavefors for the constant daped resonant filter bank case under full load 69

85 Figure 4.9: Steady state input current and voltage wavefor for the constant daped resonant filter bank case under full load In the perforance tests of the experiental set up, firstly the constant daped resonant filter for is used in the current controller of the four-wire PWM rectifier. In this for of resonant filter, the phase delays on the feedback signals are not copensated. Beside of this, the selectivity of the resonant filter increases with the increasing frequency. As a result, the perforance of the resonant filters in each frequeny coponent of the current controller becoes poor. The poor perforance of the four-wire PWM rectifier can be seen in Figure 4.7, 4.8, 4.9. In Figure 4.7, three-phase input currents and the DC bus voltage wavefors at the instant of full load transition can be seen. Since the DC voltage is regulated by the PI type controller with the paraeters obtained in the siulation results, the regulation perforance is perfect. In Figure 4.8, the steady-state three-phase input currents and the DC bus voltage wavefors under full load operation can be seen. Again in this figure, the DC voltage regulation is perfect; however, the phase currents contain high haronic content. This is the result of the poor resonant filter perforance. In Figure 4.9, the steady state input voltage and current wavefors can be seen. 70

86 Figure 4.10: Input currents and DC bus voltage wavefors for the variable daped and phase copensated resonant filter bank case at the instant of full loading Figure 4.11: Steady state input currents and DC bus voltage wavefors for the variable daped and phase copensated resonant filter bank case under full load 71

87 Figure 4.1: Steady state input current and voltage wavefor for the variable daped and phase copensated resonant filter bank case under full load operation In the second part of the perforance tests, the variable daped and phase copensated resonant filter for is used in the current controller of the four-wire PWM rectifier. As a result, the perforance of the resonant filters in each frequeny coponent of the current controller becoes perfect. In Figure 4.10, three-phase input currents and the DC bus voltage wavefors at the instant of full load transition can be seen. In Figure 4.11, the steady state three-phase input currents and the DC bus voltage wavefors under full load operation can be seen. Both the DC voltage regulation and the input current regulation of the four-wire PWM rectifier are perfect with this type of resonant filter for. In Figure 4.9, the steady state input voltage and current wavefors can be seen. 7

88 Figure 4.13: Input power and the power factor for the constant daped resonant filter bank case under full load operation Figure 4.14: Input voltage haronic content for the constant daped resonant filter bank case under full load operation Figure 4.15: Input current haronic content for the constant daped resonant filter bank case under full load operation 73

89 In Figure 4.13, 4.14, and 4.15, power analyzer results of the four-wire PWM rectifier, which eploys the constant daped resonant filter for in the current controller, can be seen. The input power factor of the four-wire PWM rectifier is 0.99, which satisfies the IEC, IEEE standards. However, the input current THD of the four-wire rectifier is 8.1%. As it can be seen fro the current haronic spectru, the 5 th, and the 7 th haronic is the ost doinant haronic coponent. The 5 th and 7 th haronics in the input voltage wavefor is the ain reason of this. Since the resonant filters in the current controller does not work properly, the 5 th, 7 th haronic voltage coponents can not be copensated in the current wavefor. Figure 4.16: Input power and the power factor for the variable daped and phase copesated resonant filter bank case under full load operation Figure 4.17: Input voltage haronic content for the variable daped and phase copensated resonant filter bank case under full load operation 74