WITH the vast rise in parallel multicore architectures, the

Transcription

1 JOURNAL OF LIGHTWAVE TECHNOLOGY, VOL. 34, NO. 12, JUNE 15, Comprehensive Design Space Exploration of Silicon Photonic Interconnects Meisam Bahadori, Student Member, IEEE, Sébastien Rumley, Member, IEEE, Dessislava Nikolova, and Keren Bergman, Fellow, IEEE, Fellow, OSA Abstract The paper presents a comprehensive physical layer design and modeling platform for silicon photonic interconnects. The platform is based on explicit closed-form expressions for optical power penalties, derived for both signal-dependent and signalindependent noise contexts. Our models agree well with reported experimental measurements. We show how the modeling approach is used for the design space exploration of silicon photonic links and can be leveraged to optimize the wavelength-division multiplexed (WDM capacity, evaluate the scalability, and study the sensitivity of the system to key device parameters. We apply the methodology to the design of microring-based silicon photonic links, including an evaluation of the impairments associated with cascaded ring modulators, as well as the spectral distortion and crosstalk effects of demultiplexer ring arrays for nonreturn-to-zero (NRZ ON OFF keying (OOK modulated WDM signals. We show that the total capacity of a chip-to-chip microring-based WDM silicon photonic link designed with recently reported interconnect device parameters can approach 2 Tb/s realized with NRZ-OOK data modulation and 45 wavelengths each modulated at 45 Gb/s. Index Terms Microring resonators, optical interconnects, silicon photonics. I. INTRODUCTION WITH the vast rise in parallel multicore architectures, the scalability of computing performance is increasingly reliant on the availability of high-bandwidth, energy efficient data communications infrastructure. Silicon photonics has attracted considerable interest [1] [3] as an emerging technology that can offer close integration of CMOS electronics with optical devices [4], [5]. This technology has the potential for delivering ultra high bandwidth interconnect solutions that are at the same time energy efficient and available at low cost. Silicon photonics has been proposed for designing efficient networks-on-chip [6] [10], for supporting global interconnects for datacenters [11], and high performance computing systems [12], [13]. It has been also presented as an enabling technology for realizing Exascale computing systems (i.e., Supercomputers capable of realizing Manuscript received July 22, 2015; revised October 4, 2015; accepted November 16, Date of publication November 22, 2015; date of current version June 1, This work was supported by DARPA Microsystems Technology Office under the Computing With Optically Enabled Data Movement Project and the Power Efficiency Revolution for Embedded Computing Technologies Program, by the U.S. Department of Energy, National Nuclear Security Administration, Advanced Simulation and Computing program under Contract PO with Sandia National Laboratories, and by Freedom Photonics, Inc. The authors are with the Department of Electrical Engineering, Columbia University, New York, NY USA ( mb3875@columbia.edu; sr3061@columbia.edu; dnn2108@columbia.edu; kb2028@columbia.edu. Color versions of one or more of the figures in this paper are available online at Digital Object Identifier /JLT operations per second [14], [15]. The chip-scale integration of electronics and photonics may further alleviate the limitations of electronic chips in terms of I/O pin count [16], [17]. Although multiple experimental demonstrations of silicon photonic devices, sub-systems or systems [18] [21] have been reported in the last decade, the technology is still young and only recently emerging for commercial systems adoption. Significant design space exploration is required to understand the performance and impact of silicon photonics for computing interconnects. The design methodology for implementing electro-optical interfaces at the transmitters and receivers must be developed. In particular, it is critical to understand how silicon photonic optical modulators, multiplexers, and demultiplexers can be combined to obtain optimized, thus efficient, WDM links. In this paper, we present a uniquely comprehensive modeling platform for efficiently exploring the design space of silicon photonic interconnects, from the physical layer to the link-level analysis. We concentrate our efforts on microring-resonator (MRR-based links, as they offer the highest bandwidth density and most energy efficient performance among current silicon photonic interconnect devices [22] [24]. MRRs can be designed to perform several interconnect functions and represent the key building blocks of silicon photonic systems [25], [26]. MRRs can be used both as active modulators and switches, as well as wavelength-selective passive filters. When acting as modulators [27] [33], they leverage plasma dispersion effects of silicon [34] [36] which provides an efficient way of changing the effective refractive index of the ring by injecting or depleting charge carriers. Such modulation scheme has proven to be capable of delivering high-speed modulations, e.g., 10 Gb/s or higher with low energy consumption [22], [24], [37] [41]. When used as demultiplexers, a ring operates as a passive drop filter whose resonance is tuned to a specific channel wavelength [23], [42], [43]. In this case, fine-tuning is carried out via integrated heaters that are implemented on top of the silicon microring [44], [45]. First order or higher order add-drop filters have been fabricated and demonstrated in recent years [46], [47]. MRRs are generally very sensitive to thermal fluctuations [48], [49], but several stabilization mechanisms (e.g., wavelength-locking schemes have been proposed in recent years to redress this vulnerability [50] [55]. MRRs can also switch multiple wavelengths from one waveguide to another and spatially route them to another path [56] [60]. Due to their small size, many microrings can be cascaded on a single on-chip bus waveguide, facilitating the dense wavelength-division-multiplexing design and operation of the link [61] [63]. However, WDM links may suffer from spectral IEEE. Personal use is permitted, but republication/redistribution requires IEEE permission. See standards/publications/rights/index.html for more information.

2 2976 JOURNAL OF LIGHTWAVE TECHNOLOGY, VOL. 34, NO. 12, JUNE 15, 2016 Fig. 1. Chip-to-Chip silicon photonic interconnect with an MRR-based WDM link. The optical interface of the transmitter chip includes MRR modulators that act upon carrier dispersion principle for high speed modulation. The optical interface at the receiver includes Demux filters, photodetectors, and electronic decision circuitry (Det: detector. Wall-plug efficiency corresponds to the electrical to optical power conversion of the laser. degradationof channels and inter-channel crosstalk [64] [72]. These impairments eventually put an upper limit on both the modulation speed of each channel and on the number of channels, thus placing an upper bound on the maximum aggregate rate of the link [73], [74]. In this paper, we present the design and modeling platform for obtaining the maximum achievable aggregate bandwidth in a short-distance silicon photonic link with microring modulators and filters. This is achieved by estimating the effects of the induced impairments and translating them into power penalties, i.e., an amount of extra optical power that permits compensation for the effects of such impairments on the bit-error-ratio (BER performance of the system [75] [79]. We also show how the spectral statistics of a modulated light changes when it goes through a ring demultiplexer and how that can be turned into power penalty. The rest of this paper is organized as follows: in Section II, we set forth our approach in analyzing the optical power budget of a silicon photonic link and provide a brief review of the most important characteristics of non-return-to-zero ON OFF keying (NRZ-OOK modulation and estimation of BER-associated power penalties. The analysis is based on how the statistics of the optical signal changes as it undergoes modulation and filtering. Sections III and IV provide details of the power penalties associated with ring modulators and demultiplexers. Several available experimental measurements of power penalties are presented to back-up theoretical estimations. In Section V, we present our discussion of optimizing the ring parameters in order to achieve minimum power penalty throughout the MRR-based link. In Section VI, the link analysis is given for a realistic power budget. Finally, conclusion is drawn in Section VII. Two appendices are also provided with details of obtaining equations presented in the paper for microring demultiplexers. II. DESCRIPTIONOFTHEAPPROACH A. Optical Power Budget and Power Penalties Consider a simple chip-to-chip silicon photonic link as shown in Fig. 1. On the transmitter side, cascaded microrings along an on-chip waveguide modulate the incoming multi-wavelength light generated by a comb laser source [80]. The incoming wavelengths, once imprinted with data, are then transmitted through an optical fiber to a receiver chip. The receiver chip consists of multiple passive microrings whose resonances are tuned to channel wavelengths. The total capacity of this link is obtained by multiplying the number of channels (wavelengths with the modulation bit rate. Intuitively, it is tempting to maximize the number of wavelengths, and/or to choose higher bit rates for each channel. This allows for higher utilization of the available spectrum in the transmission media. However, as either the number of wavelengths and/or the bit rate grows, crosstalk and other undesired impairments emerge and eventually prevent reliable transmission. The total capacity of the link is closely tied to the optical power losses and impairments that the light experiences over the entire link. Summing up all the power penalties of the link, PP db, for a single channel, the following inequality must hold [81]: [ P dbm laser 10 log 10 (N λ ] P dbm sensitivity PP db tot. (1 In general, aggregated optical power P laser (sum over all wavelengths must stay below the nonlinear threshold of the silicon waveguides at any point of the link [70], [74]. In contrast, the signal powers should stay above the sensitivity of the detectors P sensitivity (minimum number of photons or equivalently a certain amount of optical power at the receive side. A typical receiver may have a sensitivity of 12.7 dbm at 8 Gb/s operation [77], while a good receiver may exhibit a sensitivity of about 21 dbm at 10 Gb/s [75]. The difference between lower and higher thresholds results in the maximum power budget that can be exploited. This budget accounts for the power penalty per channel (how much higher should the signal hit the detector PP tot, and for the product over the N λ channels. We will show that the power impairments induced by the rings depend on the spacing between channels, which is inversely proportional to the number of wavelengths, and on the modulation rate r b.eq. (1 is therefore a nonlinear function of N λ and r b, and determination of the N λ r b product that maximizes the bandwidth directly from it is intractable. It is however possible to optimize the link, i.e., optimize its constituting devices, such that PP tot is minimized for different (N λ, r b combinations. We proceed to do this optimization for NRZ-OOK modulation format. B. Properties of the NRZ-OOK Modulation Format In an NRZ-OOK modulation, the electrical signal is a sequence of logical 0 s and 1 s that are imprinted on the envelope of the light. Therefore, the envelope electric field of the modulated

3 BAHADORI et al.: COMPREHENSIVE DESIGN SPACE EXPLORATION OF SILICON PHOTONIC INTERCONNECTS 2977 Fig. 2. Envelope electric field of the (a NRZ-OOK modulation, and its (b power spectral density. ER is the ratio of the optical power of the high level to the optical power of the low level of the electric field. light has a form of E(t = k= a k p(t kt b (2 where T b is the bit interval and its inverse (r b =1/T b isthebit rate of the modulation, a k is the amplitude for 0 s and 1 s bits, and p(t is the ideal square pulse of unit amplitude and duration T b. A picture of such envelope is shown in Fig. 2(a, where the high level is denoted by A and the low level is denoted by B. Ideally, if the modulation has an infinite extinction ratio (ER, then the low level must be zero (B = 0; however, in practice ring modulators produce a limited ER. Typical numbers for 10 G devices are 8 10 db of extinction [24]. It is a practical assumption that all bits are equiprobable and the source of information that generates these bits is a stationary ergodic source. Therefore Prob(a k = A = 0.5 and Prob(a k = B = 0.5. The average behavior, i.e., statistics, of such modulation is given by the autocorrelation of the envelope electric field over time. The average power spectral density of the envelope of modulation is then the Fourier transform of its autocorrelation, which simplifies to the following equation [82], [83]: where S E (f =μ 2 δ(f f c +σ 2 1 r b sinc 2 ((f f c /r b, (3 μ = A + B, 2 (4a σ = A B 2 (4b f c is the optical frequency of the light, and δ(f f c is the Dirac delta function centered at carrier frequency f c. A plot of this spectrum is presented in Fig. 2(b. The power of the modulated signal is given by integrating Eq. (3 over all frequencies, yielding P av = μ 2 + σ 2. We note that the power of the NRZ-OOK modulation can be decomposed based on the mean and variance of the envelope electric field. The μ 2 power is at the center frequency (represented by the delta function and the power of the variance (σ 2 is spread over a spectrum of frequencies (represented by the sinc function. We will leverage Eq. (3 to estimate the power penalty of each optical component along the link. C. Power Penalty Evaluation Based on Signal Statistics As the modulated light goes through the waveguides, transmission medium, and other optical devices, the quality of the signal degrades. To evaluate the breadth of the degradation, one approach consists of evaluating the new high and low levels (assumed to be A and B when the light reaches the receiver. Alternatively, one can analyze how the power spectral density experiences distortion due to the non-ideal frequency response of the link. In other terms, one can evaluate how μ and σ evolve into μ and σ. Once these latter values are known, the power penalty associated with the distortion can be derived using one of the two following models widely used in literature. If the receiver is limited by a signal-independent noise (SIN mechanism (such as thermal noise that does not beat against the optical signal, and considering a Gaussian distribution for the uncertainty of high and low levels, the power penalty to get the same BER performance and compensate for the signal degradations is estimated as [84] ( σ PP SIN μ 10 log 10. (5 σ μ In contrast, in amplified systems the noise is mainly due to the amplified spontaneous emission (ASE and that depends on the intensity of the input light to the amplifier (signal dependent noise, SDN. Using the model proposed by Downie [85] that relates the variance of the noise to the high and low levels of the electric field, the power penalty is estimated as { 10 log10 (μ /μ, if σ /μ > 1 PP SDN (6 10 log 10 (σ /σ, if σ /μ 1. This equation implies that the power penalty in this case is related to the worst-case decrease of μ or σ. If the center frequency loses more power than the spectral frequencies, then σ /μ > 1 and the power penalty is due to the change in μ. If the spectrum loses more power than the center frequency, then σ /μ < 1, and the penalty is due to the change in σ. In accordance with Eq. (5 and (6, two approaches can be taken to obtain power penalties: 1 Estimate the high and low levels (A and B first, and then find μ and σ using equations (4a and (4b. This is the approach we take for characterizing penalties of ring modulators. 2 Directly estimate μ and σ based on μ, σ, and the frequency response of the optical device. This is the approach we take for the demultiplexing ring filters. Also note that in this paper, we mainly work with SIN model and assume no amplification in the link. If amplification is to be considered in the receiver, as it is the case in the experiments against which this work has been compared, Eq. (6 should be employed instead of Eq. (5. III. MICRORING MODULATORS MRRs are capable of modulating light at very high speeds while operating at low powers. This property combined with the small footprint of these optical modulators turn them into very promising candidates for integrated photonics. The physics behind the operation of such modulators has been well estab-

4 2978 JOURNAL OF LIGHTWAVE TECHNOLOGY, VOL. 34, NO. 12, JUNE 15, 2016 These can be inserted into Eq. (5 or (6, depending on the noise conditions. Considering a dominant SIN in the system, so Eq. (5, the total modulation power penalty is given by ( r 1 PP SIN P ( mod Pmod Mod 10 log log r +1P 10 (8 mod P in Fig. 3. (a Schematic view of a microring modulator. The high-speed electrical signal is applied to the p-n-junction embedded inside the silicon ring. (b Modulation of the input laser by shifting the resonance of the ring to create high and low levels of optical power at the output. (c Graphical view of how ON and OFF states of the light at the output happen. (d Time-domain presentation of a non-ideal NRZ OOK modulation. lished [30]. Just like all other practical modulators, microrings are not capable of providing ideal modulation and thus some impairments and penalties on the optical signal are inevitable. In this section, we provide a general description of the power penalties that are of the most importance. As depicted in Fig. 3, the electric field envelope of the unmodulated input light has constant amplitude that we denote by A, and constant power P in = A 2. The modulation action occurs by shifting of the resonance of the ring as shown in Fig. 3(b. An ideal NRZ-OOK ring modulator traps the input light depending on whether the data bit is 1 or 0 (shown in Fig. 3(c. The mean and variance of this ideal modulation (from equations (4a and (4b is μ = A/2, and σ = A/2, which gives the average power P mod = μ 2 + σ 2 = A 2 /2=P in /2. We note that 10 log 10 (P mod /P in =3 db. As data bits are assumed equiprobable, this means obfuscating half of the power. A non-ideal modulation differs from the ideal modulation in the way that the high level is A instead of A (A <A and the low level is B instead of zero, so that P mod = μ 2 + σ 2 = (A 2 + B 2 /2 as plotted in Fig. 3(d. This causes the power penalty to grow beyond the 3 db of an ideal modulation. If the transmission of the ring for bit 0 and 1 at the laser wavelength is denoted by T 0 and T 1, then A = T 1 and B = T0. T 0 and T 1 can be estimated by considering a Lorentzian shape for the resonance spectrum of the ring [73]. The ER of modulation is the ratio of power of bit 1 to the power of bit 0, i.e., r = A 2 /B 2. Using equations (4a and (4b again, the mean and variance of the non-ideal modulation levels can be written as μ = B ( r +1/2, and σ = B ( r 1/2. After some further algebraic manipulations 1 the following relations are obtained: μ r +1 P μ = mod, r +1 P mod σ r 1 P σ = mod. (7 r +1 P mod 1 In particular, by rewriting levels in terms of average modulation powers A = 2P mod and B = 2P mod /(r +1. where the second term is the 3 db corresponding to the ideal modulation. By eliminating P mod and noting that P mod =(r + 1A 2 /2r, the power penalty can be rewritten as ( r 1 (A PP SIN 2 Mod 10 log log r log 10 ( r+1 2r P in. (9 This formulation is helpful as it permits to distinguish three contributing power penalty factors. The first term is the power penalty due to the finite ER of modulation as widely used in the literature [30]. The second term is the modulator insertion loss (IL, defined as the difference between the input power and the power of bit 1. Finally, the third term is the penalty due to the OOK nature of the imperfect modulation. For an ideal modulator (infinite ER and no loss, this last term is 3 db while the first two terms are reduced to zero. If a silicon photonic link is dominated by the ASE noise that beats against the optical signal, the power penalty of modulator is estimated from Eq. (6 by noticing that (μ /μ>(σ /σ. The result is PP SDN Mod 10 log 10 ( r 1 r +1 5 log 10 ( P mod 2P in. (10 The first term in this equation is the power penalty due to the finite ER of modulation (which is different from the one in Eq. (9 and the second term represents the average power loss due to the combination of OOK and IL of modulator. Now, we consider several ring modulators mounted on a silicon waveguide, each one modulating a specific wavelength (as in Fig. 1. Cascading all the ring modulators on a single bus waveguide induces two additional power penalties. The first one to be accounted for is the passive cumulative IL of the laser power of each channel as it passes by other rings. If the spacing between wavelengths is small and the number of modulators is large, this could become a significant loss. The second one is the array-induced multiplexing crosstalk [86]. Multiple wavelengths of the comb laser pass by each ring, but only one of them gets modulated. Due to the carrier injection or depletion of the p-n junction of the ring, the spectrum of the ring is switched between two resonance frequencies as shown in the inset of Fig. 4: f 0 (OFF state and f 1 (ON state, where f 1 >f 0 (blue shift. If this shift (Δf = f 1 f 0 is too large, then the shifted spectrum may capture some of the power of the neighboring channel that passes by the ring, hence inducing an intermodulation crosstalk. If f Δ represents the channel spacing and f Δ > 0, then a neighboring channel at f 0 + f Δ would suffer higher crosstalk than a neighboring channel at f 0 f Δ. This leads to write the

5 BAHADORI et al.: COMPREHENSIVE DESIGN SPACE EXPLORATION OF SILICON PHOTONIC INTERCONNECTS 2979 Fig. 4. PP Mod-Xtalk (see Eq. (11 for two cases: (blue when the adjacent channel is at a higher frequency (f Δ > 0, (red when the adjacent channel is at a lower frequency (f Δ < 0. Due to the unilateral shift of the ring modulator, the first case exhibits higher penalty than the second case. Measurement data [86] are denoted by circles and squares. worst-case penalty associated with the induced crosstalk as ( (2ζ/FWHM 2 + q 0 PP Mod-XTalk 5log 10 (2ζ/FWHM 2 (11 +1 where ζ = f Δ Δf for f Δ > 0, and ζ = f Δ for f Δ < 0. In the above equation, FWHM is the half-power bandwidth of the ring and q 0 is considered as the extinction of the ring resonance in the OFF state (q 0 = T min /T max, where the ring is assumed to be close to critical coupling operation [87]. This power penalty is obtained by assuming a Lorentzian shape for the resonance of the ring. Using Eq. (11, we reproduce the results of intermodulation crosstalk reported by Padmaraju et al. [86] for a microring modulator with Q 6000 at the resonance wavelength nm and q The reproduced results are plotted in Fig. 4 where circles and squares are penalties from experimental data for f Δ > 0 and f Δ < 0, respectively. At this point, we have introduced the main ingredients to estimate the total power penalty induced by a ring-resonator based modulator array. Knowing the parameters Q, coupling coefficients of ring-waveguides, and loss of each ring, its associated Lorentzian transform can be obtained, from which stems A and B. These permit to evaluate Eq. (9 or (10, the passive ILs, as well as the intermodulation crosstalk penalty. IV. MICRORING DEMULTIPLEXERS Consider an NRZ-OOK modulated signal passing through MRR based filter as shown in Fig. 5(a. The filter has two outputs: the drop port and the through port. Considering this filter as a linear time independent system, it can be characterized by two frequency responses H drop (f and H through (f that relate the output electric field present at each port to the input electric field. Knowing that the input power spectral density is S E (f, we can calculate the output power spectral density as S out (f= S E (f H(f 2 at each port [82]. Unless H(f is constant for all f, which is not realizable with ring resonators, some degree of spectral distortion will appear. This translates into eye closure in the time domain as shown in Fig. 5(b. At the receiver, the received signal is sampled in each bit interval. We consider this Fig. 5. (a Schematic representation of a demux ring filter. The input NRZ modulation is characterized by the mean and variance of its high and low levels. The dropped signal is a distorted NRZ modulation that can be characterized by new mean and variance parameters. (b Comparison between the input NRZ modulation to a filter and its output response. The output is a signal that effectively has new high and low levels A and B with different mean and variance than the original signal. This can be seen by inspecting the eye-diagrams of the input and output optical signals. as a reconstruction of the NRZ-OOK signal with new average high and low levels denoted by A and B, and new statistical mean μ and variance σ 2. Using Eq. (3, the change in the mean and variance by going through the H(f system can be written as ( μ 2 H(f c 2 (12a μ ( σ 2 σ H(f 2 1 sinc 2 ((f f c /r b df. r b (12b The relation between μ and μ is approximately independent of the shape of the filter since the power associated with the mean of the two levels is centered at a single frequency. On the other hand, the relation between σ and σ depends on the shape of the filter, indicating spectral distortion to the power spectrum of the signal. Fig. 6(a shows demux ring resonators at the receive side of a WDM silicon photonics link, where each one is tuned to drop one of the NRZ-OOK modulated channels. Each ring has an intrinsic loss that corresponds to a decay time-constant denoted by τ i. The coupling rates and the coupling decay time constants for the ring-waveguide interactions are denoted by (ξ 1, τ 1 and (ξ 2, τ 2 for the through and drop paths as depicted in Fig. 6(b. If the quality factor of the rings is high enough and the coupling of ring-waveguide is weak, one can employ the equations of the temporal coupled mode theory for MRRs [88]. By virtue of the conservation of energy in the ring-waveguide system, the relation between (ξ 1, τ 1 and (ξ 2, τ 2 are then given as ξ 1,2 τ 1,2 and the drop and through field transfer functions

6 2980 JOURNAL OF LIGHTWAVE TECHNOLOGY, VOL. 34, NO. 12, JUNE 15, 2016 Fig. 6. (a Graphical view of microring demultiplexers at the receiver. (b Intrinsic and coupling decay rates of a ring add-drop filter. (c Transmission spectrum of the drop path of a demux ring with the 3-dB bandwidth denoted as FWHM. f Δ is the possible detuning between the resonance and the channel. are H drop (Ω ξ 1 ξ 2 1 jω+1/τ (13a H through (Ω jω+1/τ (13b jω+1/τ where Ω=2π(f f 0 is the frequency deviation from the resonance of the ring (f 0, 1/τ =1/τ i +1/τ 2 +1/τ 1, and 1/τ =1/τ i +1/τ 2 1/τ 1. The full width at half maximum of the resonance is calculated as FWHM = 1/(πτ, and the quality factor of the resonance is given by Q = f 0 /FWHM. The power transfer function from the input to the drop port is H drop (Ω 2, which has a Lorentzian line shape as depicted in Fig. 6(c. The main parameters of this function are the resonance frequency of the ring (f 0, 3-dB bandwidth (FWHM, and the peak drop power (p 0 =(ξ 1 ξ 2 τ 2. Once the drop and through shapes of the filter are known, the next step consists of looking at the impact that the drop path has on the spectrum of an NRZ-OOK modulation. This means evaluating the integral of Eq. (12b for a Lorentzian function. The result is given by γ = 1 1+β 2 1 ( 1 exp( 2πν(1 jβ 2πν Re (1 jβ 2 (14 in which ν = f 0 /(2Qr b, β =2Qf Δ /f 0, and Re(... indicates the real part of the complex number. Details of calculations are provided in Appendix I. Here, f Δ denotes any possible detuning between the channel and the resonance of the demux ring as shown in Fig. 6(c. We note that the first term of Eq. (14 is not a function of bit rate, reflecting only the filtering nature of the device at a distance f Δ from its resonance. If there is no detuning between the channel and the filter, then f Δ = 0, β = 0, and γ is reduced to γ =1 (1 exp( 2πν/(2πν. The second term, on the other hand, depends on the bit rate, highlighting the fact that the bandwidth of the NRZ-OOK signal is a contributor to the spectral distortion imposed by the ring filter. If modulation speed is low enough, the second term is negligible compared to the first term. Fig. 7. (a Comparison of the estimated spectral distortion (solid curve and measured penalties [71] (circles for various bit rates. (b Estimation of the penalty due to small detunings between the resonance of the ring and the channel wavelength. Measurements are taken from [72]. (c Simulated eye-diagrams of the received signal for 10 and 20 Gb/s data rates. (d Schematic view of two adjacent channels with a fair amount of spectral overlap. A low-q ring will result in more crosstalk effect, but less spectral distortion (highlighted areas on the NRZ spectrum. f Δ denotes the spacing between channels. We proceed to employ Eq. (14 to evaluate the power penalty of the filtering action. By introducing Eq. (13 into equations (12a and (12b, we obtain (μ /μ 2 p 0 /(1 + β 2 and (σ /σ 2 p 0 γ. From there, using Eq. (5, we have PP SIN Demux 10 log 10 (p 0 5 log 10 ( 1 1+β 2 5 log 10 (γ (15 for a link with SIN. The first term in PP is simply the drop IL of the resonance spectrum of the ring. The second term reflects the IL at the channel wavelength, and the third term reflects the penalty of spectral distortion due to the filtering. If the link is dominated by an SDN, power penalty is estimated from Eq. (6 as PP SDN Demux 5log 10 (p 0 5 log 10 (γ. (16 Fig. 7(a shows the measured power penalties (circles reported by Biberman et al. [71] (at BER = 10 9 foraringof a bandwidth about 10 GHz and four different bit rates: 5 Gb/s (PP 0.3 db, 7.5 Gb/s (PP 0.5 db, 10 Gb/s (PP 0.7 db, and 12.5 Gb/s (PP 1.15 db. The solid curve on this figure is the estimated power penalty up to 60 Gb/s (PP 3.3 db with f Δ = 0 (exact tuning between the channel and the ring filter.

7 BAHADORI et al.: COMPREHENSIVE DESIGN SPACE EXPLORATION OF SILICON PHOTONIC INTERCONNECTS 2981 The model provides a good estimation of the power penalties and predicts a penalty of 3 db at about 50 Gb/s. Fig. 7(b presents the measured penalties (circles of small detunings (up to 6 GHz reported by Lee et al. [72] (at BER = 10 9 for a ring resonator with FWHM = 9.6 GHz (Q and a bit rate of 10 Gb/s (PP 0.8 db at zero detuning. The solid curve represents the estimated BER penalties for a link with SDN mechanism (see equation (16. The SDN is selected based on the fact that the experiment used optical amplification to compensate for power loss in the system. Here again, we observe good agreement between the measurements and calculations. These two comparisons against measurements let us posit that Eq. (14 can serve as a basis for quantifying the rate-dependent penalties due to sideband truncation of an NRZ-OOK channel. Fig. 7(c illustrates the effect of the spectral distortion on the eye-diagram of the received signal for 10 and 20 Gb/s data rates. As it can be observed, the eye-diagram of the 20 Gb/s signal experiences more closure in accordance with higher power penalty. Besides the spectral distortion of each channel, power penalties also arise in the demultiplexing process in the form of crosstalk. Due to the infinite tail of the Lorentzian resonance and spectral overlap of the channels, capturing and dropping some of the optical power from adjacent channels is unavoidable, as shown in Fig. 7(d. We assume that the worst-case crosstalk happens at the first demux ring in Fig. 1 [66], since all the channels are at their maximum power and none has been dropped yet. The power that is captured by the Lorentzian tail of the ring from the spectra of the adjacent channels has a negative impact on the statistics of the high and low levels at the detector and increases the BER. In order to compensate for this, the crosstalk effect on the BER is turned into a power penalty. The total electric field at the drop port of each demux ring is a superposition of the electric field of the tuned channel and all other crosstalk channels. The worst case of crosstalk happens if all the channels are within the bandwidth of the electronic filter ( ω i ω 0 /2π <B e at the receiver (coherent crosstalk. Here we can use Eq. (14 again to characterize the strength of the electric field ( E(t 2 of the crosstalk channels at the drop port. We also consider a random phase difference between the main channel and each crosstalk channel [89]. In this case, the worst-case of powers of bits 1 and 0 (due to the random phase difference have the following thresholds: P min 1 =1+ i P max 0 = i γ i γ i 2 i γi (17a (17b where γ i s are estimated for each crosstalk channel from Eq. (14. The eye-closure penalty then follows from Eq. (5 (SIN or Eq. (6 (SDN as PP SIN Demux-XTalk 10 log 10 (P min 1 P max 0 (18a ( PP SDN Demux-XTalk 10 log 10 P min 1 P max 0. (18b Fig. 8. Comparison of measured crosstalk power penalties (circles and model (solid line for two cascaded ring resonators at a bit error rate of For channel spacings larger than 1 nm ( 120 GHz, the penalty is negligible. However, penalty sharply increases as the channel spacing is reduced to below 40 GHz. Measurement data are taken from [66]. These equations along with Fig. 7(d show that penalty of crosstalk between channels depends on the bandwidth of the rings, spacing between channels, and bit rate of each channel. Experimental measurements for the crosstalk penalty between two rings at BER = have been recently presented in Ref. [66] with a ring bandwidth (FWHM of 30 GHz and a bit rate of 25 Gb/s (structure shown in the inset of Fig. 8. We compare our estimations to these measurements in Fig. 8 where the solid curve is the estimated power penalty and the circles represent the measured power penalties. A reasonable agreement is observed between measurements and the model. The model tends to overestimate the crosstalk penalty but this is acceptable in a link design context, where one aims at identifying the worst-case scenario. At this point, we can estimate the worst-case power penalties inflicted by the demux array. Worst-case of crosstalk happens at the first ring, while the worst-case of waveguide loss happens for the last ring. Equations (15 and (16 estimate the impact of truncation while Eq. (18 predicts the effect of crosstalk. Using these equations and the ones presented in the previous section for the modulator side (see Eq. (9 (11, we can proceed to the next steps, which consist of optimizing the ring parameters for minimum power penalty (see Section V and evaluating the ideal link format (N λ r b for maximum bandwidth (see Section VI. V. OPTIMIZATION OF MICRORINGS FOR MINIMUM POWER PENALTY In all of the calculations to follow, we presume an extreme FSR of 50 nm for the rings (which corresponds to approximately 4 μm diameter of the rings. It is also assumed that there is no amplifier in the link, thus the link is dominated by a SIN mechanism. For the modulators, the parameters subject to optimization are Q-factor of the rings, shift of the resonance due to the plasma dispersion effect, and the spacing between channels. Considering these parameters, trade-offs then should be made between modulator IL, ER penalty, OOK penalty, cumulative arrayed-induced IL, and the multiplexing crosstalk. Fig. 9 shows an example of calculating the total penalty of the ring modulator for a case where channel spacing is 1 nm (hence 50 rings and

8 2982 JOURNAL OF LIGHTWAVE TECHNOLOGY, VOL. 34, NO. 12, JUNE 15, 2016 Fig. 9. Estimated modulator penalty for a channel spacing of 1 nm and different resonance shifts. If the shift of the resonance is small, then the IL and ER penalty are high. On the other hand, if the shift is too big, then the crosstalk penalty is very high. The optimum shift is about half of the channel spacing. the quality factor is either 6000 (low-q, solid curve or (high-q, dotted curve. It is assumed that the rings are critically coupled to the waveguide. As this plot indicates, power penalty of a modulator is always higher than 3 db (which is the penalty of an ideal NRZ-OOK modulator and the minimum value of the penalty occurs when the shift of the resonance of the ring is approximately half of the channel spacing. Unfortunately, such a big shift of the resonance usually requires high peak-to-peak voltages for the electrical signal that drives the p-n-junction of the ring (according to the formulation presented in [73] and [76]. Consequently, for CMOS-compatible voltages (V pp 1 2 V [74], [90], the resonance shift is away from its optimum value and this leads to higher penalties on the modulators. For example in Fig. 9, a shift of about 0.5 nm leads to a penalty of 3.7 db for Q = 6000 and a shift of 0.07 nm ( 8.5 GHz leads to about 9.7 db penalty. It is important to note that here we make the assumption that the ring modulators are driven with pre-emphasized NRZ signals to effectively compensate for the limited bandwidth of the modulator [40]. Hence, we always consider that the static ER can be achieved under dynamic drive. With this assumption, higher Q factor for the ring modulators is more desirable as it reduces the penalty of the modulator for a fixed amount of resonance shift. This fact can be seen on Fig. 9. For the ring demultiplexers, the parameters subject to optimization are the Q-factor of the rings, bit rate of channels, and channel spacing. Trade-offs are then established among drop IL, spectral distortion, and crosstalk penalties. As an example, Fig. 10(a shows the optimized overall penalty of demux rings for 50 channels with 1 nm spacing between them, various bit rates, and various Q-factors. This penalty is the sum of the spectral distortion (see Eq. (15, the cross-talk penalty (see Eq. (18a and ring drop IL. The latter is function of the intrinsic loss of the ring α loss and Q-factor (as developed in Appendix II. Note that the intrinsic loss is not subject to optimization: the minimal value that fabrication can provide should simply be employed to limit IL. In contrast, the Q-factor can clearly be optimized if the objective is to minimize the overall power penalty. Looking at Fig 10(a, one notes that if the Q of the ring is small, crosstalk has a big impact on the penalty. On the other hand, if the Q is too high, then the spectral distortion penalty and Fig. 10. Optimizing the penalty of demux ring filters by including the IL as a function of Q-factor of the ring. (a Loss factor of the ring is 1 db/cm and 50 channels. (b Loss factor of the ring is 8 db/cm and 50 channels. (c Power penalty for different number of channels and bit rates (loss = 1 db/cm when using demux rings with optimized Q. (d Optimum penalty and its corresponding Q-factor for an aggregate rate of 1 Tb/s and different channel rates (loss = 1 db/cm. IL have higher impacts. Therefore, there is a minimum point for the power penalty at an optimum Q-factor. The dependence of the optimum penalty and optimum Q on the bit rate is emphasized on Fig. 10(a with the dotted black curve showing how the optimum point behaves as the bit rate increases. This indicates that higher bit rates lead to wider filters in order to compensate for higher spectral distortion. In Fig. 10(b, same optimization is done for the demux rings but with 8 db/cm ring loss [66]. It is clear that in this case, high-q rings will lead to a high IL. In Fig. 10(c we plot the penalty of different bit rates and different number of channels for optimally set Q-factors. It is seen that penalty tends to increase as the spacing between channels decreases. Finally, we perform the optimization on microring demux filter with the assumption of having a fixed amount of total aggregate rate (1 Tb/s is assumed. In this case, increasing the number of channels, e.g., from 10 to 100, is accompanied by a decrease in the modulation speed. The results are shown in Fig. 10(d where six cases are clearly marked on the plot. With optimum design of the rings, the penalties result in a narrow range db. Meanwhile, the corresponding quality factors can vary over a wide range from 1800 to VI. SCALABILITY ANALYSIS OF A MICRORING-BASED SILICON PHOTONIC LINK The scalability of a silicon photonic link in terms of the maximum total aggregate rate depends on the available optical power budget. Based on the characterization of power penalties for ring modulators at the transmitter and ring demultiplexers at the receiver, it is possible to maximize the achievable aggregation. For each particular bit-rate, the analysis is carried out by finding the maximum value of N λ such that Eq. (1 holds. This

9 BAHADORI et al.: COMPREHENSIVE DESIGN SPACE EXPLORATION OF SILICON PHOTONIC INTERCONNECTS 2983 Fig. 11. Calculation of the available optical power at the output of the transmitter and minimum required power at the input of the receiver. Both calculations reflect the worst-case. The difference is the available power budget for losses and penalties associated with fibers, or switches that are placed between transmitter and receiver. The capacity of the link is estimated to be 1.2 Tb/s. Quality factor of ring modulators is taken to be and intrinsic loss factor α loss to be 1 db/cm. is done by incorporating power penalties from Eq. (9 or (10, (11, (15 or (16, and (18 into the PP tot term in Eq. (1. Some of the power penalties of the link do not generally depend on the bit-rate of channels or on the number of channels/channel spacing. Examples include the waveguide propagation loss ( 1 db/cm and chip-to-fiber/fiber-to-chip coupling IL ( 1 db [74]. The penalties of the modulators are also approximately independent of bit-rate, but depend on the number of channels due to array-induced IL and inter-modulation crosstalk. This can be taken as an encouragement to use higher bit-rates at the modulators. However, higher bit-rates limit the sensitivity of the receiver circuitry which, as we will see, eventually puts a cap on the maximum achievable bandwidth. Overall, this engenders an interesting design space to explore. We initiate our explorations by calculating the available power at the output of the transmitter chip and the minimum required power at the input of the receiver as a function of number of channels for 10 Gb/s NRZ-OOK modulation, as shown in Fig. 11. It is assumed that laser can provide a maximum power of 5 dbm for each wavelength (after coupling into the waveguide, but the total amount of the optical power that is injected into the silicon waveguide is kept below 20 dbm to avoid the nonlinearities of the silicon waveguide. The sensitivity of the receiver for each channel is set to a value of 15.5 dbm (given by a rate-dependent model, to be described later. These assumptions set the limit of the maximum power budget to 20.5 db for each 10 Gb/s channel. From the transmitter side, we add up the modulator penalties (loss, ER, crosstalk and the waveguide loss, assuming 1 db/cm loss for the silicon waveguide. Since the number of required modulators and demultiplexers is equal to the number of channels, the length of the waveguides linearly scales up with the number of channels. Then, we add up the worst-case penalty associated with the demultiplexers (including worst-case crosstalk and worst-case waveguide loss. We also account for a 2 db extra margin to compensate jitter in clocking. In contrast, we do not account for power penalties related to chromatic dispersion. We assume this last to be negligible for short distance links with modulation rates < 100 Gb/s. Fig. 11 shows that at about 120 channels, the power budget Fig. 12. Estimated sensitivity of the receiver as a function of the modulation data rate. At low data rates, dark current dominates over the noise current and thus the sensitivity is almost constant, whereas at high data rates the noise current dominates over the dark current and sensitivity is reduced. completely closes and thus the scalability of this link for 10 Gb/s channels is limited to about 1.2 Tb/s. For lower number of channels, there is some budget remaining. It can be exploited to introduce additional elements in the link (more couplers, switches, etc.. Alternatively, launching powers per channel can be reduced to limit the power consumption of the laser. Next, the dependence of the scalability of the link on the modulation speed is investigated. In this case, we still assume the same condition on laser power (5 dbm per channel, maximum 20 dbm total, but a bit-rate dependent model for the sensitivity of Germanium-based detector is employed. The model for the sensitivity is based on the simple definition of the BER for OOK modulation with Gaussian noise at the receiver. Therefore, sensitivity is defined by the following equation: ( I dark +Inoise RMS P sensitivity 10 log 10 Q BER R 10 ER/ ER/10 1 (19 where Q BER is the argument of the complementary error function (Q BER = 7 for BER = 10 12, R is a responsivity [A/W] of the photodiode (assumed 0.7 A/W, ER is the extinction ratio of input signal [db] (assumed 10 db as a reference, I dark is the dark current [ma] (assumed 1 μa and I noise is the total RMS input-referred noise current [ma] of the receiver [19], [91] [95]. For simplicity, we also assume that the receiver has enough bandwidth to accommodate the designated bit rate (BW 0.75 DataRate. The noise current is estimated from an inputreferred RC equivalent circuit for the receiver (photodiode + TIA whose 3 db bandwidth is set to the bandwidth of the receiver [19]. Fig. 12 shows a plot of the obtained sensitivity for BER = The sensitivity is dominated by dark current at low data rates, whereas at high data rates is dominated by the noise current. A sensitivity of 15.5 dbm is then obtained for 10 Gb/s NRZ OOK modulation. We identify the highest number of channels supported for each modulation rate up to 120 Gb/s (see Fig. 13. By evaluating the link format (N λ r b product, it is found that a chip-to-chip silicon photonic link, under the considered assumptions, can support a peak aggregated bandwidth of 2.1 Tb/s, composed of 47 channels at 45 Gb/s. This corresponds to a channel spacing of 1.06 nm (i.e., 125 GHz and a modulator shift of 0.53 nm. The

10 2984 JOURNAL OF LIGHTWAVE TECHNOLOGY, VOL. 34, NO. 12, JUNE 15, 2016 Fig. 13. Plot of estimated maximum capacity of the link based on the power penalties of the link and the modulation rate of each channel. It is seen that the link can support up to 2.1 Tb/s at a modulation rate of 45 Gb/s. If 10 Gb/s modulation is used, the total capacity is 1.2 Tb/s as shown in Fig. 11. breakdown of the power penalties in this case includes 5.56 db for the modulator array (Eq. (9 + Eq. (11 + cumulative IL of the array, 1.6 db for waveguide loss and coupling at the transmitter side, negligible fiber loss and dispersion effects (considering a short-reach application, 1.2 db for waveguide loss and coupling at the receiver, 3.5 db for the optimized demultiplexing array, and 2 db for jitter penalty. The overall power penalty at 45 Gb/s for 47 channels then sums up to 13.8 db which is equal to the total power budget per wavelength (3.4 dbm laser power for each channel, and 10.4 dbm receiver sensitivity. Overall, this result is to be considered with care, as the generation and detection of 45 Gb/s electric signal might be problematic and energy consuming. Nevertheless, it is a good indicator of the general capabilities of silicon photonics ring resonators from an optical point-of-view. VII. CONCLUSION The paper provided a comprehensive design and modeling platform for a detailed link level analysis of a silicon photonic WDM link realized with microrings. It characterized power penalties of microring modulators and presented a closed-form approximation for the spectral distortion of a microring filter for NRZ-OOK modulated signals. The theoretical estimations for the penalties of spectral distortion and their dependence on the modulation speed (bit-rate were compared and verified against experimental measurements. Crosstalk effects at the transmitter and the receiver were estimated and we showed that the models for crosstalk agree with the experimental measurements. Based on the presented models, we explored optimal device parameters to minimize the overall power penalties of the link. Our design space exploration indicates that with the current state of the art photonic components and optimal microring parameters, the best possible modulation rate for each channel, and the total aggregate link rate are approximately 45 Gb/s and 2.1 Tb/s, respectively. APPENDIX I In accordance with Eq. (12b and Eq. (13, the variance of the NRZ-OOK modulation changes by getting dropped by the Fig. 14. Selected contour in the upper half-plane for evaluating the integral of the spectral distortion penalty. There is a pole at the origin and a pole at Z 0 = F Δ + jν. ring as (σ /σ 2 = p 0 γ where sinc 2 (F γ = 1+((F F Δ /ν 2 df (A1 ν = FWHM/(2r b, and F Δ = f Δ /r b. In order to calculate this integral, one has to employ the residue theorem. Rewriting the integral in the form of γ = ν2 2π 2 Re 1 exp(j2πf F 2 ((F F Δ 2 + ν 2 df (A2 and choosing the appropriate contour in the upper half plane shown in Fig. 14 indicates that there is only one pole at Z 0 = F Δ + jν inside this contour. The closed-loop integral is then calculated as 1 exp(j2πz Z 2 ((Z F Δ 2 + ν 2 dz = π νz 2 0 (1 exp(j2πz 0. (A3 Obviously, due to the higher-order denominator, the integral over C( is zero. The integral over C(ε where ε 0 on the other hand is not zero and must be calculated. Substituting Z = ε exp(jθ into the integrand of Eq. (A3, will then yield the following result C (ɛ 1 exp(j2πz 2π2 Z 2 ((Z F Δ 2 + ν 2 dz = Z 0 2. (A4 Combining the above equations finally leads to γ = ν2 Z ν ( 1 2π Re exp(j2πz0 Z0 2 which can also be cast as Eq. (14. APPENDIX II (A5 Considering the Lorentzian function presented in Eq. (13, the goal is to minimize the IL (make it as close as possible to 0 db of the drop path of the demux ring. For convenience, we introduce the normalized rate parameters as x =(1/τ 1 /(1/τ i and y =(1/τ 2 /(1/τ i. For a demux ring, the drop IL is H drop (0 2 =4xy/(1 + x + y 2. This expression is completely symmetric for both x and y. Considering a given value for y, equating the derivative of the H drop (0 2 with respect to x to zero yields x opt =1+y which is exactly the critical coupling condition (input coupling rate is equal to the intrinsic loss rate plus the output coupling rate for a ring resonator. Substituting x opt then gives